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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 10, OCTOBER 2011 2753

Transient Improvement by Window Transient

Enhancement and Overshoot Suppression Techniques

in Current Mode Boost Converter

Yean-Kuo Luo, Chao-Chang Chiou, Chun-Hsien Wu, Ke-Horng Chen, Senior Member, IEEE,

and Wei-Chou Hsu

Abstract—In this paper, a current mode boost converter using window transient enhancement (WTE) and overshoot suppression (OSS) technique is presented for digital still camera (DSC) appli-cations. The peak-to-peak transient overshoot voltage demand of a DSC motor driver is generally within 4%–5% of the regulated value. However, conventional boost converters usually fail to pass this criterion during large load transient. The OSS technique re-duces the overshoot voltage when load current changes from heavy to very light. Experimental results show that compared with the use of a conventional current mode boost converter, the use of the technique reduces drop voltage about 62% and overshoot voltage about 51% when the load current has a load step of 400 mA. More-over, the settling time improves to 43%, which is better than in the conventional case of a 400 mA load current step. The overhead of the silicon area is about 4.5% to achieve the overshoot reduction. The estimated high performance demonstrates that it is suitable for DSC applications.

Index Terms—Boost converter, constant frequency regulation, dc-dc power converter, free-wheeling switching, on-chip compen-sation, OSS technique, overshoot suppression.

I. INTRODUCTION

O

VER the past few years, portable electronic devices, such as digital still cameras (DSC), have become very popular. Today, cameras come in smaller sizes but boast of more powerful features that require highly integrated power solutions. Gener-ally, the power supply of a DSC is divided into several parts, which include the motor driver, system logic and input/output (I/O), double-data-rate-two synchronous dynamic random ac-cess memory (DDR2 SDRAM), central proac-cessing unit, charge-coupled device, CMOS sensor, and backlight unit [1]. Each part has its own specifications under specified application condi-tions. Here, a boost converter is designed to supply an output of 4.5–5 V to a motor driver, with a loading range between 0

Manuscript received October 20, 2010; revised December 21, 2010; accepted February 10, 2011. Date of current version September 21, 2011. This work was supported by the National Science Council, Taiwan, under Grant NSC 99– 2220-E-009-003 and Grant NSC 98-2622-8-009-014-A2. Recommended for publication by Associate Editor D. Maksimovic.

Y.-K. Luo and W.-C. Hsu are with the Institute of Microelectronics, National Cheng Kung University, Tainan 701, Taiwan.

C.-C. Chiou, C.-H. Wu, and K.-H. Chen are with the Institute of Electrical Control Engineering, National Chiao Tung University, Hsinchu 300, Taiwan (e-mail: khchen@cn.nctu.edu.tw).

Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2011.2116807

and 300 mA, or 400 mA. Aside from the static specification demands, dynamic transient performance, such as peak-to-peak overshoot is also important. The reason is that a motor driver is commonly used to drive the zoom lens or audio frequency shutter. The supply voltage of the motor driver can affect the driving speed of the zoom lens. Therefore, an overshoot in the supply voltage may lead to inaccurate lens movement or focus error.

Generally, a peak-to-peak overshoot at a specified loading transient should be within 4%–5% in order to not affect sys-tem performance. The right half-plane zero decrease the syssys-tem bandwidth when approaching low frequencies; it even causes instability issues at heavy loads and low-battery conditions [2]. Thus, a conventional boost converter with a limited bandwidth limited by the RHP zero usually suffers from a large transient overshoot. Many control techniques have been presented to improve transient performance in the design of buck convert-ers [3]–[16]. External components or slave transient enhance-ment systems are used to improve transient performance. How-ever, these require more off-chip components or printed-circuit board space and are more costly [5]–[7]. Some methods require special conditions to maintain effective operation. For example, the V2 control is used for fast transient but needs a large time constant composed of a large output capacitor and equivalent se-ries resistance (ESR). As a result, it suffers from a larger output voltage ripple, deteriorating the system performance [8]–[10]. A low-ESR ceramic capacitor is commonly used because of its low cost. However, it is not suitable for such design.

Other control methods used to improve transient performance may not be suitable to reduce overshoot in the design of boost converters [11]–[13]. For example, some control methods dis-charge redundant energy by turning ON the low-side power MOSFET. The control mechanisms used in boost converters may also lead to a bad overshoot, when load current changes from heavy to light [13]. The dynamic frequency control or slew rate enhancement in error amplifiers may solve the drop during current change from light load to heavy load. Unfortunately, these methods are not very effective in overshoot reduction when the load current changes from heavy to light load or very light load [11], [12]. In this paper, the proposed converter with the window transient enhancement (WTE) and overshoot sup-pression (OSS) techniques can efficiently reduce the overshoot problem at the cost of about 4.5% extra increase of silicon area. Section II describes the WTE and the OSS techniques to improve transient response and reduce the overshoot problem.

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2754 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 10, OCTOBER 2011

Fig. 1. Proposed current mode boost converter.

The description of the WTE technique is illustrated in Section III. The proposed OSS technique is illustrated in Section IV. The experimental results are presented in Section V. Finally, conclusions are given in Section VI.

II. PROPOSEDWTEANDOSS CONTROLTECHNIQUES

A conventional boost converter usually operates in discontin-uous conduction mode (DCM) at light loads. In case of a sudden large load current, a large dip in output voltage appears due to the limited bandwidth and poor recovery ability of the DCM operation. However, the extra energy induces a large overshoot voltage at the output once the load current changes from heavy to light. Unfortunately, large transient voltage variations will cause a malfunction in the next stage. The suppression of the transient output variation becomes more important.

To improve transient response in conventional boost convert-ers, this paper introduces the OSS technique and the WTE tech-nique, which is depicted in Fig. 1 [14]. The function schematic includes three main parts, namely, the original pulsewidth mod-ulation controller with a feedback network, the WTE controller, Oscillator and frequency selector, and the light-load detection used to start the OSS control mechanism.

A large variation in the output voltage caused by light-to-heavy or light-to-heavy-to-light load current transient can be detected by the feedback detection circuit. The hysteretic comparator COMP1 with a positive input reference of (VR EF-VH Y S) for

detecting sudden heavy load conditions, sets its output to high to turn ON the WTE controller. The controller then temporarily changes the compensator at the output, VEA, of the error

am-plifier to speed up the charge accumulation so that the inductor current increases rapidly to supply the heavy load. Moreover, the switching frequency is increased during the period TLH when

a drop condition is detected by the comparator, COMP1, at the feedback node. The increase of switching frequency avoids

sys-Fig. 2. Waveforms with and without the WTE technique.

tem instability when the compensator is changed and the new cross-over frequency comes too close to one-tenth of the orig-inal switching frequency. The WTE controller is turned OFF beyond the period TLH. Besides, the temporarily higher

switch-ing frequency is stopped and the compensator returns back to its original value for stable operation. Similarly, the heavy-to-light load current transient induces an overshoot voltage, which is detected by the hysteretic comparator COMP2, to turn ON the WTE controller. The controller then temporarily changes the equivalent compensator network to speed up the decrease in the ramp process of the EA output VEA. As a result, it decreases the

inductor current faster or even stops the switching. The value of VH Y Sis set to 2% of the VR EF, and thus, the feedback detection

voltage is in the range of 0.98∗ VR EF–1.02∗ VR EF. The VH Y S

voltage is set to 1% or less to obtain a better transient perfor-mance. However, a very small VH Y S value, when it is smaller

than the output ripple or the switching noise observed at the output, leads to malfunction.

The operation of the WTE technique is presented in Fig. 2. The red dotted line denotes the feedback voltage, VFB, in a

conventional boost converter. The correspondent VEA,

repre-sented by the pink dotted line, illustrates the slow transient response due to a limited system bandwidth. The blue solid line represents the behavior of the proposed WTE technique. Fast transient response is achieved because of the fast response of the correspondent VEA, which is represented by the green solid

line. The inductor current waveform with and without WTE technique is represented in solid and dotted line, respectively. During the periods TLH and TLH, the compensator is changed

by the WTE technique in the proposed converter to achieve fast transient response.

Although the WTE technique is applied to improve transient performance, overshoot problems are not solved completely us-ing only this technique. The reason is that the WTE technique is triggered when the feedback voltage exceeds VR EF+ VH Y S,

where an overshoot condition has already occurred. Thus, al-though the WTE technique prevents the inductor current from rising too quickly, it cannot discharge the excess voltage that

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LUO et al.: TRANSIENT IMPROVEMENT BY WTE AND OSS TECHNIQUES IN CURRENT MODE BOOST CONVERTER 2755

Fig. 3. (a) Proposed current mode boost converter with WTE technique. (b) Operation of Oscillator and frequency selector during light-to-heavy load tran-sient period TL H.

has already appeared at the output. Thus, a long recovery time is needed for the output to return to its steady state. Usually, a dummy load is used to reduce the settling time. Although a dummy load is applied to discharge energy, it inevitably causes loss in efficiency when an inappropriate dummy load is selected. Hence, the OSS technique is applied to replace the dummy load mechanism.

III. IMPLEMENTATION OF THEPROPOSEDWTE TECHNIQUE

The operation of the WTE technique is shown in Fig. 3(a). Generally, the transistor P34 is turned ON in a steady state un-less a large load transient is detected by the feedback detection circuit. When this scenario occurs, an off-time pulse, which is generated by logic operation, turns OFF the transistor P34 for a few microseconds. Off-time pulse at the C node is the output of the one-shot circuit. This off-time pulse temporarily mod-ifies the capacitance multiplication ratio to two times smaller than the original six times the value of the on-chip capacitance C1 [3], [4], [15], [16]. In other words, the effective small ca-pacitance results in a faster slew rate. The system bandwidth is also extended temporarily. Thus, transient performance is im-proved by the proposed mechanism. Fig. 3(b) shows that the oscillator speeds up the switching frequency clock only during the light-to-heavy load transient period TLH.

The proposed circuit of the WTE technique is shown in Fig. 4(a). The off-time pulse generation circuit is not shown for simplification. The EA implemented by a simple structure is composed of transistors P21–P25 and N21–N24. As previ-ously mentioned, transistors, P11–P17 and N11–N17, function as a rail-to-rail unity gain buffer X1. The rail-to-rail unity gain buffer responds from the ground to a maximum positive voltage without problems in dc-biasing operation. Transistors P31–P34

Fig. 4. (a) Transistor level of the WTE technique. (b) Oscillator and frequency selector for improving transient response in the WTE controller.

and the unity gain buffer X1 function as capacitance multiplica-tion. The capacitance multiplication ratio is decided by the ratio of the transistor size of P31 to the summation size of transistors P32 and P33. In this work, the size ratio of transistors P31-P33 is 1:1:4. According to the theory published in [3], [4], [16], effective capacitance can be boosted to six times the C1 capac-itance. The gates of transistors P31–P33 are connected to the ground, and the drain-source voltages are tracked by the unity gain buffer X1. Transistors P31–P33 have the same dimensions and a well-matching layout to ensure their equivalent threshold voltages.

Fig. 4(b) illustrates the internal oscillator and the frequency selector. When the WTE controller output, VW T E, is pulled high

temporarily, the internal clock is increased to a higher value to cause a larger charge/discharge current on capacitance C7 when the transistor N42 is turned ON. The signals VH and VLdefine

the boundary voltage of the sawtooth, which is generated from the reference voltage VR EF. The operational amplifiers X2–X4

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2756 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 10, OCTOBER 2011

Fig. 5. Operation of the proposed OSS technique of a boost converter.

IV. OPERATION OF THEPROPOSEDOSS TECHNIQUE

The OSS technique is turned ON when the inductor current is lower than the IL (Light). The cost is lower efficiency

perfor-mance when converter operates at the light condition. At heavy load condition, the inductor is higher than IL (Light) and the

converter operates in the continuous conduction mode (CCM) mode, as same as the conventional boost converter. When con-verter operates in CCM mode, the transistor P2 is always turned OFF and transistor P1 is turned ON after transistor N1 is turned OFF every cycle.

Once the OSS technique is turned ON and the operation of OSS technique is shown in Fig. 5. During subinterval 1, The term “subinterval” has been left italicized at the first instance, and changed to Roman at other instances in the text per style guide. Kindly check if it is OK. as shown in Fig. 5(a), the power NMOSFET N1 is on and ramps up the inductor current, whereas transistors P1 and P2 are OFF. This operation of subinterval 1 is the same as that of a conventional boost converter.

In subinterval 2, as shown in Fig. 5(b), all the power transistors are turned OFF, whereas transistor P1 is turned OFF, and the inductor current passes through the body diode of transistor P1 to charge the output capacitor. The period of subinterval 2 is fixed, which is different in comparison with that of the pseudocontinuous conduction mode control [15]. The maximum period of subinterval 2 is 20%–25% of the duty cycle in this work.

As shown in Fig. 5(c), the free-wheel transistor P2 turns ON and shorts the inductor to dissipate the extra energy in subin-terval 3. When transistor P2 is turned ON, the inductor current

Fig. 6. (a) Waveforms with and without OSS technique. (b) Normalization of excess charge with and without OSS technique.(c) Chart of total excess charge.

Fig. 7. Light load detection.

flows through transistor P2 and returns to the inductor circular. Through this method, the redundant energy is consumed by the finite resistance of transistor P2. The inductor avoids the energy to be transmitted to the output at the same time.

The concept of OSS technique is explained in two parts. First, it blocks the inductor current from transferring to the converter output. This is achieved using the free-wheel switching control. The free-wheel period becomes subinterval 3 in the proposed operation as illustrated in Fig. 6(a). Free-wheel switching traps the inductor current in a closed loop with a finite resistance. During the free-wheel period, redundant current is consumed by the finite resistance. Power dissipation is proportional to the inductor current. There is a tradeoff between power conver-sion efficiency and transient recovery time. Second, it restrains energy from transferring to the output by fixing the operation period of subinterval 2, as shown in Fig. 6(a). As a conventional

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LUO et al.: TRANSIENT IMPROVEMENT BY WTE AND OSS TECHNIQUES IN CURRENT MODE BOOST CONVERTER 2757

Fig. 8. (a) AC analysis of conventional boost converter at heavy load of 250 mA. (b) AC analysis of the proposed converter at heavy load with the activated WTE technique.

boost converter still transfers stored energy in the inductor to the output under an overshoot condition, a fixed period of subinter-val 2 is a good choice to restrain the energy and avoid system instability due to a very small value of subinterval 2.

Fig. 6(a) shows the inductor current waveforms of a converter with or without the OSS technique. The current waveform of a conventional boost converter is represented by the black solid line, and that of a boost converter with the OSS technique is rep-resented by the red solid line. When the inductor current is lower than IL (light) and the light-load detection is triggered, the OSS

technique is enabled. The main difference between the proposed converter with OSS technique and conventional boost converter is the charge transferred to the converter output. In Fig. 6(a), the yellow areas, A1–A3, represent the transferred charge of a conventional boost, and the blue areas, B1–B3, indicate that of the proposed converter. Referring to A1, the A1–A3 and B1–B3 areas after normalization are shown in Fig. 6(b). In comparison, the total excess charge is shown in Fig. 6(c). These results show that the excess charge sent by the proposed OSS technique to the

output is less than half of a conventional boost converter. They demonstrate that the overshoot voltage of the OSS technique is smaller than that of a conventional design.

Moreover, the period of subinterval 2 is decided by the equa-tion, as shown in (1)

Tsubinterval 1+ Tsubinterval 2+ Tsubinterval 3 = Tp erio d. (1)

Tsubinterval1, Tsubinterval2, and Tsubinterval3 indicate the time

periods of subinterval 1, subinterval 2, and subinterval 3, respec-tively. When a converter operates in a steady state, the relation of subinterval 1 and subinterval 2 is expressed in (2)

(Tsubinterval 1+ Tsubinterval 2) = (VO U T/VIN)∗ Tsubinterval 2.

(2) Usually, in DSC applications, the minimum input voltage, VIN, is 1.8 V, and the maximum output voltage, VO U T, is 5 V.

According to (2), the value of (Tsubinterval1 + Tsubinterval2) is

approximately equal to 2.78 times the value of Tsubinterval2. If

Tsubinterval2 is chosen as 30% of the entire switching period, then the value of (Tsubinterval1 + Tsubinterval2) is 83% of the

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2758 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 10, OCTOBER 2011

Fig. 9. (a) AC analysis of conventional boost converter at light load current of 20 mA. (b) AC analysis of the proposed converter at light load of 20 mA with the activated OSS technique.

entire switching period. The remaining period of Tsubinterval3

is only 17%. In other words, a very large value of Tsubinterval2

results in a small value of the free-wheel period Tsubinterval3. For

example, according to I/O voltage setting above, the maximum value of Tsubinterval2 is 36%, while Tsubinterval3 is 0% of the

entire switching period.

A simple method to detect light load conditions is proposed (Fig. 7). The transistor ND U M M Yis biased by a constant current

source IB IA S, and the size ratio between transistors ND U M M Y

and N1 is 1/M. Switches S1 and S2 are controlled by the signal VG L, which is the gate driver of the power MOSFET N1. When

the power MOSFET N1 is turned ON, switches S2 and S1 turn ON and OFF, respectively, to sense the drain-source voltage of transistor N1.

Ignoring the parasitic series resistance of the inductor and the transistors, the detected light load current IL (light)is expressed

as (3)

IL (light)= M× IB IA S. (3)

Fig. 10. Measurement results of load transient response of a conventional boost converter.

For example, IL (light) is 100 mA if M = 50000 and IB IA S

is 2 μA. To operate the OSS technique earlier than the DCM operation at light loads, the value of IL (light) should be larger

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LUO et al.: TRANSIENT IMPROVEMENT BY WTE AND OSS TECHNIQUES IN CURRENT MODE BOOST CONVERTER 2759

Fig. 11. Measurement results of load transient response of the proposed boost converter.

Fig. 12. Measurement waveform of a conventional boost converter from light load to heavy load.

than that of the inductor current at CCM/DCM boundary condi-tion IL (b oundary). Equation (4) denotes the boundary condition

between the CCM and DCM operations, where L is the inductor value, D is the duty cycle in steady state, and Tsis the period

IL (b oundary)=

VIN× DTS

2L . (4)

Although a large value of IL (light)can be set to make the OSS

operate much earlier, it results in more power loss and lower power conversion efficiency than the CCM mode operation at the same load current condition.

V. EXPERIMENTALRESULTS

The proposed boost converter with OSS technique is fab-ricated in a 0.4 μm CMOS technology. The converter output voltage is about 4.9 V, and the input voltage is 3 V. The total feedback resistance is 564 kΩ, with the built-in reference set to 0.8 V.

Fig. 8 is the AC analysis of conventional and the proposed boost converters in the CCM and proposed converter with the

ac-Fig. 13. Measurement waveform of the proposed boost converter from light load to heavy load.

Fig. 14. Efficiency comparison of boost converter with and without OSS technique.

Fig. 15. Chip micrograph.

tivated WTE technique at heavy-load condition of 250 mA. The proposed converter with the activated WTE technique has higher cross-over frequency than that of conventional boost converter at the cost of degraded phase margin. Fig. 9 is the ac analysis of conventional and the proposed boost converters in the DCM at light-load condition of 20 mA. The proposed converter with the activated OSS technique has similar cross-over frequency as that of conventional boost converter.

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2760 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 26, NO. 10, OCTOBER 2011

TABLE I DESIGNSPECIFICATIONS

TABLE II PERFORMANCESUMMARY

Fig. 10 shows the measured waveform of a conventional boost converter when the load current changes from 400 mA to 0 mA load. The drop voltage and overshoot voltage during load tran-sient are about 235 mV and 110 mV, respectively. The settling time is 210 μs at no-load condition. A longer settling time is observed because there is no load, and the feedback resistance at the output is high.

The same transient condition and external component are used for comparison between a conventional design and the proposed method, as shown in Fig. 11. The output drop voltage is reduced from 235 to 90 mV and the overshoot voltage is effectively reduced from 110 to 54 mV. Furthermore, the settling time is decreased to 120 μs, which is 43% better than that of a conventional boost converter.

Fig. 12 shows the transient waveform of a conventional boost converter with a gradual increase in load current. According to the load current, the converter is in the DCM and CCM modes. Fig. 13 shows the mode transient waveform with an increase in load current. At light loads, free-wheel switching is turned ON in the beginning. The CCM operation replaces the operation of the free-wheel switching when the inductor current increases high enough to push the converter into heavy load operation. Higher switching frequency waveform can be observed when the WTE technique is activated during the TLH period and shown at the

right-hand-side of Fig. 13. This mechanism is turned OFF when the converter reaches its steady state.

Fig. 14 is the efficiency comparison between the conven-tional and the proposed converters. The proposed converter has lower efficiency performance due to the free-wheel switch-ing operation. After the free-wheel switchswitch-ing is turned OFF at heavy load, the efficiency comes back to the value silar to that of the conventional boost converter. The chip mi-crograph of the proposed converter is shown in Fig. 15. The die size is 2.16 mm2. The operation frequency is 2 MHz.

Table I shows the specifications of the proposed converter and Table II summarizes the performance of the proposed converter.

VI. CONCLUSION

In this paper, a boost converter with the WTE and the OSS techniques is presented. This work provides a simple method of reducing overshoot during load transient from heavy load to very light load. The design and simulation are based on 0.4 μm CMOS technology and show a reduced overshoot volt-age that ensures the performance of DSC systems. Simulation and measurement results demonstrate how overshoot and set-tling time are effectively reduced using a combination of the WTE and the OSS techniques at light loads. The improved drop reduction and overshoot reduction are about 62% and 51%, re-spectively.

ACKNOWLEDGMENT

The authors thank Advanced Analog Technology, Inc., for its assistance.

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Yean-Kuo Luowas born in Tainan, Taiwan. He re-ceived the B.S. and M.S. degrees from the Insti-tute of Microelectronics, Nation Cheng Kung Uni-versity, Tainan, Taiwan, in 2001 and 2003, respec-tively, where he is currently working toward the Ph.D. degree.

He is a Faculty Member at the Mixed-Signal and Power Management Integrated Circuit Laboratory, Institute of Electrical Control Engineering, National Chiao Tung University, Hsinchu, Taiwan.

Chao-Chang Chiouwas born in Taoyuan, Taiwan. He received the B.S. degree from Fu Jen Catholic University, Taipei, Taiwan, in 2008, and the M.S. degree from National Central University, Taoyuan, Taiwan, in 2010, at the Department of Electrical En-gineering, where he is currently working toward the Ph.D. degree from the Institute of Electrical Con-trol Engineering, National Chiao Tung University, Hsinchu, Taiwan.

He is a Faculty Member at the Mixed-Signal and Power Management Integrated Circuit Laboratory, Institute of Electrical Control Engineering, National Chiao Tung University. His current research interests include the power management integrated circuit design and analog integrated circuit designs.

Chun-Hsien Wu was born in New Taipei City, Taiwan. He received the B.S. degree from the Na-tional Taiwan University of Science and Technology, Taipei, Taiwan, in 2007, and the M.S. degree from the National Tsing Hua University, Hsinchu, Taiwan, in 2010, both in electrical engineering, where he is currently working toward the Ph.D. degree in electri-cal and control engineering from the National Chiao Tung University, Hsinchu, Taiwan.

The main research area during his M. S. career was microelectronic and mechanical systems (MEMS), which include bio-MEMS and microsensor design. His current research in-cludes the power management integrated circuit design and mixed-signal inte-grated circuits design.

Ke-Horng Chen (M’04–SM’09)received the B.S., M.S., and Ph.D. degrees in electrical engineering from the National Taiwan University, Taipei, Taiwan, in 1994, 1996, and 2003, respectively.

From 1996 to 1998, he was a part-time Integrated Circuit Designer at Philips, Taipei. From 1998 to 2000, he was an Application Engineer at Avanti, Ltd., Taiwan, and from 2000 to 2003, he was a Project Manager at ACARD, Ltd., where he was engaged in designing power management ICs. He is currently an Associate Professor in the Department of Electrical Engineering, National Chiao Tung University, Hsinchu, Taiwan, where he orga-nized a Mixed-Signal and Power Management IC Laboratory. He is the author or coauthor of more than 80 papers published in journals and conferences, and also holds several patents. His current research interests include power management ICs, mixed-signal circuit designs, display algorithm and driver designs of liquid crystal display TV, red, green, and blue color sequential back-light designs for optically compensated bend panels, and low-voltage circuit designs.

Wei-Chou Hsu (M’87) was born in Taichung, Taiwan, on May 28, 1957. He received the B.S., M.S., and Ph.D. degrees from the National Cheng Kung University (NCKU), Tainan, Taiwan, all in electri-cal engineering, in 1979, 1981, and 1984, respec-tively. In 1979, he passed the National Higher Civil Service Examination and received the Technical Ex-pert License of the Republic of China in electrical engineering.

In 1983, he was with Four Dimensions Company, California, as an Engineer. From 1982 to 1985, he was an Instructor with the Department of Electrical Engineering, NCKU, and since 1985, he has been an Associate Professor. From 1991 to 1992, he was a Postdoctoral Researcher with the Department of Electrical Engineering, Uni-versity of Florida, Gainesville. Since 1993, he has been a Professor with the Department of Electrical Engineering, NCKU. During 2000–2005, he was an Associate Chair with the Department of Electrical Engineering, NCKU, and the Chair of the Institute of Microelectronics, NCKU. Between 2005 –and 2008, he was the Chair of the Department of Electrical Engineering, NCKU. Since 2008, he has been the Chair of the Advanced Optoelectronic Technol-ogy Center, NCKU. His research interests include metal–organic chemicalva-pordeposition and molecular-beam-epitaxy-grown pseudomorphic heterostruc-ture field-effect transistors, d-doped FETs, high-power FETs, heterojunction bipolar transistors, organic light-emitting diodes, and organic photovoltaic devices.

數據

Fig. 1. Proposed current mode boost converter.
Fig. 4. (a) Transistor level of the WTE technique. (b) Oscillator and frequency selector for improving transient response in the WTE controller.
Fig. 5. Operation of the proposed OSS technique of a boost converter.
Fig. 8. (a) AC analysis of conventional boost converter at heavy load of 250 mA. (b) AC analysis of the proposed converter at heavy load with the activated WTE technique.
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