Abstract—This paper proposes a modulated hysteretic current
control (MHCC) technique to improve the transient response of a DC-DC boost converter, which suffers from low bandwidth due to the existence of the right-half-plane (RHP) zero. The MHCC technique can automatically adjust the on-time value to rapidly increase the inductor current, as well as to shorten the transient response time. In addition, based on the characteristic of the RHP zero, the compensation poles and zero are deliberately adjusted to achieve fast transient response in case of load transient condition and adequate phase margin in steady state. Experimental results show the improvement of transient recovery time over 7.2 times in the load transient response compared with the conventional boost converter design when the load current changes from light to heavy or vice versa. The power consumption overhead is merely 1%.
Index Terms—Boost converter, hysteretic current control
(HCC), load transient response, right-half-plane (RHP) zero.
I. INTRODUCTION
L
IGHT emission diode (LED) backlights have become increasingly popular in the recent green power main-stream. LED backlights have better color gamut and consumes less power than the cold cathode fluorescent lamp (CCFL) backlights [1], [2]. In addition, start-up time is shorter than that of the latter. As a result, recent liquid crystal display (LCD) backlight systems have gradually shifted to LED backlight to enhance display quality and power consumption.One of the main characteristics that may affect image quality is backlight uniformity. Therefore, a boost converter is widely utilized for LED backlight display systems to step up low-input voltage to high-output voltage for driving a number of LEDs in series. The brightness of the LED can be easily adjusted by the digital dimming control method and by the constant current sink circuit [3]–[5], thereby achieving precise backlighting control. Additionally, the voltage across the current sink circuit must be minimized to reduce power consumption. As a result, using a minimum output voltage of the boost converter to activate the LED strings can enhance the efficiency of the LED driver. To provide good driving capability for controlling the brightness of LED, the boost converter is required to handle massive in-stant load variation to maintain the image quality. Therefore, the transient response is also an essential design issue because Manuscript received July 15, 2010; revised October 04, 2010; accepted November 26, 2010. Date of publication February 14, 2011; date of current version July 27, 2011. This paper was recommended by Associate Editor E. Alarcon.
The authors are with the Institute of Electrical Control Engineering, National Chiao Tung University, Hsinchu 300, Taiwan (e-mail: khchen@cn.nctu.edu.tw). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TCSI.2011.2106231
Fig. 1. Proposed HCC technique uses an error amplifier to enhance the regu-lation accuracy.
the large output voltage drop on the boost converter may result in the insufficient driving voltage for activating the LEDs.
Due to its low power consumption, the hysteretic current con-trol (HCC) technique is selected as the modulation method for the LED backlight [6]. However, the voltage regulation perfor-mance of the HCC technique is inferior to that of the pulsewidth modulation (PWM) method [7] because the inductor current level varies under the different load conditions. Thus, in the pro-posed HCC technique, an error amplifier is utilized to form a voltage loop to improve regulation accuracy, as illustrated in Fig. 1. In addition, a pre-defined current hysteresis window, , can limit the inductor current within it to guarantee a re-stricted output voltage ripple, as expressed in (1) [8]. is the voltage across the equivalent series resistance (ESR) and is the voltage across the capacitor Co. The regulation performance and power consumption can therefore be ensured by the pro-posed HCC technique
(1) Furthermore, the proposed HCC technique has fast transient response because the trailing and leading edges can rapidly react to the output load variation. The on-time of the pulsewidth signal adaptively increases or decreases when the load current sud-denly increases or decreases, respectively. Similarly, the off-time can be modulated to improve the transient response off-time due to the HCC window controlled by the error amplifier.
Unfortunately, unlike the design of buck converters, the tran-sient response of the boost converter is limited by the existence of the RHP zero in continuous conduction mode (CCM) because the RHP zero remains in both of the PWM control (in either voltage- or current-mode) and the HCC technique [9], [10]. In 1549-8328/$26.00 © 2011 IEEE
Fig. 2. Inductor current waveform is limited with the hysteresis window de-fined by the HCC technique.
a conventional boost converter design, the discontinuous con-duction mode (DCM) is widely used to obtain simple system compensation due to the existence of the RHP zero at high fre-quencies. However, the load transient response in DCM hardly meets the requirement of the LED backlight. As a result, elimi-nating the effect of the RHP zero is a proper solution in a boost converter to speed up the load transient response. The MHCC technique is proposed to improve the transient response in the DC-DC boost converters in order to automatically adjust the on-time value and rapidly increase the inductor current, thereby shortening the transient response time. Moreover, it can delib-erately adjust the compensation poles and zero to achieve fast transient response in case of load transient condition and yield adequate phase margin (PM) in steady state.
The paper is organized according to the following sections. Section II shows the small-signal analysis of the HCC tech-nique indicating the system compensation for ensuring stability. Section III describes the operation of the proposed MHCC tech-nique to achieve the fast transient response. A comparison is also shown to demonstrate the performance. Circuit implemen-tations are illustrated in Section IV. Experimental results are dis-cussed in Section V to prove the performance and precision of the proposed MHCC technique. Finally, a conclusion is made in Section VI.
II. SMALL-SIGNALANALYSIS ANDSYSTEMCOMPENSATION WITHHCC TECHNIQUE
A. Small-Signal Modeling of the HCC Technique
As depicted in Fig. 2, the HCC technique senses the inductor current and limits it within a hysteresis window, defining the upper and lower current bands. The inductor current rises to reach the upper band of the hysteresis window when the N-MOSFET turns on during period [11]. In contrast, the inductor current falls to reach the lower band of the hys-teresis window when the N-MOSFET turns off during the period. This simple HCC technique exhibits fast dynamic characteristics.
The switching period , as expressed in (2), is equal to the sum of the on-time and the off-time in the CCM oper-ation. The value of can be written as (3) according to the waveform in Sub-interval 2
(2) (3)
The peak inductor current, , can be expressed as (4) by and the average inductor current .
(4) Given the small-signal analysis, the value of each variable can be written as the summation of the DC term, whose perturbation as shown in (5). The duty cycle, , and its complementary value,
, also are defined in (6)
(5) (6) Hence, (2) and (3) can be rewritten as (7) and (8), respectively (7) (8) Keeping the first-order AC terms, the small-signal equation can be derived in (9)
(9)
Similarly, and , are expressed in (10) as
(10) Therefore, the small-signal duty cycle is derived as (11)
(11)
Given that and ,
(12) can be derived from (11)
(12)
Due to the equation shown in (13), is shown in (14) (13) (14)
Fig. 3. Small-signal model of the boost converter under the hysteretic current mode control.
Because of the DC equivalent equations above, (15) and (16) can be derived as follows:
(15)
(16) The HCC technique can determine the inductor peak current by the intersection of the error signal and the current-sensing signal. The small-signal model [12]–[14] of the boost converter with a hysteretic current mode control is illustrated in Fig. 3. The control-to-output transfer function is shown in (17), where is the output load impedance, is the current sensing gain, and is the equivalent series resistance of the output ca-pacitor,
(17) where
(18) Formula (17) can be simplified as (19) because
in the proposed design, according to the design values
(19) where
(20)
Fig. 4. Dip output voltage versus the ratio of the! and ! for showing the effect of the RHP zero.
The system contains one dominant pole, , and two zeros, which include one RHP zero, , and one LHP zero, . The frequency response of the HCC technique is similar to that of the current-mode PWM technique. Thus, the proportional-integral (PI) compensation [15] is suitable to achieve system compensation in the proposed structure. The transfer function of the PI compensator is shown in (21). Com-pensation zero is used to cancel the effect of the system pole . Then, forms the dominant pole to stabilize the system, and can help decrease the high-frequency gain to eliminate the effect of
(21)
The existence of the RHP zero influences the system band-width. As illustrated in (20), the effect of RHP zero becomes worse at heavy loads. The crossover frequency, , is generally designed to be smaller than 10%–20% of the when it is under a heavy load, as shown in Fig. 4. The output voltage, , has no dip voltage in case of load current variation because is remote from the crossover frequency. In other words, the RHP zero has minimal effect on the dip output voltage. How-ever, the transient response is too slow, resulting in a large un-dershoots voltage due to the small system bandwidth. Increasing the crossover frequency can shorten the transient response and obtain a small undershoot voltage; however, a large dip voltage is also derived because of the existence of the RHP zero. As a result, there exists an optimum ratio between and
for obtaining the smallest dip output voltage, which is labeled as at Point C, as well as the shortest transient re-sponse period. Nevertheless, the PM at this optimum dip voltage is not desirable because of the existence of the and generated from the PI compensator. Therefore, the system band-width needs to be extended to ensure a stable transient response that removes the restriction of 10%–20% RHP zero. The de-signed value of this work is set at Point B, which is approxi-mately 30% of the RHP zero aiming for a better PM.
B. Closed-Loop Analysis With the PI Compensation
A closed-loop diagram of the boost converter with the HCC technique is shown in Fig. 5. The loop gain is illustrated in (22). Here, is the sensor gain, which is equal to
Fig. 5. Simplified feedback system of the HCC regulator.
compensator that represents the transfer function of the compen-sation network. The PI compensator contributes the compensa-tion poles and zero, as described in (21)
(22) Due to the decrease of the frequency of RHP zero
at heavy loads, the compensation zero, , in the PI compen-sator is designed to cancel the effect of the system pole, , to extend the system bandwidth. However, at heavy loads, the crossover frequency is limited by , resulting in an im-proper system compensation, as depicted in Fig. 6(a). According to Fig. 4, the ratio of and has an optimum value when the dip output voltage is the major concern.
In contrast, the bandwidth worsens due to the decrease in at light loads. As illustrated in Fig. 6(b), the compensation zero can be adaptively adjusted within a stable region. The maximum value of is determined by the PM because there are two poles at low frequencies once the compensation zero moves to-ward high frequencies. In addition, the minimum value of is determined by the minimum value between and because a decrease in also causes a decrease in when using the PI compensator. However, the minimum value of is no longer decided by because is smaller than
at light loads. To obtain a better PM at light loads, must be adaptively moved to effectively cancel the effect of the system pole under the different output load conditions.
An adaptive compensation zero can ensure a good PM; how-ever, improving the transient response is difficult because of the limitation in the low-frequency RHP zero. Therefore, the MHCC technique includes an adaptive compensation control (ACC) to obtain a fast transient response and a good PM in steady state.
III. PROPOSEDMHCC TECHNIQUE FOR FAST TRANSIENTRESPONSE
The proposed MHCC architecture, as shown in Fig. 7, can ensure a limited output ripple and adjust the trailing and leading edges for fast transient response. Moreover, the ACC technique can rapidly regulate the output of the error amplifier to speed up the transient response and guarantee good PM in the steady state. The difference in the voltages between the upper band and the lower band forms the hysteresis current window, which is the product of and . Here, is equal to the output voltage of the error amplifier for improving the accuracy of load regulation. The ACC technique is composed of the adaptive resistance and capacitance, which are controlled by the ACC controller. As a result, the compensation poles and
Fig. 6. Compensated loop gainT (s) (a) at heavy loads and (b) at light loads.
zero can be adaptively adjusted in the load transient and steady state.
Fig. 7. System architecture of the proposed boost converter with the proposed MHCC technique.
Fig. 8. PI compensator (a) with a small on-chip capacitor, (b) with and without the ACC technique.
A. Adaptive Compensation Control (ACC) Technique
As presented in Fig. 8(a), the PI compensator is connected at the output of the error amplifier. Given that only a small on-chip capacitor is allowed in the IC fabrication, the small feedback voltage may cause a large voltage variation at when the compensation zero is located at high fre-quencies, resulting in insufficient PM.
A pseudo large compensation capacitance is derived through the adaptive capacitance , as shown in Fig. 8(b). The large capacitance can be generated through a mirrored AC current, , connected to the ground [16] that only allows a small current to flow into the small on-chip capacitor . Thus, the on-chip compensation can be achieved by the implementation of the small capacitor . In addition, when the load tran-sient response occurs, would drop as well as , triggering fast transient operation [17] with the ACC control. Here, is the control signal, indicating the period of the fast transient re-sponse, for the auxiliary switches to return the large compen-sation capacitance to the small on-chip capacitor in the transient period. Consequently, the response at is acceler-ated for rapid reaction to the output load variations in the boost converter.
Moreover, can have a stable settling behavior due to the large transient current, which can be redirected to the ground. Similar to the off-chip compensation zero, a pseudo low-fre-quency compensation zero is used to cancel the effect of the output system pole. As mentioned above, the bandwidth is small because of the RHP zero. Furthermore, the ACC technique can deliberately control the switches and to improve the tran-sient response. This modifies the pseudo compensation poles and zero to obtain a small dip output voltage and fast transient response time, as shown in Fig. 8(b). Thus, the PI compensator can generate two poles, and , and one zero, , as shown in (23), to compensate the proposed structure.
(23) According to Fig. 4, the ratio of to can be in-creased for a short period by the ACC technique to obtain a higher bandwidth. Thus, the dip output voltage would not be significantly increased when the operation points from B toward a higher ratio of to . That is, the fast transient pe-riod will push the operation point from Point B to Point C, and then to Point D. After the fast transient period, it is pushed back
Fig. 9. The compensation poles and zeros controlled by the ACC technique (a) when load current changes from light to heavy and (b) when load current changes from heavy to light.
to Point B to ensure a better PM value. The fast transient pe-riod only occupies one fraction of the transient pepe-riod. The dip output voltage can be smaller compared to that without the ACC technique. The value of can be rapidly settled to its stable level; thus the MHCC technique can achieve a fast transient re-sponse and a small dip output voltage without being limited by the RHP zero.
The operation of the ACC technique needs careful control of the compensation poles and zeros during the fast transient pe-riod. Basically, the fast transient period contains two stages, as indicated by the signal in Fig. 8(b): Transient 1 and 2. At the Transient 1 stage, the mirrored AC current is fully redirected to the small on-chip capacitor to recover rapidly the voltage level of . Then, the Transient 2 stage needs to detect the valley point of the dip output voltage to pull back the compensa-tion poles and zeros to the posicompensa-tion that ensures that the system has a better PM value. Moreover, to achieve the exact system pole-zero cancellation, the compensation resistance is ad-justed according to the different load conditions. This ensures the adequate system PM under different output load conditions. The locations of the compensation poles and zeros controlled by the ACC technique are shown in Fig. 9. When the load changes from light to heavy [Fig. 9(a)], the ACC technique activates the fast transient procedure. In the Transient 1 period, the system bandwidth can be largely extended by moving
, and to the high frequencies. This is because the adaptive capacitance circuit returns the compensation capaci-tance to a small value, thereby triggering the fast loop response
of the proposed structure. Additionally, the period of Transient 2 can pull back the compensation poles and zero to ensure system stability. In the final status, the adaptive capacitance is set to a steady-state value as the initial status. The adaptive resistance circuit can also move the compensation zero to cancel the effect of for yielding the adequate PM. A similar operation is activated when the load changes from heavy to light as depicted [Fig. 9(b)].
B. Compensation of the Modulation Techniques
The switching frequency of the PWM mode is fixed, and the switch has to turn on and off every switching cycle. The recovery time takes a longer time, as shown in Fig. 10(a), when the output current changes from light to heavy. In contrast, the inductor current in the HCC technique is controlled within two bound-aries: , which is determined by the error amplifier, and , as depicted in Fig. 10(b) where is the current hysteresis window [18]–[20]. The recovery time can be shortened because the on-time value is not limited by the maximum on-time value in the HCC control technique. The switching period can be ex-tended; thus, the inductor current can be rapidly increased to the regulated level. The current of obtains additional time to reach the boundary, . The switching frequency of the boost converter slows when the load current changes from light to heavy. Therefore, the inductor charging period is extended to raise rapidly the inductor current level, shortening the transient response period and minimizing the transient dip voltage. This is the reason the recovery time of the HCC technique is faster than that of the current-mode PWM technique.
The proposed MHCC technique rapidly regulates the output of the error amplifier to shorten the transient response time. However, the system bandwidth is limited by the RHP zero in the boost converter [21]–[23]. The MHCC technique utilizes the characteristics of the RHP zero to initiate a large control signal due to the inferior PM to recover rapidly the voltage level at the output of the error amplifier. However, the inferior PM cannot ensure a stable operation in the steady state. The MHCC tech-nique can adaptively adjust the compensation poles and zeros to low frequencies to guarantee system stability. During the tran-sient response, if is able to rise and fall quickly, the recovery time is shortened, as shown in Fig. 10(c).
IV. CIRCUITIMPLEMENTATIONS
The proposed MHCC technique contains two main blocks: the HCC and the ACC circuits. The current sensor and the fixed hysteretic current window circuit constitute the HCC circuit. The adaptive compensation resistance, the capacitance circuits, and the ACC controller constitute the ACC circuit.
A. Current Sensor
The HCC technique needs to sense the full-range inductor current. Thus, an accurate current sensor is required. As de-picted in Fig. 11, a small value-sensing resistor is connected in series with the inductor to sense the full-range inductor cur-rent. A slight reduction in the power conversion efficiency may ensue in this simple procedure. The transistors and are
biased by the same bias current . Thus, .
de-Fig. 10. Recovery time during light load to heavy load. (a) Waveforms controlled by the PWM technique. (b) Waveforms controlled only by the HCC technique. (c) Waveforms controlled by the proposed MHCC technique.
Fig. 11. Schematic of current sensor.
rived and simplified as (25) if under a good layout matching result
(24) (25) The sensing signal can be expressed in (26) and used to represent the full-range inductor current. Hence, the value of can be scaled by the ratio of to . According to the operation of the HCC technique, the value of is limited
Fig. 12. Fixed hysteretic current window circuit.
within the hysteresis window, which is controlled by the fixed hysteretic window circuit
(26)
B. Fixed Hysteretic Current Window Circuit
The fixed hysteretic current window circuit is designed to control accurately the output ripple for ensuring the regulation performance, as shown in Fig. 12. The lower band of the fixed hysteretic current window is controlled by the output of the error
Fig. 13. (a) Schematic of the ACC controller. (b) Adaptive capacitance cir-cuit. (c) Time diagram of the switch control signals. (d) Waveforms of the peak detector.
amplifier, . Thus, a unity-gain buffer used to filter out the switching noise can generate the lower band . The hysteretic window is easily generated by adding an IR-drop to . The value of the IR-drop is derived by a constant current flowing through a hysteresis resistor . As a result, the upper band
can be expressed by (27). The cascaded current mirror and can suppress the channel length modulation effect to obtain higher accuracy
(27)
C. Adaptive Capacitance and Resistance Circuits and the ACC Controller
The fast transient mechanism is triggered by the threshold detector in Fig. 13(a). Once the feedback voltage is higher or lower than or , the ACC controller begins to control the fast transient procedure. The ACC technique can speed up the transient response by two transient procedures: Transient 1
and Transient 2 stages. The Transient 1 stage is simply decided by a one-shot circuit to increase rapidly the voltage of . As depicted in Fig. 13(a), the one-shot period can be determined by (28)
–
(28) The one-shot period is inversely proportional to the input voltage because the one-shot value should be shortened under a large input voltage. Moreover, bigger overshoot and undershoot output voltages need longer one-shot values. The adaptive re-sistance is shown in Fig. 13(b), which is equal to the value of
two resistances in parallel (i.e., = ).
The maximum equivalent value of is designed as
at light loads because the transistor operating in the cut-off region causes to be sufficiently large to be ignored. The adaptive capacitance contains the voltage follower designed by the low-voltage operational amplifier to ensure the accurate current mirror constituted by the transistors – . The input common-mode range can be set from 0.4 to 1.9 V. The time diagram of the switch control signals is shown in Fig. 13(c). The control signal , shown in Fig. 8(b), represents the fast tran-sient period including Trantran-sients 1 and 2. During Trantran-sient 1, the switches in the switch array of Fig. 13(b) are all enabled by the detail control signals of and from the ACC controller. During Transient 2, the is still on to derive the correct oper-ation of the ACC technique.
Consequently, the current mirror array in Fig. 13(b) can de-termine the value of the pseudo capacitance . The ratio of the current mirror array is 1:M:N:(100-M-N). In this paper,
and . Thus, changes from
to the value, as shown in (29). Here, is the enlarged factor
(29) In other words, 95% of the mirrored current is directed to the small capacitor to rapidly increase the voltage level of . As a result, the system bandwidth can be extended, and the drop voltage can be reduced. However, the PM is not sufficient to ensure a stable operation. Thus, the one-shot timing control in (28) depends on and to avoid oscillation.
After the Transient 1 stage, the Transient 2 stage is deter-mined by the valley of the output voltage. Thus, a peak de-tector in Fig. 13(a) is required to decide the period of the Tran-sient 2 stage. As illustrated in Fig. 13(d), the feedback signal can be filtered by two low-pass filters, and , to generate two output signals, and . In the Transient 2 stage, the switches in Fig. 13(b) are conser-vatively set to increase to the value, as shown in (30)
TABLE II
LOCATIONLIST OFALLPOLES ANDZEROS INMHCC STRUCTUREWITHTRANSIENTFROMHEAVYLOAD TOLIGHTLOAD
Fig. 14. Bode plot of the proposed boost converter with the HCC technique only.
During the Transient 2 stage, the compensation poles and zeros are pulled toward the origin to obtain a higher PM value. Once is higher than , the Transient 2 stage ends. After the detection of the peak value, the compensation poles and zeros are set to the positions that can guarantee an adequate PM
value. Thus, is again equal to .
The Bode plot of the proposed boost converter with the HCC technique only is shown in Fig. 14. The loop gain varies with the different load conditions in the range of 65 to 80 dB. The crossover frequency is fixed at 50 kHz. The Bode plot of the proposed MHCC technique is shown in Fig. 15. The ACC tech-nique activates the fast transient response when load current changes from light to heavy, as illustrated in Fig. 15(a). The loop crossover frequency is extended to 80 kHz during the fast transient period. The positions of all poles and zeros are listed
TABLE III
DESIGNSPECIFICATION OF THEPROPOSEDMHCC STRUCTURE
Fig. 16. Simulated results of the load transient response with the ACC tech-nique (a) when load current changes from 50 to 250 mA and (b) when load current changes from 250 to 50 mA.
value of at heavy loads, as presented in Fig. 4. Simi-larly, the Bode plot when the load current changes from heavy to light is shown in Fig. 15(b). The loop crossover frequency is also extended to 80 kHz during the fast transient period. Table II lists the relationship of the poles and zeros. At light loads, the value of is set to approximately 30% of the value of . The adaptive resistance moves both and toward the origin. Thus, is smaller than at light loads. In other words,
determines the position of .
Simulation results of the load transient response in the pro-posed boost converter with the MHCC technique are shown in Fig. 16. When the load changes from 50 to 250 mA, as shown in Fig. 15(a), the ACC technique can ensure the system stability, voltage regulation in steady state, and achieve fast transient re-sponse with periods of Transients 1 and 2. Similarly, when the load changes from 250 to 50 mA, as shown in Fig. 15(b), the fast transient mechanism also works precisely to improve the load transient response.
Fig. 17. Chip micrograph.
V. EXPERIMENTALRESULTS
The proposed boost converter with the MHCC technique was fabricated by the TSMC 0.25 m CMOS process. The off-chip inductor and output capacitor are 6.8 H and 10 F, respec-tively. The output voltage is 12 V to drive three white LEDs in series. The specification is listed in Table III. The chip micro-graph is shown in Fig. 17. The chip area is approximately 1480
m 2780 m, including the test pads.
Experimental results of conventional boost converter are shown in Fig. 18. The input voltage is 4 V. The load current changes from 70 to 270 mA. The slew-rate load current is 100 mA/ s. The undershoot and overshoot voltages are 338 and 308 mV, respectively. The recovery time is 141 s when the load current changes from light to heavy or from heavy to light. The load regulation is 0.16 mV/mA. The bandwidth is limited by the existence of the RHP zero. Under the low-bandwidth design, the transient response cannot be speeded up. Thus, the slow-response output voltage may cause the LEDs in series to have a little luminance variation when the dimming control is applied on the LED brightness control.
A comparison of the slow response of the conventional design and the experimental results of the proposed boost converter is shown in Fig. 19.
The undershoot and overshoot voltage are 300 and 234 mV, respectively. The recovery time of light to heavy load is 19.5 s, and the recovery time of heavy to light load is 18.8 s. Experi-mental results show that, compared with the conventional boost converter design, the improvement in the transient response is 7.2 times higher when the load current changes from light to heavy or vice versa. The improvement comes from the proposed HCC and the ACC techniques.
Fig. 18. Waveforms in conventional boost converter with HCC technique (a) when load current changes from 70 mA to 270 mA within 2s and (b) when load current changes from 270 to 70 mA within 2s.
The fast transient performance can be easily noted because the MHCC technique can speed up the transient response over a wide load current range, as shown in Fig. 20. The load cur-rent step decreases to approximately 100 and 50 mA, as shown in Figs. 20(a) and (b), respectively. The transient response time has only a slight improvement compared with that of the large load current step because the ACC technique has only a min-imal effect on the whole system. Fortunately, the ACC technique consumes little power because it works only when the output voltage varies significantly. The power conversion efficiency is illustrated in Fig. 21. The power conversion efficiency is hardly influenced with or without the ACC technique. The power con-sumption overhead is merely 1%. The maximum power conver-sion efficiency is approximately 90%. A summary of the com-pared performances between the conventional and the proposed MHCC techniques is listed in Table IV.
VI. CONCLUSION
This paper proposes an MHCC technique to improve tran-sient response of DC-DC boost converters. The ACC technique can settle the low-band value of the hysteresis generated by the error amplifier to further improve the transient response. Thus, the low bandwidth and large overshoot and undershoot voltages due to the RHP zero can be improved. The proposed
Fig. 19. Waveforms in the boost converter with the MHCC technique (a) when load current changes from 70 to 270 mA within 2s and (b) when load current changes from 270 to 70 mA within 2s.
Fig. 20. Waveforms in the proposed boost converter with the MHCC technique when load current changes from light to heavy within 2s.
TABLE IV
COMPARISON OF THECONVENTIONAL AND THEPROPOSEDMHCC TECHNIQUES
Fig. 21. Power conversion efficiency.
boost converter with the MHCC technique can demonstrate fast transient performance to verify its advantage in the driving capability of the LEDs in series. Good driving capability and accuracy can be guaranteed by the proposed MHCC technique according to the fast transient performance in the experimental results.
REFERENCES
[1] H.-J. Chiu and S.-J. Cheng, “LED backlight driving system for large-scale LCD panels,” IEEE Trans. Ind. Electron., vol. 54, no. 10, pp. 2751–2760, Oct. 2007.
[2] I.-H. Oh, “A single-stage power converter for a large screen LCD back-lighting,” in Proc. Appl. Power Electron. Conf. Expo., Mar. 2006, pp. 1058–1063.
[3] C.-Y. Hsieh and K.-H. Chen, “Boost DC-DC converter with fast refer-ence tracking (FRT) and charge-recycling (CR) techniques for high-ef-ficiency and low-cost LED driver,” IEEE J. Solid-State Circuits, vol. 44, no. 9, pp. 2568–2580, Sep. 2009.
[4] C.-H. Liu, Y.-C. Hsieh, T.-J. Tai, and K.-H. Chen, “SAR-controlled adaptive (SARCA) off-time control without sensing resistor for achieving high efficiency and Accuracy LED lighting system,” IEEE
Trans. Circuits Syst. I, Reg. Papers, vol. 57, no. 6, Jun. 2010.
[5] C.-Y. Hsieh, C.-Y. Yang, and K.-H. Chen, “A charge-recycling buck-store and boost-rebuck-store (BSBR) technique with dual outputs for RGB LED backlight and flashlight module,” IEEE Trans. Power Electron., vol. 24, no. 8, pp. 1914–1925, Aug. 2009.
[6] G. Jurasek, G. Levin, P. Sisson, and S. Repplinger, “High efficiency automotive power supply with hysteretic current mode controller,” in Proc. Appl. Power Electron. Conf. Expo., Mar. 1996, vol. 2, pp. 861–868.
[7] R. W. Erickson and D. Maksimovic, Fundamentals of Power
Elec-tronics, 2nd ed. Norwell, MA: Kluwer, 2001.
[8] C. Song and J. L. Nilles, “Accuracy analysis of hysteretic current-mode voltage regulator,” in Proc. Appl. Power Electron. Conf. Expo., Mar. 2005, vol. 1, pp. 276–280.
[9] W. Tang, F. C. Lee, and R. B. Ridley, “Small-signal modeling of av-erage current-mode control,” IEEE Trans. Power Electron., vol. 8, no. 2, pp. 112–119, Apr. 1993.
[10] H.-W. Huang, K.-H. Chen, and S.-Y. Kuo, “Dithering skip modulation, width and dead time controllers in highly efficient DC-DC converters for system-on-chip applications,” IEEE J. Solid-State Circuits, vol. 42, no. 11, pp. 2451–2465, Sep. 2009.
[11] H.-H. Huang, C.-L. Chen, and K.-H. Chen, “Adaptive window control (AWC) technique for hysteresis DC-DC buck converters with improved light and heavy load performance,” IEEE Trans. Power Electron., vol. 24, no. 6, pp. 1607–1617, Jun. 2009.
[12] B. Bryant and M. K. Kazimierczuk, “Small-signal duty cycle to in-ductor current transfer function for boost PWM DC-DC converter in continuous conduction mode,” in Proc. IEEE Int. Conf. Electron.
Cir-cuits Syst., May 2004, vol. 5, pp. 856–859.
[13] R. D. Middlebrook, “Modeling current-programmed buck and boost regulators,” IEEE Trans. Power Electron., vol. 4, no. 1, pp. 36–52, Jan. 1989.
[14] R. B. Ridley, “A new, continuous-time model for current-mode con-trol,” IEEE Trans. Power Electron., vol. 6, no. 2, pp. 271–280, Apr. 1991.
[15] K.-H. Chen, C.-J. Chang, and T.-H. Liu, “Bidirectional current-mode capacitor multipliers for on-chip compensation,” IEEE Trans. Power
Electron., vol. 23, no. 1, pp. 180–188, Jan. 2008.
[16] G. A. Rincón-Mora, “Active capacitor multiplier in Miller-compen-sated circuits,” IEEE J. Solid-State Circuits, vol. 35, no. 1, pp. 26–32, Jan. 2000.
[17] B. Arbetter and D. Maksimovic´, “DC-DC converter with fast transient response and high efficiency for low-voltage microprocessor loads,” in Proc. Appl. Power Electron. Conf. Expo., Feb. 1998, vol. 1, pp. 156–162.
[18] F. Su, W.-H. Ki, and C.-Y. Tsui, “Ultra fast fixed-frequency hysteretic buck converter with maximum charging current control and adaptive delay compensation for DVS applications,” IEEE J. Solid-State
Cir-cuits, vol. 43, no. 4, pp. 815–822, Apr. 2008.
[19] H. Guldemir, “Sliding mode control of DC-DC boost converter,” J.
Appl. Sci., vol. 5, no. 3, pp. 588–592, 2005.
[20] D. Ge and Z. Chen, “On-chip boost DC-DC converter in color OLED driver & controller ICs for mobile application,” in Proc. IEEE ASICON, Oct. 2005, vol. 1, pp. 459–463.
[21] T.-J. Tai and K.-H. Chen, “Switching loss calculation (SLC) and positive/negative compensated dynamic droop scaling (PNC-DDS) technique for high-efficiency multiple-input single-output (MISO) systems,” IEEE Trans. Power Electron., vol. 24, no. 5, pp. 1386–1398, May 2009.
[22] V. Gupta and G. A. Rincón-Mora, “Analysis and design of monolithic high PSR, linear regulators for SOC applications,” in Proc. IEEE SOC
Conf., Sep. 2004, pp. 311–315.
[23] S. K. Hoon, S. Chen, F. Maloberti, J. Chen, and B. Aravind, “A low noise, high power supply rejection low dropout regulator for wireless system-on-chip applications,” in Proc. IEEE Custom Integr. Circuits
management circuit designs, PFC ICs, and analog integrated circuit designs.
Chi-Lin Chen received the B.S. degree in Depart-ment of Electrical Engineering from Yuan Ze University, Taiwan, in 1995 and the M.S. degree in Department of Electrical Engineering from National Central University, Taiwan, in 1998. He is working toward the Ph.D. degree in the Department of Electrical and Control Engineering National Chiao Tung University, Hsinchu, Taiwan.
His interests include switching power circuit, mixed-signal circuit designs and analog integrated circuit designs.
Yu-Huei Lee (S’09) was born in Taipei, Taiwan. He received both B.S. and M.S. degrees from the Department of Electrical and Control Engineering, National Chiao Tung University, Hsinchu, Taiwan, in 2007 and 2009, respectively. He is currently pursuing the Ph.D. degree in the Institute of Elec-trical Control Engineering, National Chiao Tung University, Hsinchu, Taiwan.
He is a Faculty Member at the Mixed Signal and Power Management IC Laboratory, Institute of Elec-trical Control Engineering, National Chiao Tung Uni-versity, Hsinchu, Taiwan. His current research interests include the power man-agement integrated circuit design, light-emitting diode driver IC design, and analog integrated circuits.
Ming-Shen Hsu was born in Hsinchu, Taiwan. He re-ceived the B.S. degree in electrical engineering from Feng Chia University, Taiwan, in 1993, and the M.S. degree in electrical and control engineering, Chiao Tung University, Hsinchu, Taiwan in 2009.
He is now with Tontek Design Tech. Inc., Hsinchu, Taiwan, where He works on analog integrated-circuit design. His interests include analog and mixed-signal integrated circuit designs, power IC designs, and LED driver ICs.
Ke-Horng Chen (M’04-SM’09) received the B.S., M.S., and Ph.D. degrees in electrical engineering from National Taiwan University, Taipei, Taiwan, in 1994, 1996, and 2003, respectively.
From 1996 to 1998, he was a part-time IC De-signer at Philips, Taipei. From 1998 to 2000, he was an Application Engineer at Avanti, Ltd., Taiwan. From 2000 to 2003, he was a Project Manager at ACARD, Ltd., where he was engaged in designing power management ICs. He is currently an Associate Professor in the Department of Electrical Engi-neering, National Chiao Tung University, Hsinchu, Taiwan, where he organized a Mixed-Signal and Power Management IC Laboratory. He is the author or coauthor of more than 80 papers published in journals and conferences, and also holds several patents. His current research interests include power management ICs, mixed-signal circuit designs, display algorithm and driver designs of liquid crystal display (LCD) TV, red, green, and blue (RGB) color sequential backlight designs for optically compensated bend (OCB) panels, and low-voltage circuit designs.
