Circuit implementation and measurement…

2.3.2 Circuit implementation and measurement

Fig2.3-7 and Fig2.3-8 are the physical layout and measured response of the CPW balun doubler respectively. The circuit size is 100mil x 66mil.

Fig2.3-7 Physical layout of CPW balun-type doubler

-15

Conversion loss (dB) & in/out power (dBm)

fo pow er (dBm) 2fo pow er (dBm) conv loss

MA/COM 4E2037 beamlead diode

(a) Broad band measurement of conversion loss response

0

fundamental power isolation (dB)

MA/COM 4E2037 beamlead diode

(b) Broad band measurement of isolation response Fig2.3-8 Measurement of CPW balun-type doubler

The conversion loss is typically 11dB from 20 to 34GHz, 13dB from 40 to 56GHz, 16dB from 75 to 81GHz. Some resonances occur around frequencies of 38, 62, 72, and 93GHz. The measured isolation is at least 20dB in the whole band except few frequencies. The resonant phenomenon happens due to several possible reasons.

First, although the simulated result of the CPW balun alone without diodes shows good performance in broad bandwidth, the termination of the CPW balun-type doubler is diode, not 50Ω. When the diode is reversed biased, the balun sees a capacitor at the termination. This non-50Ω termination will reflect the signal back to the T junction and cause resonance. Fig2.3-9 is the simulated resonance.

(a) Circuit model of doubler resonance

(b) Simulated response of resonance Fig2.3-9 Simulation of doubler resonance

Second, a loop, resulted from the twisted line, would also form resonance. Third, the parasitic inductor of bond wire and the effect of the CPW bend implemented in the physical layout would degrade the performance and even resonate. The overall effect is simulated in Fig2.3-10 to demonstrate the resonance.

(a) Circuit model of doubler including parasitic effect

(b) Simulated conversion of CPW balun-type doubler including parasitic effect Fig2.3-10 Non-linear simulation of CPW balun doubler including bond wire parasitic

effect and twisted line

Chapter 3 Switch

Switches, which direct the signal or power flow, are extensively used in radars, instruments, antennas, and communication systems etc. When switch is discussed, several issues about it are concerned such as bandwidth, switching speed, power handling, high isolation, low voltage operations, and high operating frequency. Two kinds of semiconductor devices can be used for switches: transistors and diodes. A PIN diode switch has lower loss and handles high power levels. A MESFET switch consumes negligible power and has very fast switching speed. A MESFET switch can even provide possible power gain if the transistor is worked in active mode.

3.1 Theory of Switch design

The performance of a practical switch can be expressed by specifying its insertion loss and isolation as the basic design parameters [10].

Insertion loss: It is defined as the ratio of the power delivered to the load in the

“ON” state of the ideal switch to the actual power delivered by the practical switch. It is usually expressed in decibels and is a positive number.

Isolation: It is defined as the ratio of the power delivered to the load for an ideal switch in the ON state to the actual power delivered to the load then the switch is in the “OFF” state. This is also expressed in decibels and is a positive quantity.

When the switch is in ON state, the signal should be able to pass through the switch. Insertion loss measures the loss as the signal penetrates through it. When the switch is in OFF state, the signal should not be able to pass through the switch.

Isolation measures the ability about how the switch blocks the signal.

There are several types of switches are used most frequently in microwave frequencies as Fig3.1 depicts. They are single-pole-single-through (SPST), single- pole-double-through (SPDT), and single-pole-triple-through (SP3T) etc. The switch controls which path the signal goes through. Besides, the switches are further classified into two categories: reflective and non-reflective. In reflective switch, the signal is reflected to the source terminal at OFF state. This causes significant standing waves between components which might be a problem in some applications. In non-reflective switch, the signal would be absorbed in excess path and termination.

Thus, more devices and 50Ω terminations should be added into the circuit. Fig3.2 represents the examples of reflective and non-reflective switches in SPST configuration. The designed W-band SPST and SPDT adopts reflective switch.

In Out

(a) SPST

Out1 In

Out2 (b) SPDT

Out1

In Out2

Out3 (c) SP3T

Fig3-1 Switch configuration

VC1 VC1

In Out In Out

50Ω

VC2= VC1*

VC2= VC1*

(b) non-reflective type (a) reflective type

Fig3-2 Two classifications of switches

Moreover, there are three types of reflective SPDT switch configurations as shown in Fig3-3. They are series, shunt, and series-shunt configurations separately.

(a) series

λg/4 λg/4

λg/4 λg/4

(a) shunt

Fig3.3 Reflective switch configurations for SPDT (c) series shunt

From above configuration and circuit theory, it’s already enough to derive individual insertion loss and isolation. For simplicity, the derivation is based on SPST.

(a) Series configuration

ZO Z=R+jX

(b) Shunt configuration

2

2

Isoshunt= ILshunt

2

(c) Series-shunt configuration

Zse ILseries=Isoseries =

(3.3)

The derivation of insertion loss and isolation has been done for three types of reflective switch. It looks somewhat strange that the formulas for insertion loss and isolation are the same equations. This is natural since we represent the diode just as an impedance Z. The switch has different states, ON and OFF states. Correspondingly, the impedance of diode would appear two different levels under different bias conditions. For example, when the switch is at ON state, the impedance of the diode would be low impedance under forward biasing for series configured reflective switch.

For ideal cases, the Z approaches zero so that the ideal insertion loss (3.1) of series configured switch is 1 (in linear scale). On the other hand, when the series configured switch is at OFF state, the impedance of the diode would become high impedance under reverse biasing. Then, the isolation (the same equation 3.1) would be infinite (in linear scale) by setting Z infinity in ideal cases.

For shunt configured switch, when the switch is at ON state, the admittance of the diode should be zero in ideal cases. The similar analysis could be done by inspecting the insertion loss (3.2) by setting Y zero. For OFF state, the admittance of the diode should be infinite so that the ideal isolation would be infinite. The same validation could be done by setting Y infinite in (3.2).

The series-shunt switch is the combination of the series and the shunt configuration. The same analysis could be done by inspecting the equation (3.3). We can discover that the series or the shunt configured switch is one of the special cases by letting Zsh infinite or Zse zero.

The mechanism of switch is done by different level of impedance of diode. Thus, the measurement of the impedance of the diode under forward and reverse biasing would be the first step. The switch for vehicle-collision-avoidance-radar-system

same GaAs Schottky diode as frequency doubler used. Besides, the fast switching speed of GaAs Schottky diode switch is suitable for radar systems. Fig3-4 indicates the measured impedance of the GaAs Schottky diode under forward and reverse biasing individually.

Fig3-4 Measured impedance of Schottky diode under forward and reverse biasing

Fig3-4 depicts the measured impedances under two different biasing from 50GHz to 110GHz. The forward biasing is defined when the current of diode is 20mA and the reverse biasing is defined as the reverse applying voltage is 1V. The measurement was done by a Agilent 8510 network analyzer with waveguide measuring head. Slightly jump between V-band and W-band measurement is observed due to unavoidable measurement error. The normalized impedances of forward and reverse biasing at 77GHz is about (0.2-0.69j) Ω and (0.36+0.95j) Ω. Take these values into (3.1) and (3.2). The insertion loss and isolation for series configuration could be derived as 1.24dB and 2.09dB correspondingly. For shunt configuration, the insertion loss and isolation are 2.027dB and 2.76dB respectively. It is apparent that the insertion loss for each configuration is still acceptable but the isolation is too terrible.

The reason why the switch performs awfully at OFF state is that the impedance at reverse biasing is not high enough. Thus, the diode couldn’t be used directly for series or shunt configuration. Matching network must be inserted to make the impedance as high (or low) as possible at OFF (or ON) state. The matching network can be realized in the following procedures.

First, apply matching network so that the impedances of forward and reverse biasing are, complex conjugate pairs, symmetrical to the real axis in the Smith chart as Fig3-5 shows. Second, add a quarter-wavelength transformer to the matching network to transform two impedances with 180° difference like Fig3-6. Finally, the matching is completed by rotating these two impedances to real axis with a one-eighth wavelength transmission line. Fig3-7 depicts the final impedances of forward and reverse biasing. The high and low impedances are around 0.14Ω and 5.34Ω at ON and OFF state. The insertion loss and isolation are correspondingly 0.588dB and 11.3dB

Fig3-5 Complex conjugate pairs in the Smith chart

Fig3-6 Transformed impedances with 180° difference

Final matching

t k

Fig3-7 Final impedance of forward and reverse biasing

The switch applied with matching network could have acceptable performance in both insertion loss and isolation. The next step would be the implementation of the needed matching network. The matching network is mainly composed of transmission line and shunt open stub. Both of them have the same impedance which is constructed from 3mil wide line width and 1.4mil wide gap spacing. The matching network is simulated in 3-D full-wave simulator HFSS, as Fig3-8 depicts.

Fig3-8 Simulated matching network in HFSS

In fact, the open stub has a small capacitance due to the fringing field at the open end which is reported in [9]. Besides, the parasitic effects of T-junction and bond wire should also be considered. Therefore, fine adjustment must be done for circuit simulation. The fitting curve is shown in Fig3-10. Then, the shunt configured SPST and SPDT could be realized utilizing the simulated matching network as Fig3-11 and Fig3-12 show.

Fig3-9 Curve fitting between ideal and practical matching network

(a) SPST simulation

(a) Simulated insertion loss and isolation of SPST Fig3-10 Shunt configured SPST simulation

(a) SPDT simulation

(a) Simulated insertion loss and isolation of SPDT Fig3-11 Shunt configured SPDT simulation

3.2 Circuit implementation and measurement

Fig3-13 indicates the circuit layouts and measurements of SPST and SPDT. The circuit sizes of them are 73mil x150mil and 150mil x150mil separately. The DC bias is applied to diodes through a low-pass filter which is formed by two long bond wires and a metal-insulator-metal capacitor. Additionally, a long bond wire, which behaves high reactance at RF, is needed for DC ground.

(a) Shunt configured SPST

(b) Shunt configured SPDT

Fig3-12 Physical layout of shunt configured SPST and SPDT

The SPST and SPDT are measured by a probe station. Fig3-14 and Fig3-15 indicate the measured results of the ON and OFF states of them. A 50Ω termination is required when SPDT is measured.

(a) Measured isolation and insertion loss of shunt configured SPST

(b) Measured return loss at ON and OFF state of shunt configured SPST Fig3-13 Measurement of shunt configured SPST

(a) Measured isolation and insertion loss of shunt configured SPDT

(b) Measured return loss at ON and OFF state of shunt configured SPDT Fig3-14 Measurement of shunt configured SPDT

The measured insertion loss and isolation of SPST are 1.44dB and 11.69dB at 77GHz separately. The best isolation, 16.41dB, occurs at 75.8GHz. Although slightly frequency shift exists between simulation and measurement, the calculation value, circuit simulation, and measurement of shunt configured SPST do not differ from each other very much as TableⅠ shows. The return losses of the ON and OFF state are 17.75dB and 1.867dB at 77GHz respectively. The worse return loss at OFF state is due to the reflective type switch configuration.

TableⅠ Insertion loss and isolation comparison

Insertion loss(dB) Isolation(dB)

Calculation value 0.588 11.3

Circuit simulation 1.152 12.73

Measurement 1.44 16.41

The measured insertion loss and isolation of SPDT are 2.455dB and 18.43dB at 77GHz individually. The best isolation, 24.89dB, occurs at 75.7GHz. The return loss should be the same for SPDT since there are always one ON path and one OFF path.

The return loss is about 18dB at 77GHz.

While it seems that the simulation in HFSS is credible in implementation of switch, one important subject should noted here for CPW simulation. The bond wire parasitic effect is not negligible about the size, location, height, and coupling effect of the bond wire. For instance, the different location of bond wire would result in different simulation result as Fig 3-16 shows. As a result, the implementation and simulation layout should closely coincide with each other.

(a) Location Variation of bond wire

(b) Various simulation result correspond to different bond wire location Fig3-15 The effect of the bond wire location