Chapter 4 Suppressing Side Lobe of Tapered Short Leaky-Wave Antenna
4.2 Antenna Design
The structure of the proposed short length leaky-wave antenna is shown in Fig. 4-2. The antenna is printed on a 1.6 mm thick FR4 substrate, the dielectric constant and the loss tangent of which are 4.1 and 0.02. It is fed by microstrip feed line with a matching stub. The antenna is contained of a tapered microstrip radiator with a shorting pin and two rectangular slots (Slot 1 and Slot 2). The total length LB1 of the antenna is chosen to be 10.0 cm (about 1.5 λB0 at 4.5 GHz, λB0 is the free space wavelength), and the length of each section of the tapered LWA is 10.0 mm. The width of each section is listed in Table 4-1. The sizes of SB1B×SB2 (Slot 1) and SB3B×SB4B (Slot 2) are respectively 6.0×12.0 mm2 and 6.0×7.5 mm2. In this chapter, we utilize Ansoft High Frequency Structure Simulator to simulate our proposed antenna.
In general, the tapered leaky-wave antenna is utilized to increase the impedance bandwidth [4-1]. However, this method of the tapered short leaky-wave antenna excites serious side lobes, and even the main lobe is replaced by side lobe at higher frequency. The measured normalized radiation patterns of the conventional tapered short LWA are exhibited in Fig. 4-3. As can be seen from Fig. 4-3, the gain of the side lobe is greatly increased as the frequency is increased, and the gain of side lobe has been equal to the main lobe at 6.4 GHz.
In order to suppress the side lobe, the method of using two slots (Slot 1 and Slot 2) and a shorting pin is proposed. The Slot 1 is mainly used to suppress the excitation of the dominant mode. The simulated return losses of the conventional tapered short LWA and the
conventional tapered short LWA with Slot 1 structure are illustrated in Fig. 4-4. From these simulated results, they can be clearly seen that the dominant mode is successfully suppressed by embedded Slot 1 on the tapered short LWA. The Slot 2 and the shorting pin are designed to affect the reflected wave and reduce the radiation of side lobe. The structures of LWA and simulated radiation pattern in YZ-plane at 6.0 GHz are shown in Fig. 4-5. As can be seen from Fig. 4-5(e), the level of the side lobe is successfully suppressed by utilizing these methods.
The simulated surface current distributions of the conventional tapered short LWA, with Slot 1, with Slot 1 and 2, and the proposed LWA at 6.0 GHz are plotted in Fig. 4-6(a), (b), (c), and (b), respectively. According to Fig. 4-6(a), the serious side lobe is generated by the strong power of current distribution at the end of the tapered short LWA. As the Slot 1 is embedded on the LWA, although the current distribution in Fig. 4-6(b) can be changed, the power at the end of the LWA with Slot 1 still radiate a serious side lobe. From Fig. 4-6(c) it can be shown that the path and the direction of current at the end of the LWA with Slot 1 and 2 are varied, and then the serious side lobe of the LWA with Slot 1 is separated into two weak side lobes, which is shown in Fig. 4-5(e). In order to reduce the power at the end of LWA with Slot 1 and 2, the shorting pin is design near the Slot 2. According to Fig. 4-6(d), because the power at the end is guided to the ground, the side lobe can be reduced. By using this method, the side lobe of the conventional tapered short LWA can be suppressed successfully. To achieve the impedance matching, a matching stub is added along the feeding line.
Figure 4-7 and 4-8 show the effects of the distance between Slot 1 and Slot 2 on radiation pattern in YZ-plane and current distribution at 6.0 GHz. As the distance is 0.75 λ0, the side lobe level is increased in Fig. 4-7, mainly because the power at the end of the LWA is larger in Fig. 4-8(a). However, the power and the direction of current at the end of the LWA can be varied in Fig. 4-8(b) as the distance is 1.0 λ0. According to these simulated results, the side lobe level is mainly varied by the distance between Slot 1 and Slot 2.
Slot 1
Fig. 4-2. Structure of the proposed short length LWA.
TABLE 4-1
DIMENSIONSOFTHE PROPOSED TAPERED SHORT LWA.
Width of Section 1 15.0 mm Width of Section 6 11.7 mm Width of Section 2 14.5 mm Width of Section 7 11.0 mm Width of Section 3 13.8 mm Width of Section 8 10.3 mm
Width of Section 4 13.1 mm Width of Section 9 9.6 mm
Width of Section 5 12.4 mm Width of Section 10 8.9 mm
0
Fig. 4-3. Measured normalized radiation patterns of the conventional tapered short LWA.
0
Fig. 4-4. Simulated return losses of the conventional tapered short LWA and the conventional tapered short LWA with Slot 1 structure.
(a)
Fig. 4-5. Structures of LWA and simulated radiation pattern in YZ-plane at 6.0 GHz: (a) conventional tapered short LWA; (b) LWA with Slot 1; (c) LWA with Slot 1 and 2; (d) proposed LWA; (e) Radiation pattern.
(a)
(b)
(c)
(d)
Fig. 4-6. Simulated surface current distributions at 6.0 GHz: (a) conventional tapered short LWA; (b) LWA with Slot 1; (c) LWA with Slot 1 and 2; (d) proposed LWA.
0
45
90
135
180 225
270 315
-10 -8 -6 -4 -2 0
YZ-Plane at 6 GHz 0.75 λ0 1.00 λ0 1.25 λ0 Z
Y
Fig. 4-7. Simulated radiation pattern in YZ-plane at 6.0 GHz.
(a)
(b)
Fig. 4-8. Simulated surface current distributions at 6.0 GHz: (a) 0.75 λ0 between Slot 1 and Slot 2; (b) 1.0 λ0 between Slot 1 and Slot 2.
4.3 Simulation and Measurement Results
The measured normalized radiation patterns of the proposed LWA at 5.9, 6.2, and 6.4 GHz are shown in Fig. 4-9. Comparing the measured radiation pattern of Fig. 4-3 with that of Fig. 4-9, it is found that the tapered short LWA and the proposed LWA are very similar in the characteristics of radiation angle and 3 dB radiation beamwidth within 6.2 GHz. However, the measured side lobe level (SLL) of the proposed LWA have been significantly improved from -0.01 to 6.13 dB at 6.4 GHz by using two slots and a shorting pin. The main lobe scanning angle of the proposed LWA is from 14° to 57° between 4.6 to 6.4 GHz. Fig. 4-10 exhibits the measured SLL, the SLL of the proposed LWA is less than -5 dB, and the variation of the SLL is independent on the frequency. Fig. 4-11 illustrates the measured maximum gains of the tapered LWA and the proposed LWA. The variation of maximum gain of the proposed LWA is slightly affected by this method, and the gains are larger than 4 dBi from 4.6 to 6.4 GHz. Fig.
4-12 plots the simulated and measured return losses. The 7-dB impedance bandwidth of measured result is about 1.6 GHz from 4.58 to 6.18 GHz. Although the bandwidth of the proposed LWA is narrower than the conventional tapered LWA (see Fig. 4-4), the proposed LWA can largely suppress side lobe level; therefore, it avoids the main lobe being replaced by the side lobe at higher frequency.
0 Fig. 4-9. Measured normalized radiation patterns of the proposed LWA.
-7
Fig. 4-10. Comparison of the measured side lobe level (SLL) of tapered LWA and proposed LWA.
0 2 4 6 8
4.6 5.2 5.8 6.4
Frequency (GHz)
Gain (dBi)
Tapered LWA Proposed LWA
Fig. 4-11. Comparison of measured maximum gains of the tapered LWA and the proposed LWA.
0
10
20
30
40
4 4.5 5 5.5 6 6.5
Frequency (GHz)
Return Loss (dB)
Simulated Measured
Fig. 4-12. Simulated and measured return losses of the proposed LWA.
4.4 Summary
In this chapter, a method is proposed to reduce the serious side lobe excited by the conventional tapered short LWA. By embedding two slots and a shorting pin, it can change the current distribution at the end of the tapered short LWA at higher frequency to reduce the radiation of the side lobe. According the measured results, the SLL of the proposed LWA maintains less than -5 dB from 6.0 to 6.4 GHz. The scanning range covers 43° from 14° to 57°, and the impedance bandwidth is achieved about 30%with respect to the center frequency at 5.38 GHz. Compared to the conventional tapered short LWA, the proposed LWA not only suppresses the side lobe, but also remains a wide impedance bandwidth. Furthermore, this method does not use any parasitic element or circuit, thus we can avoid enlarging antenna size.
4.5 References
[4-1] W. Hong, T. L. Chen, C. Y. Chang, J. W. Sheen, and Y. D. Lin, “Broadband tapered microstrip leaky-wave antenna,” IEEE Trans. Antennas Propag, vol. 51, no. 8, pp.
1922-1928, Aug. 2003.
[4-2] V. Nalbandian and C. S. Lee, “Tapered leaky-wave ultra wide-band microstrip antenna,” in Proc. IEEE AP-S Int. Symp., 1999, pp. 1236-1239.
C HAPTER 5
E ND -F IRE R ADIATED O N -C HIP M ONOPOLE A NTENNA FOR
WPAN A PPLICATION
An end-fired radiated on-chip monopole antenna for wireless personal area network (WPAN) application is designed. The feeding network of coplanar waveguide (CPW) structure is proposed to feed the monopole antenna. The on-chip antenna is fabricated in TSMC 0.18-μm CMOS process. The architecture of this antenna inherits rectangular monopole antenna except for its asymmetric-fed, slit, and shorting path approaches. The asymmetric-fed is to provide dual-band around 60 GHz and end-fire radiation. In addition, by embedding a slit on the monopole antenna and a shorting pin on the ground plane, this antenna can achieve wide impedance bandwidth. According to the simulation results, the impedance bandwidth is 7.7 GHz for 10-dB return loss, which covers the range from 56.4 to 64.1 GHz. The simulated maximum gain is about 3.4 dBi, and the gain of end-fire direction is about 0.75 dBi at 60 GHz.
5.1 Coplanar Waveguide (CPW) Theory
In recent years, coplanar waveguide (CPW) structure is very widely used by microwave integrated circuits (MICs), monolithic microwave integrated circuits (MMICs), or RFIC system [5-1]~[5-3]. CPW, which was first proposed by C. P. Wen [5-4] in 1969, is fabricated on a dielectric substrate. Conventional CPW structure (see Fig. 5-1) consists of a center strip conductor with two semi-infinite ground planes on a dielectric substrate for the single input, called GSG. Furthermore, the metals of conventional CPW are on the same plane, and via holes are not required to fabricate [5-5] and [5-6]. CPW structure offers several advantages to replace microstrip structure [5-5] and [5-7]:
1. Cross talk effects are very week because the signal line is existed between ground planes.
2. CPW has low dispersion to offer the wide band circuits and components.
3. As the signal is delivered on CPW, the radiation loss will be reduced.
4. Series and shunt connections are easy achieved.
The characteristic impedance of CPW is determined by the width of the signal line, w, and the width of the gap on either side, s. The approximate formula [5-8] is shown as
where
where εr is the dielectric constant of the CPW structure. If the thickness of substrate is approximately infinite, the effective dielectric constant can be defined as
2
1 r
re
ε = +ε (5-06)
According these equations, the size of CPW structure can be designed.
w s s
h
Substrate
G ro u n d G ro u n d
S ig n al l in e
Fig. 5-1. 3D structure of conventional coplanar waveguide (CPW).
5.2 Antenna Design
The structure of the proposed on-chip monopole antenna is shown in Fig. 5-2. The on-chip antenna is fabricated in TSMC 0.18-μm CMOS process. The proposed antenna is etched on a silicon substrate with relative permittivity εr = 11.9, thickness H = 0.7 mm, and 100 Ω-cm material. The CPW structure is used for feeding network of this antenna. Besides, the gap and the width of the CPW matching feed line are respectively 40 and 80 μm. The antenna consists of three parts: a rectangular monopole antenna with a slit, an asymmetric-fed, and a shorting path which connects the monopole antenna and the ground plane. In this chapter, Ansoft High Frequency Structure Simulator is utilized to simulate the proposed antenna.
The length and the width of rectangular monopole antenna are respectively 0.35 mm and 0.75 mm. According to the monopole basic theory, the resonant frequency is excited at about 78.6 GHz, and the radiated direction is at the broadside direction. However, in order to reduce the resonant frequency and the radiated direction, an asymmetric-fed, which can cause different surface current distribution on the antenna, is applied. Figure 5-3 presents the surface current distribution of the monopole antenna with central-fed and asymmetric-fed. In Fig. 5-3(a), the current distribution of the central-fed can be divided into vertical and horizontal current, and two components with 180° out of phase are excited at the horizontal direction. Therefore, radiation at the horizontal direction in the far field is very weak, and
radiated direction is at the broadside direction. Asymmetric-fed, on the other hand, also generates the vertical and the horizontal currents in Fig. 5-3(b). Due to the current level, radiated direction is mainly caused by the horizontal current to generate end-fire radiation.
Figure 5-4 shows the simulated normalized radiation patterns of the monopole antenna with asymmetric-fed at 60 GHz. It can be seen from Fig. 5-4 that the antenna generates the end-fire radiation. The simulated return losses of central- and asymmetric-fed are plotted in Fig. 5-5.
The central-fed resonates a mode at 78.6 GHz, and the asymmetric-fed excites dual mode around 60 GHz to reduce the antenna size. In order to achieve the impedance matching, the method of using a slit and a shorting path is proposed. By using these methods, the wide impedance bandwidth and end-fire radiation can be achieved successfully. Detailed dimensions are listed in Table 5-1. The chip size is about 0.62 × 1.00 mm2.
W W1
W2 W3
L1 L2
L3
S1 S2
S3
G G1
G2
Z X
Y
Fig. 5-2. Structure of the proposed on-chip monopole antenna.
(a)
(b)
Fig. 5-3. Surface current distribution of the monopole antenna with central-fed and asymmetric-fed: (a) central-fed at 78.6 GHz; (b) asymmetric-fed at 60 GHz.
0
Fig. 5-4. Simulated normalized radiation patterns of the monopole antenna with asymmetric-fed at 60 GHz: (a) XY-Plane; (b) YZ-Plane.
0
5
10
15
20
25
45 55 65 75 85
Frequency (GHz)
Return Loss (dB)
Central-fed
Asymmetric-fed
Fig. 5-5. Simulated return losses of central-fed and asymmetric-fed.
TABLE 5-1
DIMENSIONSOFTHE PROPOSED MONOPOLE ANTENNA.
W 80 μm G 40 μm
W1 0.75 mm G1 30 μm
W2 0.24 mm G2 80 μm
W3 0.15 mm L1 0.35 mm
S1 0.3 mm L2 0.23 mm
S2 0.27 mm L3 70 μm
S3 0.14 mm
5.3 Simulation Results
The layout photo and the micrographic of the on-chip antenna are shown in Fig. 5-6. Due to the dummy of 0.18-μm CMOS process, the structures of the layout photo and the proposed antenna have some different parts. However, these parts affect slightly the bandwidth and radiation pattern. Figure 5-7 plots the simulated return loss of the proposed antenna. The 10-dB impedance bandwidth of measured result is about 7.7 GHz from 56.4 to 64.1 GHz, and the minimum return loss is about 32 dB. The current distributions of the on-chip antenna at 58 and 63 GHz are shown in Fig. 5-8. The main current distribute is over the path of the slit at 58 and 63 GHz. The current distribute of the shorting path is less, so that the shorting path is used to match the impedance.
The 2D and 3D simulated normalized radiation patterns at 58 and 63 GHz are shown in Fig. 5-9 and 5-10. At the two frequencies, we can see that the direction of the pattern results is at end-fire direction. However, the simulated maximum gain of the proposed antenna is not at end-fire direction in YZ-Plane. As the result of the silicon substrate with 10 Ω-cm material, the input power is attracted to the substrate, and then the direction of the maximum gain is affect. Fig. 5-11 and Fig. 5-12 illustrate variation of the maximum gain and the gain at end-fire direction. Due to the silicon substrate of 100 Ω-cm material, the maximum gains and the gain at end-fire direction can be larger than 2.1 dBi and -0.5 dBi from 57 to 64 GHz.
(a)
(b)
Fig. 5-6. Proposed on-chip antenna: (a) layout photo; (b) micrographic.
0
5
10
15
20
25
30
35
50 60 70 80
Frequency (GHz)
Return Loss (dB)
Simulated
Fig. 5-7. Simulated return loss of the proposed antenna.
(a)
(b)
Fig. 5-8. Current distribution of the on-chip antenna: (a) 58 GHz; (b) 63 GHz.
0
Fig. 5-9. Simulated normalized radiation patterns of on-chip antenna at 58 GHz: (a) XY-Plane; (b) YZ-Plane; (c) 3D.
0
Fig. 5-10. Simulated normalized radiation patterns of on-chip antenna at 63 GHz: (a) XY-Plane; (b) YZ-Plane; (c) 3D.
1 2 3 4
57 58 59 60 61 62 63 64
Frequency (GHz)
Gain (dBi)
Proposed Antenna
Fig. 5-11. Maximum simulated gain of the proposed LWA.
-1 0 1 2
57 58 59 60 61 62 63 64
Frequency (GHz)
Gain (dBi)
Proposed Antenna
Fig. 5-12. Simulated gain of the proposed LWA at end-fire direction.
5.4 Summary
In this chapter, an on-chip CMOS monopole antenna is proposed to excite wide impedance bandwidth for wireless personal area network application and radiate to the end-fire direction. By using the asymmetric-fed, a slit on the monopole antenna, and a shorting pin on the ground plane, this antenna can achieve wide impedance bandwidth and end-fire radiation. The on-chip antenna size is only about 0.62 × 1.00 mm2. According the results, the impedance bandwidth is about 12.8 % with respect to the center frequency at 60.25 GHz, the direction of radiation pattern approximates end-fire direction, and the maximum gains are larger than 2.1 dBi from 57 to 64 GHz. This design can reduce the radiated power of back side to interfere with the frond-end circuit.
5.5 References
[5-1] S. S. Hsu, K. C. Wei, C. Y. Hsu, and H. R Chuang, “A 60-GHz millimeter-wave CPW-fed Yagi antenna fabricated by using 0.18-μm CMOS technology,” IEEE Electron Device Lett., vol. 29, no. 6, pp. 625-627, Jun. 2008.
[5-2] A. Shamim, L. Roy, N. Fong, and N. G. Tarr, “24 GHz on-chip antennas and Balun on bulk Si for air transmission,” IEEE Trans. Antennas Propag., vol. 56, no. 2, pp.303-311, Feb. 2008.
[5-3] C. Cao, Y. Ding, X. Yang, J. J. Lin, H. T. Wu, A. K. Verma, J. Lin, F. Martin, and K.
K. O, “A 24-GHz transmitter with on-chip dipole antenna in 0.13-μm CMOS,” IEEE J. Solid-State Circuits, vol. 43, no. 6, pp.1394-1402, Jun. 2008.
[5-4] C. C. Lin, S. S. Hsu, C. Y. Hsu, and H. R. Chuang, “A 60-GHz millimeter-wave CMOS RFIC-on-chip triangular monopole antenna for WPAN applications,” in Proc. IEEE AP-S Int. Symp., Jun. 2007, pp. 2522-2525.
[5-5] C. H. Doan, S. Emami, A. M. Niknejad, and R. W. Brodersen, “Design of CMOS for 60 GHz applications,” in IEEE Proc. Solid-State Circuits Conf., 2004, pp.
440-449.
[5-6] D. Bhattacharya, “Characteristic Impedance of Coplanar Waveguide,” Electron.
Lett., vol. 21, no. 13, pp. 557, Jun. 20, 1985.
[5-7] F. L. Lin and R. B. Wu, “Computations for radiation and surface-wave losses in coplanar waveguide bandpass filters,” IEEE Trans. Microw. Theory Technol. Vol.
47, no. 4, 385-389, Apr. 1999.
[5-8] R. N. Simons, Coplanar Waveguide Circuits, Components, and Systems, John Wiley
& Sons, Inc., 2001.
C HAPTER 6 F UTURE S TUDY
The proposed designs have achieved the CP monopole antenna, reduced the LWA size, solved the problem of serious side lobe, and designed on-chip antenna. Therefore, some topologies will be proposed in this section.
6.1 Radiation of Dual-Beam
In chapter 3, the slot-lobe is radiated though the slot of the proposed LWA and the measured gain of the slot-lobe is about 4 dB lower than the main beam. If the slot-lobe level is increased, the top and the bottom position will be simultaneously scanned by difference operating frequency. We propose a topology to approach this requirement. Figure 6-1 shows the schematic configuration of the topology. Some stubs are added into bigger slots of ground plane to couple the power of LWA. The power is coupled into these stubs to increase the radiation of bottom position. Furthermore, we can use switches [6-1], which are shown in Fig.
6-2, to achieve large scanning region.
6.2 Integration of Front-End
In chapter 5, an on-chip CMOS monopole antenna is proposed to excite wide impedance bandwidth for wireless personal area network application and radiate to the end-fire direction.
In order to further the application, the on-chip antenna can be integrated into front-end circuit such as receiver and transmitter front-end in [6-2] and [6-3] (see Fig. 6-3). The impedance matching and the effects of electromagnetic interference (EMI) will be considered in the integrated process. These problems are not alike the commercial produce, front-end circuit and antenna of which are individually designed, so that the effects of EMC and EMI can be solved by many methods and experiences. However, because the on-chip antenna and the front-end circuit are integrated into a chip, these problems will be a period of critical challenge.
Ground Plane Z
X Y
Stub
LWA Feed-line
Slot elements
Fig. 6-1. Schematic configuration of the topology for dual-beam radiation.
Fig. 6-2. Configure of a beam-switchable scanning LWA [6-1].
(a)
(b)
Fig. 6-3. Configure of frond-end: (a) transmitter [6-2], (b) receiver [6-3].
6.3 References
[6-1] C. J. Wang, Y. C. Shin, and C. F. Joe, “Beam-switchable scanning leaky-wave antenna,” Electron. Lett., vol. 36, no. 7, pp. 596-597, Mar. 30, 2000.
[6-2] C. Cao, Y. Ding, X. Yang, J. J. Lin, H. T. Wu, A. K. Verma, J. Lin, F. Martin, K. O.
Kenneth, “A 24-GHz transmitter with on-chip dipole antenna in 0.13-μm CMOS,”
IEEE J. Solid-State Circuits, vol. 43, no. 6, pp.1394-1402, Jun. 2008.
[6-3] C. S. Wang, J. W. Huang, S. H. Wen, S. H. Yeh, and C. K. Wang, “A CMOS RF front-end with on-chip antenna for V-band broadband wireless communications,” in IEEE European Solid State Circuits Conference, Sep. 2007, pp. 143-146.