Chapter 3 Dual‐band Branch Line Coupler with Wideband switching
3.2 Design of Wideband Switching Circuit
An RF choke circuit is usually implemented in order to block the RF signals and make the circuit operation correctly. The conventional circuit of RF choke is shown in Fig. 3‐5a. If the power is incident into RF port 1, because the quarter‐wavelength TL is a narrow‐band element operating in single band, and can be considered a very high inductive impedance, the power will not travel to DC port and be received at RF port 2. However, the inductor must be kept in the high impedance state in the dual‐bands we desire to block the RF signal and provide a path for DC bias. As a result, the circuit of wideband RF choke is sketched as a solution in Fig. 3‐5b. As shown in Fig. 3‐5b, the power is incident from the RF port1, and there is only little leaky power traveling to DC port because of the high inductive impedance Ls. Most power will therefore travel to RF port 2. Based on this concept, we design a wideband switching circuit as shown in Fig. 3‐6. By changing the state of pin diodes, we can control the power travels to either RF2 port or 50Ω termination. As the D1 is at “ON” state, D2 is at “OFF” state and the power incident from RF1, the power cannot travel to RF2. As the D1 is at
“OFF” state, D2 is at “ON” state and the power incident from RF1, most of the power will travel to RF2. The final chosen values of capacitor, inductor and resistor are 66nH, 30pF and 50Ω, respectively.
Figure 3‐5 The schematic circuit of RF choke.
3.3 Simulation and Measurement Results
In this section, we will combine the dual‐band BLC and the wideband switching circuit. The circuit schematic of the combination circuit is shown in Fig. 3‐7. As the power is incident from Port1 and wideband switching circuit (1) and circuit (2) in Fig.
3‐7 is at “Through” state and “Termination” state, only Port2 have output power. As the power is incident from Port1 and wideband switching circuit (1) and circuit (2) are at “Through” state, the power at Port2 and Port3 is equal and the phase difference between two ports is 90°. In Table 3‐2, we list four operating senses resulting from the combination of the dual‐band BLC and wideband switching circuit.
We utilized HFSS to simulate the dual‐band BLC and the simulated performances of Case 2 and Case 4 in Table 3‐2 are shown In Fig. 3‐8 and Fig. 3‐9. From Fig 3‐8 and Fig 3‐9, the power difference between two output ports of the BLC is 0.47 dB and 0.52 dB respectively in the center frequency of WiFi and WiMAX operation in Case 2 state. Moreover, the simulated angle difference between two output ports is 84 deg and ‐86 deg in center frequency of WiFi and WiMAX operation in Case 2 state, respectively as shown in Fig.3‐10.
The measured performances of Case 2 and Case 4 in Table 3‐2 are shown In Fig.
3‐11 and Fig. 3‐12. In Fig. 3‐11 and Fig. 3‐12, we find the bandwidth and the two equal‐power branches with 90° phase difference of this circuit are good for WiFi and WiMAX applications. From Fig 3‐11 and Fig 3‐12, the power difference between two output ports of the BLC is 0.69 dB and 0.77 dB respectively in the center frequency of WiFi and WiMAX operation in Case 2 state. Moreover, the measured angle difference between two output ports is 83 deg and ‐98 deg in center frequency of WiFi and WiMAX operation in Case 2 state, respectively as shown in Fig. 3‐13. Because of the value deviation of lumped elements and the soldering effect, the experimental
results are slightly different from the simulation results. Fortunately, the measured bandwidth in each operating band is about 200MHz, so it is still feasible for covering each of the required bandwidth. From Fig. 3‐8 to Fig. 13, the two output power in WiMAX operation are lower than the two output power in WiFi operation. It’s caused by the wavelength in WiMAX operation is shorter than the wavelength in WiFi operation, so the discontinuity area is increased and may affect the performance in WiMAX operation.
Figure 3‐7 Combination of the dual‐band BLC and two wideband switching circuits.
Table 3‐2 THE OPERATING SENSE OF FOUR STATES OF COMBINATION OF THE DUAL‐BAND BLC AND TWO WIDEBAND SWITHCING CIRCUITS.
Intput Switching circuit (1) Switching circuit (2) Output
Case 1 Port 1 Through Termination Port2
Figure 3‐8 The sim
(a)
(b) W
mulated resu
WiFi opera
WiMAX oper
ults of the d ation.
ration.
dual‐band BBLC in Case
2 state.
(a)
(b) W
WiFi opera
WiMAX oper ation.
ration.
Figure 3‐10 The
(a)
(b) W
e simulated
WiFi opera
WiMAX oper
angle differ ation.
ration.
rence of thee dual‐band
d BLC.
(a)
(b) W
WiFi opera
WiMAX ope ation.
ration.
Figgure 3‐12 Measured s with two
(a)
(b) W scattering p
wideband s
WiFi opera
WiMAX ope parameters switching ci
ation.
ration.
of combina rcuits in Ca
tion of the se 4 state.
dual‐band BLC
(a) WiFi operation
Chapter 4 Dual‐band Patch Antenna
In this chapter, we will propose a square patch antenna fed by two L‐shaped probe with four slots. The two L‐shaped probes can effectively reduce the coupling effect between two ports. Moreover, the square patch with four slots can make the structure of the antenna symmetric and provide dual‐band operation.
4.1 Antenna Design
The structure of slot is often adopted in the antenna design for either dual‐band excitation or bandwidth enhancement. As shown in Fig. 4‐1 [18], they propose a structure which consists of a rectangular patch loaded by two slots which are etched close to the radiating edges where the bandwidth is about 2% in two operating bands. In this structure, the slots are used to change the current of the TM30 mode on the rectangular patch and make the current distribution become more similar with the current distribution of TM10 and the current distribution in both operating band are shown in Fig. 4‐2. However, because the patch is of the rectangular shape, it is hard to integrate two orthogonal linear polarization modes in a single patch.
Moreover, two directly‐fed coaxial probes in a single patch may cause serious coupling as well.
We therefore propose a square patch antenna fed by the L‐shaped probe with four slots as shown in Fig. 4‐3. The two slots along the X‐direction are parallel to the X‐direction current distribution on the square patch, so they will only disturb the X‐direction current distribution slightly. Because it is a symmetric structure, we can therefore utilize two L‐shaped probes feeding in orthogonal directions. This technique allows one square patch to set up two L‐shaped probes along X and Y direction for producing two linear polarizations in a single patch and will not cause
serious coupling effect as well. The geometric size of AH, AG, Lp, Pw, SL, SG, Sw we implemented is 3mm, 0.5mm, 12.5mm, 47mm, 32.5mm, 2mm and 0.6mm respectively.
Figure 4‐1 The probe‐fed patch antenna with a pair of slots.
Figure 4‐3 Patch antenna fed by L‐shaped probe with four slots.
4.2 Simulation and Measurement Results
The simulation and measurement results are shown in Fig. 4‐4. The performance of low S21 means coupling effect between two ports can be observed in both of the desired frequency bands. The simulated E‐plane radiation pattern at 2.45GHz and 3.5GHz are shown in Fig. 4‐5. Because the magnitude of cross‐polarization pattern is lower ‐10 dB, it can be a good antenna candidate of linear polarization.
Fig. 4‐6 shows the measured radiation patterns. The low cross polarization represents that the additional two X‐direction slots only affect the X‐direction current slightly. The measured bandwidth and maximum gain are 2.44 % and 9.23 dBi at 2.45GHz, respectively. On the other hand, the measured bandwidth and maximum
FR4 Substrate FR4 Substrate
Air
Feeding Line
Ground Rectangular Patch
Ground Feeding Line
Rectangular Patch L-shaped probe
FR4 Substrate FR4 Substrate
(a) Side view of L-shaped probe fed patch antenna
(b) Top view of L-shaped probe fed patch antenna with four slots
Slot
Slot
Slot
Slot
X Y Pw SL
Lp AH
AG
Sw SG
gain haped prob the patch m
The simulat
Bi at 3.5GH ted results.
bes and the may not be f
‐shaped pro metric mann
(c) Th
(d) Th
Figure 4‐4
he simulatio
he measure
The scatter
on result of
ed result of t
ring parame two ortho
the dual‐ba
the dual‐ba
eters of the ogonal L‐sh
and antenna
and antenna
patch ante haped probe
a in WiMAX
a in WiMAX
enna with fo es.
X band.
X band.
our slots fedd by
-20 -15 -10 -5 0 5 10
(a) The simulated E‐plane radiation pattern at 2.45GHz.
(b) The simulated E‐plane radiation pattern at 3.5GHz.
Figure 4‐5 The simulated patterns of the patch antenna with four slots fed by two
Linear Polarization E‐plane
－Co‐ －Cross
Linear Polarization E‐plane
－Co‐ －Cross
-20 -15 -10 -5 0 5 10
(a) The measured E‐plane radiation pattern at 2.45GHz.
(b) The measured E‐plane radiation pattern at 3.5GHz.
Figure 4‐6 The measured patterns of the patch antenna with four slots fed by two orthogonal L‐shaped probes.
Linear Polarization E‐plane
－Co‐ －Cross
Linear Polarization E‐plane
－Co‐ －Cross
Chapter 5 The Dual‐band Reconfigurable Quadri‐Polarization Diversity Antenna
We further combine the three parts (dual‐band BLC, wideband switching circuit and dual‐band patch antenna) which we have designed in the previous chapters and present its performances. The system block is shown in Fig. 5‐1. As the wideband switching circuit 1 and switching circuit 2 operate at “Through” and “Termination”
state respectively and the power is incident from port 1, the antenna can generate an X‐directional linear polarization sense. As the wideband switching circuit 1 and switching circuit 2 both operate at “Through” state and the power is incident from port 1, the antenna can generate a left‐hand circular polarization sense in WiFi operation. In the same manner where the power is incidence from port4, the antenna can generate a right‐hand circular polarization sense in WiFi operation. The operating modes of the proposed antenna are shown in Table 5‐1. From Table 5‐1, we find as the wideband switching circuits change their states, the antenna structure can provide quadri‐polarization in both operating bands.
5.1 Simulation and Measurement Results
The top view of the whole simulated antenna structure in HFSS is shown in Fig.
5‐2. The simulated scattering parameters in Case 1 (port 1 excitation) and Case 3 (port 2 excitation) state are shown in Fig. 5‐3, and simulated scattering parameters of Case 2 (port 1 excitation) and Case 4 (port 4 excitation) state are shown in Fig. 5‐4.
From the simulated scattering parameters we know the whole antenna structure is
is 2.7dBi in WiFi and 2.2dBi in WiMAX operation, respectively. From the simulation results, since the cross‐polarization pattern is very small compared to the co‐polarization pattern, the antenna is convinced to have good operation in either linear polarization or circular polarization.
Figure 5‐1 The system block of the dual‐band antenna structure.
Table 5‐1 STATUTES OF THE DUAL‐BAND ANTENNA STRUCTURE.
Input Switching circuit (1) Switching circuit (2) Polarization sense
Case 1 Port 1 Through Termination X-direction
Linear Polarization
Case 2 Port 1 Through Through LHCP (WiFi operation)
RHCP (WiAX operation)
Case 3 Port 4 Termination Through Y-direction
Linear Polarization
Case 4 Port 4 Through Through RHCP (WiFi operation)
LHCP (WiAX operation)
The photograph of the antenna structure is shown in Fig. 5‐6. The measured performances of scattering parameters are shown in Fig. 5‐7 and Fig. 5‐8, and the bandwidth is 5.7% and 6% in WiFi and WiMAX operation respectively.The measured radiation patterns are shown in Fig. 5‐9 to Fig. 5‐13.In Fig. 5‐9 and Fig. 5‐10, since the cross‐polarization pattern is very small compared to the co‐polarization pattern, the dual‐band antenna is good for operating in linear polarization.In Fig. 5‐11a, the maximum gain is 2.5dBi and the 2‐dB axial‐ratio beamwidth is 86° at 2.45GHz in Case 2. In Fig. 5‐11b, we can observe the maximum gain is about 1.9dBi and the 2‐dB axial‐ratio beamwidth is 78° at 3.5GHz in Case 2. The measured performances of the dual‐band structure are listed in Table 5‐2. In Table5‐2, the quadri‐polarization beamwidth means the axial‐ratio of LHCP and RHCP are lower than 2dB and the magnitude of cross polarization of LP in Case 1 and Case 3 are lower than ‐10dB in this beamwidth in each operating band.
The measured maximum gain of Case 2 is little larger than the measured
by the dual‐band patch antenna. Therefore, we have to get the scattering parameters of the dual‐band patch antenna and dual‐band BLC individually. In Section 3‐3, we have got the simulated and measured scattering parameters of S21 and S31, which can be considered A1 and A2 respectively, the power incident to two L‐shaped probes of the dual‐band patch antenna. In Section 4‐2, we had simulated and measured scattering parameters of the dual‐band patch antenna, which can be considered a set actually radiated by the dual‐band patch antenna. We first discuss the simulated results of PANTi. In simulated results we get the parameters of A1, A2 and SANT and substitute them into Equation (5‐2). We can therefore obtain b1 and b2, and the normalized power (PANTi) actually radiated by the dual‐band patch antenna is 0.686 and 0.685 in Case 2 and Case 4 state respectively in 2.45GHz. In the same manner, we can get the normalized power (PANTi) actually radiated by the dual‐band patch antenna is 0.554 and 0.548 in Case 2 and Case 4 state respectively in 3.5GHz. From the computation results in both frequency bands, because 1) the antenna structure is symmetric, 2) the lumped elements in HFSS are lossless and 3) the values of lumped element don’t vary with frequency, the power radiated by the dual‐band patch antenna is almost equal in both of the two circular polarization states.
Different from the simulated ones, the measured A1, A2 and SANT can be gotten from the previous chapter 3 and chapter 4 and we can use the same analysis sequence to find the solutions of PANTi. The normalized power (PANTi) actually radiated by the dual‐band patch antenna is 0.508 and 0.403 in Case 2 and Case 4 state respectively in 2.45GHz, and the normalized power (PANTi) actually radiated by the dual‐band patch antenna is 0.462 and 0.448 in Case 2 and Case 4 state respectively in 3.5GHz. Obviously, the measured result PANTi is lower than the simulated result of PANTi, and it’s caused by the practical lumped elements provide extra power loss. Moreover, the measured result of PANTiin Case 4 is lower than the measured result of PANTiin Case 2. The calculated difference between Case 2 and Case 4 is 0.06 in 2.45GHz and 0.08 in 3.5GHz, so the measured maximum gain in Fig.
5‐11 is slightly larger than the measured maximum gain in Fig. 5‐12 by 0.22dBi in 2.45GHz and 0.24dBi in 3.5GHz. The main reason of the imbalance may be that A1 and A2 of Case 2 is a little different from those of Case 4 at both frequency bands, so the actual radiated power of Case 2 and Case 4 will not be the same and result in the asymmetric pattern gain at both frequency bands.
Figure 5‐33 The simul
(a)
(b) W
lated scatte
WiFi opera
WiMAX ope
ering param ation.
ration.
eters in Casse 1 and Cas
se 3 state.
Figure 5‐44 The simul
(a)
(b) W
lated scatte
WiFi opera
WiMAX ope
ering param ation.
ration.
eters in Casse 2 and Cas
se 4 state.
-20 -15 -10 -5 0 5
Linear Polarization E‐plane
－Co‐ －Cross
Linear Polarization E‐plane
－Co‐ －Cross
(a) (b)
Figure 5‐6 The photographs of the dual‐band quadri‐polarization diversity patch antenna. (a) Front side of the structure. (b) Back side of the structure.
Figgure 5‐7 Thee measured
(a)
(b) W
d scattering
WiFi opera
WiMAX ope parameters and Case 3
ation.
ration.
s of the ant .
enna struct
ture for Case 1
(a)
(b) W
WiFi opera
WiMAX ope
ation.
ration.
-20 -15 -10 -5 0 5 45
-135
90
-90 135
-45
180 0
-20 -15 -10 -5 0 5 45
-135
90
-90 135
-45
180 0
(a) The measured E‐plane radiation pattern at 2.45GHz.
(b) The measured E‐plane radiation pattern at 3.5GHz.
Figure 5‐9 The measured E‐plane radiation patterns in Case 1 state.
Linear Polarization E‐plane
－Co‐ －Cross
Linear Polarization E‐plane
－Co‐ －Cross
-20 -15 -10 -5 0 5
(a) The measured E‐plane radiation pattern at 2.45GHz.
Linear Polarization E‐plane
－Co‐ －Cross
Linear Polarization E‐plane
－Co‐ －Cross
-20 -15 -10 -5 0 5
(a) Measured circular polarization radiation pattern at 2.45GHz.
Circular Polarization
－ Gain
Circular Polarization
－ Gain
-20 -15 -10 -5 0 5
(a) Measured circular polarization radiation pattern at 2.45GHz.
(b) Measured circular polarization radiation pattern at 3.5GHz.
Circular Polarization
－ Gain
Circular Polarization
－ Gain
(a) 2.45GHz.
(b) 3.5GHz.
Figure 5‐13 The measured axial‐ratio of the proposed antenna.
Table 5‐2 THE MEASURED PERFORMANCES OF THE DUAL‐BAND ANTENNA STRUCTURE.
Frequency Polarization
Sense
Axial Ratio < 2 Beamwidth (Deg)
Quadri‐Polarization Beamwidth (Deg)
Maxinum Gain (dBi)
2.45GHz
RHCP (Case 4) 65
65
2.18
LHCP (Case 2) 90 2.4
LP (Case 1) XXX 2.3
LP (Case 3) XXX 2
3.5GHz
RHCP (Case 2) 95
70
1.8
LHCP (Case 4) 80 1.56
LP (Case 1) XXX 1.76
LP (Case 3) XXX 1.63
Chapter 6 Conclusion
We have provided another even‐odd mode method to design the dual‐band BLC and combined it with the wideband switching circuits successfully in Chapter 3. With the wideband switching circuit, we can easily control the power to either go through the circuit or be absorbed by 50‐Ohm termination from 2 GHz to 6 GHz. The dual‐band BLC provides two equal power signals with 90° phase difference in two operating bands. In Chapter 4, a dual‐band patch antenna with low cross polarization and coupling has been completely fabricated. Because of the symmetry of the antenna structure we proposed, two L‐shaped probes can be easily integrated in a single patch, and the four additional slots are etched close to each radiation side to produce another radiation frequency. Finally, we combine the above three parts to fabricate a single patch antenna structure with dual‐band reconfigurable quadri‐polarization diversity. The measured results in Chapter 5 have shown our proposed dual‐band quadric‐polarization diversity antenna meets our expectation.
WiFi and WiMAX systems are becoming more popular in wireless communication applications. And one antenna structure operating in these two bands is becoming more important. Our antenna design can be one of the best solutions for enhancing the communication quality. The idea we present in this thesis is good but there is still room for improvement to reach better performance. First, we may design a better switching circuit to increase the antenna gain and reduce the lump elements.
Moreover, the design of the dual‐band quadri‐polarization diversity antenna is convinced a rising topic and we provide one as a milestone for this demand. We believe the design of dual‐band quadri‐polarization diversity antenna will greatly bring contribution to the communication technology.
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