Chapter 2 Broadband 180° Hybrid Ring
2.4 Fabrication and Measurements
Figure 2.18 shows the photograph of designed 180° hybrid ring, of which overall
circuit size is 52×52 mm
2
. The measured amplitude responses of the circuit for in-phase operation are shown in Figure 2.19, and the measured responses of the circuit for out-of-phase operation are shown in Figure 2.20. The return loss is better than 10 dB from 1.10 to 4.12 GHz (120%) for in-phase operation and from 1.18 to 4.10 GHz (116%) for out-of-phase operation. The measured isolation is better than 25 dB in the passband for both operations.
Figure 2.18: Photograph of the broadband 180° hybrid ring.
Frequency, GHz
Figure 2.19: Measured amplitude response of the 180° hybrid ring for in-phase operation.
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
Figure 2.20: Measured amplitude response of the 180° hybrid ring for out-of-phase operation.
The amplitude and phase balance of the coupling parameters (S21/S31) are shown in Figure 2.21 and Figure 2.22. The amplitude and phase balance in the passband are less than 0.3dB and 3° for in-phase operation.
1.5 2.0 2.5 3.0 3.5
-2 -1 0 1 2
s21/s31_simulation s21/s31_measurement
Frequency, GHz
Amplitude balance, dB
Figure 2.21: Measured amplitude balance of the 180° hybrid ring for in-phase operation.
1.5 2.0 2.5 3.0 3.5 -10
-5 0 5 10
s21/s31_simulation s21/s31_measurement
Phase balance,degree
Frequency, GHz
Figure 2.22: Measured phase balance of the 180° hybrid ring for in-phase operation.
The amplitude and phase balance of the coupling parameters (S24/S34) are shown in Figure 2.23 and Figure 2.24. The amplitude and phase balance in the passband are less than 0.5dB and 3° for out-of-phase operation.
1.5 2.0 2.5 3.0 3.5
Figure 2.23: Measured amplitude balance of the 180° hybrid ring for out-of-phase operation.
1.5 2.0 2.5 3.0 3.5
Figure 2.24: Measured phase balance of the 180° hybrid ring for out-of-phase operation.
Chapter 3
180° Reflection Type Phase Shifter
3.1 Introduction
A microwave phase shifter [9] is a device that the main function of which is to change the transmission phase of a microwave signal by means of a control or switch.
Any lumped element placed in series or shunt with a transmission line results in phase shift and, hence, there is a virtually immeasurable ways to design possible phase shifter circuit configurations. Nevertheless, adding the requirements that the device has minimum insertion loss, amplitude balance and any phase shift reduces the number of practical circuits to a little. Each offers its own combination of advantages with respect to circuit size, bandwidth and the phase shift, etc.
In order to know the working principals of the phase shifter, we need to define the special terms such as transmission phase and insertion loss first. From Figure 3.1, it can be noted that phase usually is defined using the generator voltage (V0) phase as a reference, rather than that of the input voltage (V1) of the system. This choice happens because that control networks are not perfectly matched; accordingly, the input voltage does not remain constant when the state of the device is changed. For practical measurement, a phase referenced to the internal voltage of the generator is
obtained by using a directional coupler to sample the forward-going wave ahead of the two port network, as shown in Figure 3.1. These couplers give output voltages proportional in amplitude and phase to V0 and V2, individually, even in the presence of mismatches, of course, that the coupler directivity is high enough. While the loss of a phase shifter is often overcome using an amplifier stage, the less loss, the less power amplifier that is needed to overcome it. Phase shift is the change in transmission phase of a network. Hence, we just care about relative phase shift. For instance, if a two port network has two discrete states having individual transmission phases of φ1 and φ2, its phase shift is Θ = −φ φ1 2.The sign to be identified with Θ depends both
Transmission phase Argument of V V
IL Insertion loss V V
φ= =
= =
Figure 3.1: Definition of transmission phase and insertion loss.
For infinitely long sinusoids, a change in θ is the same as a shift in time, such as a time delay. For example, V2 delays by
2
π compared to V0 of its cycle as shown
in Figure 3.2.
.
Figure 3.2: Sample sinusoid waveforms showing that a time delayed signal has negative phase.
3.2 Theory
3.2.1 Reflection Type Phase Shifter Using 90° Hybrid Coupler
There are several methods to realize phase shift by electronics. The most
widespread phase shifter circuit is the one where PIN diodes (or varactors) are
connected to a 3 dB 90° hybrid ring, as shown in Figure 3.3 [10]. By providing a
positive or negative bias to PIN diode, two phase states can be obtained from the base
elements. The range of phase shifted can be designed from a few degrees to 360
degrees.
Figure 3.3: Generic reflection-type phase shifter.
The term “hybrid” has come to mean a coupler which provides a 3dB power split for the two output ports and there must be a 90° phase difference in its output signals.
The scattering matrix has the following:
In the use of phase shifting, the port 2 and port 3 should be connected with a loading circuit consisting of a PIN diode. Therefore, V2 and V3 caused by V1 are reflected on the port 2 and port3.
The relationship between the signals reflected from port 2 and port 3 and the signals input to port 2 and port 3 are:
V2'= Γ = ΓV2 V e2 jφ (3.4)
For observing the effects of these reflected voltages on port 1 and port 4, now we can apply V2' and V3' in (3.6), (3.7) to (3.3), yield: (3.8) and (3.9) shown that, the signal inputting port 1, reflected at port 2 and port 3, is equivalent to signals V2' and V3' that input from port 2 and port 3 individually.
The effect of two signals summing on port 1 is zero, it means no effect on port 1;
however, a signal that amplitude differs from V1 to ΓV1 and phase differs by
2
φ+π at output.
If the loading circuit is made by a microstrip line with terminated by PIN diode, then, as the diode provided with positive or negative bias, marked by “f” or “r”, the Γ would have a double value, noted by Γf and Γr respectively as shown in Figure 3.4. Furthermore, the φ also has double values, noted by φf and φr. Hence, when
the PIN diodes biased positively and negatively, the output signal from port 4 should be:
V4,f' = ΓV1 f ejπ φ2+j f (3.10)
V4,r' = ΓV1 r ejπ φ2+j r (3.11) The relative phase shift is:
Θ =φ φf − r (3.12)
Γf
Γf
Γr
Γr
Figure 3.4: Generic reflection-type phase shifter using PIN diode.
There are many advantages of the reflection type phase shifter:
1) It only needs the least number of diodes or varactors compared to the other types of phase shifter in the use of big phase shifting.
2) Any phase shift increment can be obtained with proper design of the reflective loading circuits.
3) The transmission match of the bit is dependent only upon the design of the hybrid coupler and is separate from the design of the reflective loading circuits.
4) The reflective loading circuits can be optimized with respect to phase shift, insertion loss, amplitude balance in two bias states, or power handling capacity without regard to transmission match. In reality, it is impossible to optimize all of these functions separately in one design; hence, some compromises must be made.
3.2.2 180° Reflection Type Phase Shifter Using 180° Hybrid Ring
In this thesis, we need to realize a 180° reflection type phase shifter with the same insertion loss in two different states. However, according to (3.10) and (3.11),
Γf and Γr are usually not the same. It is not proper to connect the PIN diode to a 3 dB 90° hybrid ring. We replace the 90° hybrid ring with the 180° hybrid ring, and two different loading circuits made by microstrip lines provided with positive and
negative bias, respectively. In the other state, the PIN diodes are provided with negative and positive bias as shown in Figure 3.5. By doing this, we can achieve our goal.
Γf
Γf
Γr
Γr
Figure 3.5: Generic 180° reflection-type phase shifter using PIN diode.
We discuss the 180° hybrid ring clearly in the previous chapter. The scattering
2 1
signals reflected from port 2 and port3 and the signals input to port 2 and port3 are:
V2' = ΓfV2 = Γf V e2 jφf (3.16)
For observing the effects of these reflected voltages on port 1 and port 4, now we can apply V2' and V3' in (3.18), (3.19) to (3.15), yield:
'
Γ so the relationship between the signals reflected from port 2 and port3 and the
signals input to port 2 and port3 are:
V2"= ΓrV2 = Γr V e2 jφr (3.22) same amplitude and the relative phase shift is:
Θ = − =φ φ" ' 180 (3.28)
3.3 Design Procedure and Simulation
3.3.1 PIN Diode Equivalent Circuit
To realize the reflection type phase shifter, we will need an equivalent circuit for the PIN diode [11]. The PIN diode structure is a PN junction separated by an intrinsic region. Therefore comes the name P-I-N; P-type Intrinsic N-type. The PIN diode is merely an extension of the PN diode. When a forward bias is applied to the PIN, the result is a short circuit as in the PN diode. And the difference transpires when reverse biased. If the PIN diode is reverse biased, the depletion layer will increase and the junction capacitance decreases. Because of the added intrinsic region, its depletion region will be larger than the PN structure. The increased depletion width results in a smaller than normal junction capacitance when reverse biased.
Consequently, the PIN is designed to work as either forward or reverse biased.
When forward biased, the series resistance ought to be as small as possible so that the diode will better represent a short circuit. When reversed biased, the junction capacitance ought to be as small as possible and close to constant over a wide range of reversed biased voltages. This allows the PIN diode to more effectively represent two distinct states; open circuit (forward bias) and short circuit (reverse bias).
The proposed equivalent circuit for the PIN diode is shown in Figure 3.6. We know that the circuit can be switched by DC to its two distinct states. In Figure 3.6,
Rf is the series resistance of the forward biased diode. When reversed biased, Cj represents the junction capacitance and Rr represents the series resistance. The parametersCp, Lp and Ls are the elements brought by the packaging of the PIN diode. The capacitance Cp which appears in shunt is a combination of the
capacitance that exists between the upper contact and the metallic ends of the semiconductor and the insulating packaging. The capacitance C2 arises from the gap in two transmission lines across which the diode will be mounted.
Besides, all metallic ends of the package will bring about inductance, too. The inductance is divided into two components Lp and Ls. The inductance Lp appears
in series with the junction capacitance. The most considerable contributions of the inductance come from the metallic contacting strap and lie on which the element is soldered. The inductance Ls accounts for the series inductance of the outlying end parts to the external contacting points. It can become very enormous if long leads are needed for bounding to the circuit.
Figure 3.6: Equivalent circuit of the packaged PIN diode.
3.3.2 Tuning Procedure of Loading Circuit
In this thesis, the PIN diodes we used are the MADP-007436 Series Case Style 1279 from M/A COM Silicon Products. From the data sheet, the value for Cj is quoted as 1 pF at -20V. A typical value for Rf is 0.5 Ohms at 10 mA. The values for
the package parameters can be determined from the case style considerations. A value of 0.1 pF and 0.6nH is quoted for Cp and Ls, respectively.Rr and Lp are not
quoted in the data sheet.
Figure 3.7 shows two different ground structures the PIN diode mounted. The PIN diode is mounted between the microstrip line and the ground area, and its bias voltage is applied on the right side. Figure 3.7(a) shows that the top view of the ground structure composed of a fan stub which is a virtual ground. A fan stub can achieve a “perfect short” over a moderate bandwidth. Figure 3.7(b) shows a modified ground structure in order to decrease the size of the circuit. Via holes connect the top metal to the bottom, which is a metallic ground plane.
microstrip
PIN diode
microstrip
PIN diode
via hole capacitor
(a)
(b)
Figure 3.7: Two ground structures of the loading circuits.
Figure 3.8 illustrates the method to measure the reflection coefficient of the PIN diode at two different states (15mA and -10V). We fabricate two 50Ω microstrip lines first, the length of which are L and 2L, respectively. The former is connected to the PIN diode in series .We measure the reflection coefficient and obtain S11a, and get S21b from the other microstrip line. Finally, we can acquire the reflection
coefficient R=S11a/S21b. By using this method; two different coefficients in two different states can be obtained.
S11a
S21b 11a/ 21b
R=S S
Figure 3.8: Method to measure the reflection coefficients of the PIN diode.
According to (3.20) and (3.26), in order to eliminate the term V and 1' V , we 1''
need to design so that the two states of the loading circuit satisfy the condition
f r 180
φ φ− = at the center frequency. We will use two different ground structures
illustrated in Figure 3.7. The ground structure of the loading circuit Ι is fan stub, and
the other is loading circuit Π.
Take loading circuit Ι for example, Figure 3.9 shows the measured results of the
reflection coefficients of the PIN diode. The Γ of the PIN diode is about 0.93~0.95.
The initial phase shift is about 156°.
freq (2.000GHz to 3.000GHz)
Figure 3.9: Measured results of the reflection coefficients of the PIN diode.
Now we add a 50Ω microstrip line to rotate the Γ to the condition which is symmetric to the real number axis of the Smith chart as shown in Figure 3.10.
freq (2.000GHz to 3.000GHz)
Figure 3.10: Rotation of the reflection coefficients of the PIN diode.
Finally, we add a 4
λ transformer to increase the phase shift to 180° at the center
frequency as shown in Figure 3.11.
freq (2.000GHz to 3.000GHz)
Figure 3.11: Simulated results of the reflection coefficients of the PIN diode.
The layout of the loading circuit Ι with physical dimensions are shown in Figure
3.12 and listed in Table 3.1.
Figure 3.12: Circuit layout of the loading circuit Ι.
W1(mil) W2(mil) L1(mil) L2(mil) L3(mil)
44 54 699 93 704
Table 3.1: Physical dimensions of the loading circuit Ι.
We can still design the loading circuit Π by the same way mentioned previously.
The layout of the loading circuit Π with physical dimensions are shown in Figure 3.13 and listed in Table 3.2.
L
1L
2W
1W
2L
3W
3C
1C
2W
4L
4L
5W
5D
Figure 3.13: Circuit layout of the loading circuit Π.
W1(mil) W2(mil) W3(mil) W4(mil) W5(mil)
44 62 250 6 100
L1(mil) L2(mil) L3(mil) L4(mil) L5(mil)
695 79 250 698 100
D(mil) C1(pF) C2(pF)
40 12 18
Table 3.2: Physical dimensions of the loading circuit Π.
3.4 Fabrication and Measurements
3.4.1 Loading Circuit of 180° Reflection Type Phase Shifter
Figure 3.14 and Figure 3.15 illustrates the photograph of loading circuit Ι and loading circuit Π, respectively. The measured return loss responses are displayed in Figure 3.16 and Figure 3.18. It has average 0.3 dB amplitude imbalance between two states in the passband. After tuning mentioned in the previous section, two states have almost 180° phase shift at the center frequency shown in Figure 3.17 and Figure 3.19.
Figure 3.14: Photograph of the loading circuit Ι.
Figure 3.15: Photograph of the loading circuit Π.
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
Figure 3.16: Measured amplitude response of the loading circuit Ι.
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
Figure 3.17: Measured phase shift of the loading circuit Ι.
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
Figure 3.18: Measured amplitude response of the loading circuit Π.
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
Figure 3.19: Measured phase shift of the loading circuit Π.
3.4.2 180° Reflection Type Phase Shifter
Figure 3.20 and Figure 3.21 shows the photograph of designed 180° reflection type phase shifters, which connect loading circuit Ι and loading circuit Π separately to the broadband 180° hybrid ring mentioned in the previous chapter. The former circuit
size is 89×63 mm
2
, and the latter which use reduced size ground structure is 72×68 mm
2
.
Figure 3.20: Photograph of the 180° reflection phase shifter Ι.
Figure 3.21: Photograph of the 180° reflection phase shifter Π.
The measured amplitude responses of the reflection type phase shifter Ι are shown in Figure 3.22. The return loss is better than 10 dB from 1.46 to 3.58 GHz (85%).The insertion loss is about 1.4dB.
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
-50 -40 -30 -20 -10 0
s11_a s21_a s11_b s21_b
Frequency, GHz
Return loss and insertion loss,dB
Figure 3.22: Measured amplitude response of the 180° reflection phase shifter Ι.
The amplitude balance and phase shift for two states are shown in Figure 3.23 and Figure 3.24. The amplitude and phase balance in the passband are less than 0.15dB and 5°.
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
Figure 3.23: Measured amplitude balance of the 180° reflection phase shifter Ι.
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
Figure 3.24: Measured phase shift of the 180° reflection phase shifter Ι.
The measured amplitude responses of the reflection type phase shifter Π are shown in Figure 3.25. The return loss is better than 10 dB from 1.47 to 3.47 GHz (80%).The insertion loss is about 1.3dB.
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
-40 -30 -20 -10 0
s11_a s21_a s11_b s21_b
Frequency, GHz
Return loss and insertion loss,dB
Figure 3.25: Measured amplitude response of the 180° reflection phase shifter Π.
The amplitude balance and phase shift for two states are shown in Figure 3.23 and Figure 3.24. The amplitude and phase balance in the passband are less than 0.15dB and 5°.
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
Figure 3.26: Measured amplitude balance of the 180° reflection phase shifter Π.
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5
Figure 3.27: Measured phase shift of the 180° reflection phase shifter Π.
Chapter 4 Conclusion
In this thesis, the 180° reflection type phase shifters using two different ground structures were proposed. Both of the phase shifters have amplitude balance within 0.15dB and phase balance within 5° in the passband. Compared to the conventional reflection type phase shifter that we connect PIN diodes to the 90° hybrid ring, such 180° reflection type phase shifter has the minimum amplitude and phase imbalance at the broadband. In the future, we will use this useful device to be the switch of the even- and odd-mode dual band bandpass filter and the switch of the even- and odd-mode loaded line phase shifter.
A filter structure has two passbands while exciting differential or common signals is shown in Figure 4.1. In microstrip coupled lines, there are different characteristic impedances and phase velocities for even- and odd-mode signals. Using the characteristics, the filters can be realized with both stepped-impedance and uniform -impedance resonators. The two passbands can be separated effectively by adjusting the coupling strength of the coupled lines. By controlling the bias of the PIN diodes of the phase shifters, we can decide the signals we want to excite. Even- and odd-mode equivalent circuits of the dual band bandpass filter are shown in Figure
4.2(a) and (b).
Figure 4.1: Generic even- and odd-mode dual band bandpass filter.
Figure 4.2: (a) Even- and (b) Odd-mode equivalent circuits of the dual band bandpass filter.
A loaded line phase shifter structure has phase shifts while exciting common or differential signals is shown in Figure 4.3. Because of the symmetry or antisymmetry of the excitation, the network can be decomposed into a set of two port networks, as shown in Figure 4.4(a) and (b). The susceptance, B, is usually implemented with an SPST PIN diode switch between two states. By exciting common or differential
signals, however, the susceptance of the load is exactly equal to the characteristic admittance of the transmission line. Without the complexity of constructing the model of PIN diode, we can easily improve the accuracy to design the loaded line phase shifter.
Figure 4.3: Generic even- and odd-mode loaded line phase shifter.
Figure 4.4: (a) Even- and (b) Odd-mode equivalent circuits of the loaded line phase shifter.
To sum up, the broadband 180° reflection type phase shifter with almost the same insertion loss can be applied in many balanced communication system, which is really a useful device in microwave engineering.
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