A Low Voltage High Current EV Drive Using Inverter Low Side Switches as Current Sensors
Ting-Yu Chang Ching-Tsai Pan Emily Fang
Department of Electronic Engineering Department of Electrical Engineering Department of Electrical Engineering Ta-Hwa Institute of Technology National Tsing-Hua University Chin-Min Institute of Technology Hsin-Chu, Taiwan, R. O. C. Hsin-Chu, Taiwan, R. O. C. Miao-Li, Taiwan, R. O. C.
eychang@et4.thit.edu.tw ctpan@ee.nthu.edu.tw emily@ms.chinmin.edu.tw
Abstract -- In this paper, first, an equivalent scalar dynamic model for the three-phase brushless dc (BLDC) motor is derived to simplify the control of the electric vehicle (EV) drive. Then, an equivalent armature current sensing technique is proposed for synthesizing the three-phase line current by using the low-side switches of the inverter as current sensors. No additional current sensors are required and no extra power loss is generated. Third, the proposed equivalent current sensing technique is dedicatedly integrated with the PWM control for convenient closed-loop torque control implementation. Finally, a prototype EV drive system is constructed by using a 36V, 100A, 8-poles BLDC motor and a CPLD. Experimental results show that the proposed EV drive can indeed achieve the desired performance. Moreover, it is seen that with the above simplification and integration, the proposed EV drive controller circuit can be easily integrated into a single chip.
Index Terms-- Brushless machines, Current measurement, Motor drives, Torque control
I. I NTRODUCTION
With the rapid economic growth, high population density as well as the subtropical climate, scooters/motorcycles are the most popular vehicles in most southeast Asia countries [1]-[3]. However, the resulting noise and air pollution indeed cause great impact to the environment. Considering both depleting fossil fuels and environment, development of zero-emission EV is considered as a viable solution in these countries. In fact, in Taiwan, R.O.C. alone, there are more than 11 million scooters/motorcycles. Also the Environment Protection Agency had offered a subsidy program to promote the use of EV. Moreover, in the transportation system of a large city, EV can be considered as an ideal popular interface traffic vehicle among the rapid transit, high speed rail system.
Basically, an EV drive system consists of a high power density BLDC motor, a battery, a transmission system and an electric drive controller. Due to the low voltage system design, rather high current rating, more than 100 amps, are required to generate sufficient high torque for accelerating and hill climbing. Considering safety, reliability and stability of the EV drive system, high torque (or current) control performance becomes very essential. However, partly because of the higher cost, and partly because of the bulky volume which is not suitable for chip integration, the traditional Hall effect current sensors are not practical choices. On the other hand, to achieve high drive
performance, all line currents information of the BLDC motor should be available for the torque loop control. In view of the above facts, this paper proposes a novel current sensing technique for obtaining the three-phase line currents by using the low-side switches of the inverter without using additional current sensors. In fact, the current sensing technique is integrated simultaneously with the PWM control for convenient closed-loop torque control.
The remaining contents of this paper may be outlined as follows. First, an equivalent scalar model for the BLDC motor is proposed in section II for simplifying the drive system control. Then, the proposed equivalent armature current sensing technique is explained and the integration of the proposed PWM control with the proposed current sensing is described in detail in section III. As an illustrative example and demonstration of the feasibility, implementation of a prototype and the corresponding experimental results are presented in section IV. Finally, some conclusions are offered in the last section.
II. A N E QUIVALENT S CALAR M ODEL FOR BLDC M OTORS In order to simplify the three-phase control and reduce the drive cost, an equivalent scalar dynamic model for a surface-mounted brushless DC motor with trapezoidal air gap flux distribution will be derived. First, the three phase dynamic model [4] of the BLDC motor is given as follows
0 0 0 0 0 0 0 0
0 0 0 0
*
⎡ ⎤ −
⎡ ⎤ ⎡ ⎤ ⎡ ⎤
⎢ ⎥
⎢ ⎥ = ⎢ ⎥ ⎢ ⎥ ⎢ + − ⎥
⎢ ⎥ ⎢ ⎥ ⎢ ⎥
⎢ ⎥
⎢ ⎥ ⎢ ⎥ ⎢ ⎣ − ⎥ ⎦
⎣ ⎦ ⎣ ⎦ ⎣ ⎦
⎡ ⎤ ⎡ ⎤
⎢ ⎥ ⎢ ⎥ +
⎢ ⎥ ⎢ ⎥
⎢ ⎥ ⎢ ⎥
⎣ ⎦ ⎣ ⎦
s
a a
b s b
c s c
a a
b b
c c
R
v i L M
v R i L M
v R i L M
i e
d i e
dt i e
(1)
( ) /
e a a b b c c r
T = e i + e i + e i ω (2) where
a , , b c
v v v : stator phase voltages
a b c , ,
i i i : line currents
a , , b c
e e e : back emfs
R s : phase winding resistance L : self-inductance
M : mutual inductance
Fig. 1. Typical waveforms of back emfs and phase currents of the BLDC motor.
ω r : mechanical angular frequency T e : electromagnetic torque
For reference, Fig. 1 also shows the typical waveforms of back emfs and phase currents. From Fig. 1 it is seen that the a-phase back emf for one period can be represented by the following equation
e
6 ( ) if 0
6 3
if ( ) 3
6 7 4
( ) if
6 3
if 4 2 3
φ
φ φ
φ
λ ω
⎧ ω − π ≤ ω ≤ π
⎪ π
⎪ ⎪ π
⎪ λ ω ≤ ω ≤ π
ω = ⎨ ⎪
λ ω
⎪ − ω − π π ≤ ω ≤ π
⎪ π
⎪ ⎪ π
⎪− λ ω ≤ ω ≤ π
⎩
m e
e e
m e
a e
m e
e e
m e e
K t t
K t
e t
K t t
K t
(3)
( ) ( 2 )
3 ω = ω − π
b e a e
e t e t (4)
( ) ( 2 )
c e a e 3
e ω = t e ω + t π (5) where ω is the electrical frequency, e K φ is a constant, and λ is the flux linkage due to the rotor permanent magnet. m
Also, to achieve ideal constant torque control, the corresponding ideal line currents are shown in the same figure. For convenient description, commutation functions
( )
Sa t , Sb t and ( ) Sc t are defined as follows ( )
0
( ) [ ( 2 ) ( 2 )
3
- ( 4 2 ) ( 2 2 )]
3
∞
=
≡ ∑ ω − − π − ω − π − π π
ω − π − π + ω − π − π
e e
n
e e
Sa t u t n u t n
u t n u t n
(6)
( ) ( 2 )
3 e Sb t = Sa t − π
ω (7)
( ) ( 4 )
3 e Sc t = Sa t − π
ω (8)
where u t ( ) denotes the unit step function and n is an integer. Thus, by synthesizing the three line currents of the BLDC motor one can obtain an equivalent armature current as follows
[ ][ ]
( ) 1 ( ) ( ) ( ) ( ) ( ) ( ) 2
eq a b c T
i t = Sa t Sb t Sc t i t i t i t (9) Similarly, one can define an equivalent back emf and equivalent terminal voltage as follows
[ ][ ]
( ) 1 ( ) ( ) ( ) ( ) ( ) ( ) 2
T
eq a b c
e t Sa t Sb t Sc t e t e t e t (10-a)
[ ][ ]
( ) 1 ( ) ( ) ( ) ( ) ( ) ( ) 2
eq a b c T
v t Sa t Sb t Sc t v t v t v t (10-b) It follows from (9), (10) and (2) that
( ) = φ λ ω
eq m e
e t K (11)
( ) / /
e a a b b c c r eq eq r m eq
T = e i + e i + e i ω = e i ω = pK φ λ i (12) where
e 2 p r
ω = ω and p is the pole number of the motor.
Finally, from (1) and definitions of (9) and (10) one can get the following equivalent scalar dynamic model
( ) eq
eq s eq eq
v R i L M di e
= + − dt + (13)
From the above results one can see that by synthesizing the three line currents, three back emfs and three phase voltages one can obtain an equivalent scalar dynamic model.
In fact, (13) has exactly the same form as that of the traditional DC motor with brushes. Hence, the three-phase control of the drive can be reduced to a scalar control of the equivalent DC brush motor.
III. T HE P ROPOSED E QUIVALENT A RMATURE C URRENT S ENSING T ECHNIQUE
As mentioned in the previous section, by synthesizing the
three line currents, three back emfs and three phase voltage
Fig. 2. The schematic diagram of the proposed equivalent armature current sensing integrated in the PWM control for the low voltage high current EC drive.
one can obtain an equivalent scalar dynamic model for the BLDC motor. In the scalar dynamic model, the equivalent armature current plays an important rule and governs the generating torque of the motor. Though, many methods [5]- [12] can be used to realize the implementation of the equivalent armature current synthesizing. However, in order to achieve a low cost and high rated current drive, the existing phase current sensing approach should be improved first. Also, the improved method should take advantages of the available components of the controller as possible. In other words, the realization of the equivalent armature current sensing should be integrated as much as possible with the PWM control of the BLDC motor drive, the proposed equivalent armature current sensing scheme is shown in Fig. 2. From Fig. 2 one can see that basically there are five parts, namely the BLDC motor, the full- bridge inverter consisting of Qu , Qv , Qw and
Qx , Qy , Qz MOSFET switches, the current sensing circuit, the coding table of timing sequence of phase current block and the gating signal generator. The proposed current sensing circuit consists of Qx ' , Qy ' , Qz ' small power transistor for sensing the corresponding drain-to-source voltage drops of Qx , Qy , Qz respectively. They are controlled by Sx , Sy and Sz for integration with the PWM control. Also, to synthesize the equivalent armature current from three line currents, three transmission gates Mx , My and Mz are adopted for selecting the correct line current. In addition, an RC filter is added to filter out the switching noise.
First, the output signals hu , hv and hw of the three Hall sensors are used to generated the u , v , w and x , y , z signals through the contents of the coding table as shown in Table I. From Table I, it is seen that through this coding one
can obtain the positive stator current signals u , v , w and the negative stator current signals x , y , z for each winding of the stator as can be observed from Fig. 1.
Second, the corresponding u , v , w and x , y , z signals together with the signal Vpwm generating from the proportional torque control loop shown in Fig. 3 are input to the gating signal generator to generate Su , Sv , Sw and
Sx , Sy , Sz signals for controlling the power switches of the inverter. Meanwhile, the input signals x , y , z can be applied to control the three transmission gates directly. The input output relations of the gating signal generator can be implemented using low cost logical devices or a simple table. Table II shows the contents of this generator. As an illustrative example, consider the second row of Table II where the Vpwm signal is at high level. The magnet position of the motor is encoded as (hu, hv, hw) = (0,0,1) . A l s o , t h e c o r r e s p o n d i n g s t a t u s o f (u, v, w, x, y, z) = (1,0,0,0,1,0) . Thus, under this state one has y 1 = , Sy 1 = , Su 1 = and the corresponding circuit can observe that under this state that the line current flows equivalent circuit of Fig. 2 becomes Fig. 4(a). From Fig. 4(a) one through phase-a winding and flows out through phase-b winding as well as Qy and back to the voltage source. Also, since Sy 1 = then Qy' is turned on to sense the voltage drop of Qy . Meanwhile since y = 1 , hence My is turned on to select this signal and pass to the RC filter to get the equivalent armature current signal. Next, when the Vpwm signal is at low level one can see from the first row of Table II that the resulting gating signals (Su,Sv,Sw,Sx,Sy,Sz)
= (0,0,0,0,1,0) . Hence, under this state one has y 1 = ,
Sy 1 = , Su 0 = and the corresponding circuit of Fig. 2
GATING
