行政院國家科學委員會專題研究計畫 期末報告
微型且寬頻的平面傳輸線至矩形波導轉接
計 畫 類 別 ： 個別型
計 畫 編 號 ： NSC 101-2221-E-011-084-
執 行 期 間 ： 101 年 08 月 01 日至 102 年 10 月 31 日 執 行 單 位 ： 國立臺灣科技大學電子工程系
計 畫 主 持 人 ： 王蒼容
計畫參與人員： 碩士班研究生-兼任助理人員：黃保仁 碩士班研究生-兼任助理人員：王思翰 碩士班研究生-兼任助理人員：李元鈞 碩士班研究生-兼任助理人員：廖婕宇
報 告 附 件 ： 出席國際會議研究心得報告及發表論文
公 開 資 訊 ： 本計畫可公開查詢
中 華 民 國 102 年 11 月 12 日
中 文 摘 要 ： 本研究計畫完成三個縮小化且寬頻的平面傳輸線至矩形波導 轉接，研究可以區分為三個部分。在第一部份，我們使用高 低阻抗線來達成縮小化且寬頻的槽線至矩形波導轉接。在第 二部分，我們使用電感補償的槽線來達成縮小化且寬頻的共 面波導至矩形波導轉接。在第三部分，我們使用電容補償的 寬邊耦合微帶線來達成縮小化且寬頻的微帶線至矩形波導轉 接。這三個部分都有實做驗證，如此可以證明這些概念的可 行性。
中文關鍵詞： 微型、寬頻、槽線、共面波導、微帶線、矩形波導
英 文 摘 要 ： In this research, three compact and broadband planar transmission line to rectangular waveguide
transitions are proposed. The first design uses the high-low impedance line to achieve a compact and broadband slotline to rectangular waveguide
transition. The second design uses the inductance- compensated slotline to achieve a compact and
broadband coplanar waveguide to rectangular waveguide transition. The third design uses the capacitance- compensated broadside coupled microstrip line to achieve a compact and broadband microstrip line to rectangular waveguide transition. These transitions are fabricated and measured where the measurement results are in good agreement with the simulation results.
英文關鍵詞： Compact, broadband, slotline, coplanar waveguide, microstrip line, rectangular waveguide
1
行政院國家科學委員會專題研究計畫成果報告
微型且寬頻的平面傳輸線至矩形波導轉接
Compact and broadband planar transmission line to rectangular waveguide transitions
計畫編號：NSC 101-2221-E-011-084 執行期限：101 年 8 月 1 日至 102 年 7 月 31 日
主持人：王蒼容 台灣科技大學電子系
計畫參與人員：王思翰、黃保仁、李元鈞、廖婕宇
一、中文摘要
本研究計畫完成三個縮小化且寬頻的 平面傳輸線至矩形波導轉接，研究可以區 分為三個部分。在第一部份，我們使用高 低阻抗線來達成縮小化且寬頻的槽線至矩 形波導轉接。在第二部分，我們使用電感 補償的槽線來達成縮小化且寬頻的共面波 導至矩形波導轉接。在第三部分，我們使 用電容補償的寬邊耦合微帶線來達成縮小 化且寬頻的微帶線至矩形波導轉接。這三 個部分都有實做驗證，如此可以證明這些 概念的可行性。
關鍵詞：微型、寬頻、槽線、共面波導、
微帶線、矩形波導
Abstract
In this research, three compact and broadband planar transmission line to rectangular waveguide transitions are proposed. The first design uses the high-low impedance line to achieve a compact and broadband slotline to rectangular waveguide transition. The second design uses the inductance-compensated slotline to achieve a compact and broadband coplanar waveguide to rectangular waveguide transition. The
third design uses the capacitance-compensated broadside coupled
microstrip line to achieve a compact and broadband microstrip line to rectangular waveguide transition. These transitions are fabricated and measured where the measurement results are in good agreement with the simulation results.
Keywords: Compact, broadband, slotline,
coplanar waveguide, microstrip line, rectangular waveguide 二、緣由與目的矩形波導具有阻絕外部雜訊、低損耗 及高功率傳輸的優點，因此，被廣泛地應 用在高品質因素的微波元件上。另一方 面，由於印刷電路製作技術成熟且製作成 本低，因此，平面傳輸線被廣泛地用來整 合積體電路、低雜訊放大器、震盪器及表 面黏着元件，以組成多功能的射頻模組。
為了整合上述兩種不同傳輸結構所形成的 電路模組，許多學者提出各種平面傳輸線 至矩形波導轉接，包含微帶線至矩型波導 轉 接 [1]–[6] 、 共 面 波 導 至 矩 型 波 導 轉 接 [7]–[17]、槽線至矩型波導轉接[18]、[19]。
在這些文獻當中，大部分的學者都致力於 達成寬頻的平面傳輸線至矩形波導轉接，
但是為了達成寬頻的特性，卻因此增加了 電路的面積。近年來，由於電路的製作成 本被要求降低，以及電路的可攜性被要求 提高，因此如何縮小電路來達成這兩個目 標，已成為一個必然的趨勢。本研究致力 於如何縮小平面傳輸線至矩形波導轉接，
並且維持其寬頻的特性，研究共區分為三 部分，包含微帶線至矩型波導轉接、共面 波導至矩型波導轉接、槽線至矩型波導轉 接。
關於微帶線至矩型波導轉接，許多學 者已經提出各種不同的技巧，來實現微帶 線至矩形波導轉接[1]–[6]。在 1985 年，S.J.
Nightingale 等學者使用領結型天線來實現 微帶線至矩形波導轉接[1]，但是這個轉接 的頻寬並不寬。為了增加轉接的頻寬，有
2
學者使用平面探針來激發矩形波導的 TE
10
模態，以達成寬頻的微帶線至矩形波導轉 接[2]，但是這個轉接的特性很容易受到矩 形波導背板的影響。除此之外，有學者使 用鰭線陣列來實現寬頻的微帶線至矩形波 導轉接[3]，但是由於需要很長的鰭線來達 成寬頻的特性，電路面積因此增加很多。另外，亦有學者使用 end-fire 天線來達成寬 頻的微帶線至矩形波導轉接[4]。但是由於 這個轉接需要一個額外的微帶線至共面帶 線的轉接，電路面積因此增加。還有學者 使用末端開路的共面帶線來實現寬頻的微 帶線至矩形波導轉接[5]，但是由於這個轉 接也需要一個額外的微帶線至槽線轉接，
因此同樣增加了電路的面積。為了縮小電 路面積，有學者使用矩形 patch 來實現縮小 化的微帶線至矩形波導轉接[6]。雖然電路 面積縮小了，但是頻寬也因此變窄了。由 以上的文獻我們可知，寬頻與縮小化兩者 是相衝突的，為了同時達成縮小化且寬頻 的特性，我們使用電容補償的寬邊耦合微 帶線，來實現縮小化且寬頻的微帶線至矩 形波導轉接。
關於共面波導至矩形波導轉接，已有 許多學者提出各種不同的方法，來實現共 面波導至矩形波導轉接[7]–[17]。首先，在 1990 年，G. E. Pohchak 等學者使用 tapered ridge 來 實 現 共 面 波 導 至 矩 形 波 導 轉 接 [7]。這個轉接的的優點是不需要磅線，但 是必須使用機械加工製作，因此其製作成 本相當高。近年來亦有學者使用半導體製 程來實現共面波導至矩形波導轉接[8]、
[9]，但是由於使用半導體製程技術的關 係，其製作成本仍然相當高。為了降低成 本，許多學者使用平面傳輸線電路，來實 現共面波導至矩形波導轉接[10]–[17]。在 這些文獻當中，有學者使用末端開路的平 面探針來實現寬頻的共面波導至矩形波導 轉接[10]。雖然這個轉接的頻寬很寬，但是 由於共面波導很靠近封裝的邊緣，因此很 容易和封裝耦合。除此之外，有學者使用 漸變鰭線來實現寬頻的共面波導至矩形波 導轉接[11]，但是因為需要很長的漸變鰭線 來達成寬頻的特性，電路面積因此增大許
多。另外，有學者使用末端開路的共面帶 線來實現寬頻的共面波導至矩形波導轉接 [12]、[13]。雖然這個轉接的尺寸比使用漸 變鰭線轉接的尺寸來得小，但是其仍然需 要一個額外的共面波導至槽線轉接，因此 電路面積仍然很大。再者，也有學者使用 end-fire 天線來實現寬頻的共面波導至矩形 波導轉接[14]，但是這個轉接仍然需要一個 額外的共面波導至共面帶線轉接，因此面 積仍然是很大。最後，有學者使用四分之 一波長的短路槽線，來實現寬頻的共面波 導至矩形波導轉接[15]，但是這個轉接也需 要一個額外的共面波導至槽線轉接，因此 面積亦是很大。為了縮小電路面積，有學 者使用槽線天線，來實現縮小化的共面波 導至矩形波導轉接[16]，但是頻寬卻因此變 窄了。除此之外，有學者亦使用彎曲槽線，
來實現縮小化的共面波導至矩形波導轉接 [17]，雖然這個轉接的頻寬有稍許增加，但 是仍然無法涵蓋整個 X-band 的頻段。在 此，為了縮小電路面積並維持寬頻的特 性，我們使用電感補償的槽線，來達成縮 小化且寬頻的共面波導至矩形波導轉接。
關於槽線至矩形波導轉接，截至目前 為止，並沒有被單獨探討。槽線至矩形波 導轉接，一般以中繼轉接的形式，出現在 微帶線至矩形波導轉接[1]、[4]、[5]，以及 共面波導至矩形波導轉接[11]–[13]、[15]、
[17]當中。此外，有學者使用四分之一波長 的共面帶線，來達成寬頻的槽線至矩形波 導轉接[18]、[19]。為了進一步縮小轉接的 尺寸並且維持寬頻的特性，我們使用高低 阻抗線，來實現縮小化且寬頻的槽線至矩 形波導轉接。
三、研究方法
本研究計畫的目的，在於達成縮小化 且寬頻的平面傳輸線至矩形波導轉接，研 究可以區分為三個部分。首先，第一部分 我們使用高低阻抗線，來達成縮小化且寬 頻的槽線至矩形波導轉接。第二部分我們 使用電感補償的槽線，來達成縮小化且寬 頻的共面波導至矩形波導轉接。第三部分 我們使用電容補償的寬邊耦合微帶線，來
3
達成縮小化且寬頻的微帶線至矩形波導轉 接，詳細的內容如下所述：
I. 槽線至矩形波導轉接：
在這個部分，我們使用高低阻抗線，
來實現縮小化且寬頻的槽線至矩形波導轉 接，如圖一所示，圖上所使用的共面帶線
L h
+Ll
長度為 190 mil。接著我們使用 Ansoft HFSS 來模擬這個架構，其返回與插入損耗 的頻率響應圖如圖二所示，由圖我們可 知，返回損耗大於 20 dB 的比例頻寬為 36.57%，幾乎涵蓋整個 X-band (8.20–12.40 GHz)，可謂相當寬頻；另外，在這個頻率 範圍內，其插入損耗皆小於 0.14 dB。為了驗證這個結果，我們實際製作一 個背對背電路，如圖三所示。接著我們使 用校準後的網路分析儀來量測這個電路，
其模擬與量測的返回及插入損耗比較圖如 圖四所示。由圖我們可以發現模擬與量測 結果滿一致的，如此可以驗證模擬結果的 正確性。
與使用四分之一波長共面帶線的槽線 至矩形波導轉接比較[19]，使用高低阻抗線 的槽線至矩形波導轉接，其 20-dB 的比例 頻寬為 36.57%，幾乎涵蓋整個 X-band，可 謂相當的寬頻，並且其可以將共面帶線的 尺寸從 230 mil 縮小至 190 mil。
II. 共面波導至矩形波導轉接：
在這個部分，我們使用電感補償槽 線，來實現縮小化且寬頻的共面波導至矩 形波導轉接，如圖五所示，圖上所使用的 共面帶線 L
C
長度為 230 mil。接著我們使用 Ansoft HFSS 來模擬這個架構，其返回與插 入損耗的頻率響應圖如圖六所示，由圖我 們可知，返回損耗大於 15 dB 的比例頻寬 為 38.06%，幾乎涵蓋整個 X-band，可謂相 當寬頻；另外，在這個頻率範圍內，其插 入損耗皆小於 0.21 dB。為了驗證這個結果，我們實際製作一 個背對背電路，如圖七所示。接著我們使 用校準後的網路分析儀來量測這個電路，
其模擬與量測的返回及插入損耗比較圖如
圖八所示。由圖我們可以發現模擬與量測 結果滿一致的，如此可以驗證模擬結果的 正確性。
與使用四分之一波長短路槽線的共面 波導至矩形波導轉接比較[15]，使用電感補 償槽線的共面波導至矩形波導轉接，其 15-dB 的比例頻寬為 38.06%，幾乎涵蓋整 個 X-band (8.20–12.40 GHz)，可謂相當寬 頻，並且其可以將槽線的尺寸從 280 mil 縮小至 230 mil。
III. 微帶線至矩形波導轉接：
在這個部分，我們使用電容補償的寬 邊耦合微帶線，來實現縮小化且寬頻的微 帶線至矩形波導轉接，如圖九所示。圖上 所使用的電容補償寬邊耦合微帶線的長度 為 205 mil。接著我們使用 Ansoft HFSS 來 模擬這個架構，其返回與插入損耗的頻率 響應圖如圖十所示，由圖我們可知，返回 損耗大於 15 dB 的頻率範圍為 8.81–12.73 GHz，幾乎涵蓋整個 X-band，換算所得的 比例頻寬為 37.33%，可謂相當寬頻；另外，
在這個頻率範圍內，其插入損耗皆小於 0.16 dB。
為了驗證這個結果，我們實際製作一 個背對背電路，如圖十一所示。接著我們 使用校準後的網路分析儀來量測這個電 路，其模擬與量測的返回及插入損耗比較 圖如圖十二所示。由圖我們可以發現模擬 與量測結果滿一致的，如此可以驗證模擬 結果的正確性。
與文獻[1]、[4]、[5]比較，因為這個轉 接不需要額外的中繼轉接，因此所佔的電 路 面 積 比 較 小 ， 並 且 其 比 例 頻 寬 為 37.33%，幾乎涵蓋整個 X-band，可謂相當 寬頻。
四、結果與討論
本研究計畫完成三個縮小化且寬頻的 平面傳輸線至矩形波導轉接，研究可以區 分為三個部分。在第一部份，我們使用高 低阻抗線來達成縮小化且寬頻的槽線至矩 形 波 導 轉 接 ， 其 20-dB 的 比 例 頻 寬 為
4
36.57%，可謂相當寬頻，並且其電路長度 只有 190 mil。在第二部分，我們使用電感 補償的槽線來達成縮小化且寬頻的共面波 導至矩形波導轉接，其 15-dB 的比例頻寬 為 38.06%，可謂相當寬頻，並且其電路長 度只有 230 mil。在第三部分，我們使用電 容補償的寬邊耦合微帶線來達成縮小化且 寬頻的微帶線至矩形波導轉接，其比例頻 寬為 37.33%，可謂相當寬頻，並且其電路 長度只有 205 mil。為了驗證以上的結果，
我們實際製作與量測三個背對背轉接，量 測與模擬的結果滿一致的，可以驗證這些 結果的正確性。
五、參考文獻
[1] S. J. Nightingale, M. A. G. Upton, B. K.
Mitchell, U. K. Mishra, S. C. Palamateer, and P.
M. Smith, “A 30-GHz monolithic single balanced mixer with integrated dipole receiving element,” IEEE Trans. Microw.
Theory Tech., vol. MTT-33, no. 12, pp.
1603–1610, Dec 1985.
[2] T. Q. Ho and Y.-C. Shih, “Spectral-domain analysis of E-plane waveguide to microstrip transitions,” IEEE Trans. Microw. Theory Tech., vol. 37, no. 2, pp. 388–392, Feb. 1989.
[3] Y. Lou, Q. Xue, and C. H. Chan, “A broadband waveguide-to-microstrip transition power splitter using finline arrays,” IEEE Microw.
Wireless Compon. Lett., vol. 17, no. 4, pp.
310–312, Apr. 2007.
[4] N. Kaneda, Y. Qian, and T. Itoh, “A broad-band microstrip-to-waveguide transition using quasi-Yagi antenna,” IEEE Trans. Microw.
Theory Tech., vol. 47, no. 12, pp. 2562–2567, Dec. 1999.
[5] T.-H. Lin and R.-B. Wu, “A broadband microstrip-to-waveguide transition with tapered CPS probe,” in Proc. 32th Eur. Microw. Conf., Milan, Italy, Sep. 2002, pp. 1–4.
[6] H. Iizuka, K. Sakakibara, and N. Kikuma,
“Millimeter-wave transition from waveguide to two microstrip lines using rectangular patch element,” IEEE Trans. Microw. Theory Tech, vol. 55, no. 5, pp. 899–905, May 2007.
[7] G. E. Ponchak and R. N. Simons, “A new rectangular waveguide to coplanar waveguide transition,” in Proc. IEEE MTT-S Int. Microw.
Symp. Dig., Dallas, USA, May 1990, vol. 1, pp.
491–492.
[8] Y. Li, B. Pan, J. P. Becker, J. R. East, and L. P.
B. Katehi, “Fully micromachined finite-ground coplanar line-to-waveguide transitions for
W-band applications,” IEEE Trans. Microw.
Theory Tech., vol. 52, no. 3, pp. 1001–1007, Mar. 2004.
[9] Y. Li, B. Pan, C. Lugo, M. Tentzeris, and J.
Papapolymerou, “Design and characterization of a W-band micromachined cavity filter including a novel integrated transition from CPW feeding lines,” IEEE Trans. Microw. Theory Tech., vol.
55, no. 12, pp. 2902–2910, Dec. 2007.
[10] V. S. Möttönen and A. V. Räisänen, “Novel wide-band coplanar waveguide-to-rectangular waveguide transition,” IEEE Trans. Microw.
Theory Tech., vol. 52, no. 8, pp. 1836–1842, Aug. 2004.
[11] V. S. Möttönen, “Wideband coplanar waveguide-to-rectangular waveguide transition using fin-line taper,” IEEE Microw. Wireless Compon. Lett., vol. 15, no. 2, pp. 119–121, Feb.
2005.
[12] T.-H. Lin and R.-B. Wu, “CPW to waveguide transition with tapered slotline probe,” IEEE Microw. Wireless Compon. Lett., vol. 11, no. 7, pp. 314–316, Jul. 2001.
[13] C.-F. Hung, A.-S. Liu, C.-H. Chien, C.-L. Wang, and R.-B. Wu, “Bandwidth enhancement on waveguide transition to conductor backed CPW with high dielectric constant substrate,” IEEE Microw. Wireless Compon. Lett., vol. 15, no. 2, pp. 128–130, Feb. 2005.
[14] N. Kaneda, Y. Qian, and T. Itoh, “A broadband CPW-to-waveguide transition using quasi-Yagi antenna,” in Proc. IEEE MTT-S Int. Microw.
Symp. Dig., Boston, USA, Jun. 2000, vol. 2, pp.
617–620.
[15] R.-Y. Fang and C.-L. Wang, “A broadband coplanar waveguide to rectangular waveguide transition using a truncated bow-tie antenna,” in Proc. 37th Eur. Microw. Conf., Amsterdam, the Netherlands, Oct. 2008, pp. 468–471.
[16] R.-Y. Fang, C.-T. Wang, and C.-L. Wang,
“CPW-to-rectangular waveguide transitions using slot antennas,” IEEE Trans. Compon., Packag., Manuf. Technol., vol. 1, no. 5, pp.
681–688, May 2011.
[17] R.-Y. Fang, J.-K. Chuang, and C.-L. Wang,
“Coplanar waveguide-to-rectangular waveguide transition using meander slotline,” Asia-Pacific Microwave Conference 2011, pp. 399–402.
[18] R.-Y. Fang and C.-L. Wang, “Wideband slotline-to-rectangular waveguide transition using truncated bow-tie antenna,” in Proc.
Asia-Pacif. Microw. Conf., Yokohama, Japan, Dec. 2006, vol. 2, pp. 1395–1398.
[19] R.-Y. Fang and C.-L. Wang, “A broadband slotline-to-rectangular waveguide transition using a truncated bow-tie antenna,” IEEE Trans.
Compon., Packag., Manuf. Technol., vol. 1, no.
8, pp. 1154–1159, Aug. 2011.
5
圖一 使用高低阻抗線的轉接。
-0.6 -0.4 -0.2 0.0
8 9 10 11 12 13
-30 -20 -10 0
R et u rn Los s (d B )
Frequency (GHz)
Transition using the 75o SIRTransition using the quarter-wavelength transformer
Inse rtion L o ss (dB)
RL
IL
圖二 返回及插入損耗頻率響應圖。
圖三 背對背實際電路圖。
1.50 1.25 1.00 0.75 0.50 0.25 0.00
8 9 10 11 12 13
60 50 40 30 20 10 0
IL
Return Loss (dB)
Frequency (GHz)
Simulation IL Measurement IL Simulation RL Measurement RL RL
Insertion Loss (dB)
圖四 模擬與量測比較圖。
圖五 使用電感補償槽線的轉接。
8 9 10 11 12 13
40 30 20 10 0
IL
Frequency (dB)
R et u rn L o ss (d B )
RL
0.5 0.4 0.3 0.2 0.1 0.0
Insertion Lo ss (d B)
圖六 返回及插入損耗頻率響應圖。
6
圖七 背對背實際電路圖。
8 9 10 11 12 13
70 60 50 40 30 20 10 0
RL
Measurement RL Simulation RL Measurement IL Simulation IL
Frequency (GHz)
Re turn Lo ss ( d B)
IL
2.0 1.6 1.2 0.8 0.4 0.0
In sert ion Los s (d B)
圖八 模擬與量測比較圖。
圖九 使用電容補償寬邊耦合微帶線的轉 接。
8 9 10 11 12 13
30 20 10 0
IL
RL
Frequency (GHz)
Return Loss (d B)
2.0 1.5 1.0 0.5 0.0
Inser tion Los s (dB)
圖十 返回及插入損耗頻率響應圖。
圖十一 背對背實際電路圖。
8 9 10 11 12 13
60 50 40 30 20 10 0
Measurement RL Simulation RL Measurement IL Simulation IL
Frequency (GHz)
Re tu rn L o ss (d B )
4 3 2 1 0
IL
RL
In se rt ion Lo ss ( d B )
圖十二 模擬與量測比較圖。
出席國際學術會議心得報告
計畫編號 NSC 101-2221-E-011 -084
計畫名稱 微型且寬頻的平面傳輸線至矩形波導轉接 出國人員姓名
服務機關及職稱
王蒼容
台灣科技大學 電子工程系 副教授 會議時間地點 Oct. 24~Oct. 27, 2013, 羅馬尼亞 加拉茨
會議名稱 2013 IEEE 19th International Symposium for Design and Technology in Electronic Packaging (SIITME 2013)
發表論文題目
1. High-Low Impedance Transformer Using Transmission Line Method 2. Common-Mode Noise Suppression Using SMD Capacitor with Grounded
Via 一、參加會議經過
Oct. 25 (FRI):
參加 SIITME 2013 Opening ceremony, welcome words。
參加 SIITME 2013 Oral Session I。
參加 SIITME 2013 Oral Session II。
參加 SIITME 2013 Poster Session I。
參加 SIITME 2013 The Impact of Digital Era on Engineering Education。
參加 SIITME 2013 AFCEA Romania Poster Session。
參加 SIITME 2013 Poster Session II，並在會議中發表我們的文章：
1. High-Low Impedance Transformer Using Transmission Line Method
2. Common-Mode Noise Suppression Using SMD Capacitor with Grounded Via
Oct. 26 (SAT):
參加 SIITME 2013 Industrial Oral Session。
參加 SIITME 2013 Poster Session III。
參加 SIITME 2013 Poster Exercise Session。
Oct. 27 (SUN):
參加 SIITME 2013 Closing ceremony, looking forward to SIITME 2014。
二、與會心得
這次承蒙貴會補助，到羅馬尼亞加拉茨參加 2013 IEEE 19th International Symposium for Design and Technology in Electronic Packaging (SIITME 2013)，在會議中發表論文並參與數場 討論會，著實地令我獲益良多。首先，我們在會議中發表自己的論文，展現我們創新的研究 成果；在這個過程當中，我們亦和相關領域的專家學者互相討論，並且接受專家的建議與指
教，改良我們研究上的不足之處，使我們的研究更加地精進。另外，除了本身的研究領域之 外，我亦參加了數場討論會，以瞭解其他研究領域的進展，並且從中吸收其他領域的知識。
除此之外，大會也舉辦了廠商展覽會，讓我們可以對產業界的趨勢有所的瞭解。這樣子 的安排，除了讓我們可以學習到專業的知識，也幫助我們瞭解產業界的動態。因此能夠參與 這次的會議，真的是相當的值得。
最後深切地感謝貴會的補助，讓這次的國際研討會可以成行，希望在未來的年度當中，
可以再度受到貴會的補助，讓我們的研究可以在國際上發表，並且從參與國際會議之中學習 到更多的新知，以充實我們的研究與產業知識。
b 1 ( 1 )(B)
2013/11/12(B) b
Dear Authors,
I am pleased to inform you that your papers went through the paper review process with success and the authentication check resulted in similarity indices lower than 25%, therefore your paper would be offered for inclusion into the IEEE Xplore database, if your paper were presented by you or one of your coauthors at the SIITME2013 conference in Galati.
Please find attached the list of papers accepted until now.
Instructions for the poster preparation and the additional short oral presentation are displayed at the
www.siitme.ro website, and please find them copied below:
"Please take into consideration the following updated information about the POSTER requirements:
We are looking forward to your valuable contributions to SIITME2013 in Galați!
Z solt Illyefalvi-V itez
S IITM E Publication C hairB udapest U niversity of Technology and Econom ics
The poster must not exceed the A0 standard dimensions ( A0 – 841mm x 1189mm or 33.13″
x 46.84″)
The poster presentation format must be PPT, PPTX or PDF and the short oral presentation of the
poster must not exceed 4 minutes."
High-Low Impedance Transformer Using Transmission Line Method
Ruei-Ying Fang, Chia-Fen Liu, Chieh-Yu Liao, and Chun-Long Wang
Department of Electronic Engineering, National Taiwan University of Science and Technology Taipei, Taiwan
Abstract—In this paper, a simple transmission line method is proposed to precisely predict the center frequency of the compact high-low impedance transformer. It has been shown that the center frequency of the return loss stays at 3 GHz even though the electrical length θ
Treduces greatly. Besides, the compact high-low impedance transformer with θ
T= 60° yields 35.89%
area-saving and maintains the same 15-dB bandwidth as compared with the conventional quarter-wavelength transformer.
In order to verify the simulation results, back-to-back transformers are fabricated and measured where the measurement results are in good agreement with the simulation results.
Keywords—Compact, quarter-wavelength, transformer, high- low impedance
I. I NTRODUCTION
Quarter-wavelength transformers have been widely used in various microwave and millimeter-wave circuits, such as transitions [1], antennas [2], and etc, due to its simple structure and design rule. Although the quarter-wavelength transformer has been well developed and can be found in many literatures or textbooks [3]–[5], it occupies an electrically large area since its dimension is equal to a quarter-wavelength.
Many researchers have proposed various techniques to miniaturize the quarter-wavelength transformer [6]–[11]. First of all, the simplest way to miniaturizing the quarter-wavelength transformer is to spiral the quarter-wavelength transformer [6].
However, the insertion loss is increased due to the mutual coupling between adjacent spiral lines. Besides, artificial transmission lines [7]–[8] are used to miniaturize the quarter- wavelength transformer. However, the fabrication process is costly since multilayer structures are used.
Compact quarter-wavelength transformer can also be achieved by using the broadside coupled line [9]. However, the fabrication process is also costly due to the adoption of the multilayer structures. Besides, series or T-shape transmission lines are used to form the compact quarter-wavelength transformer [10]–[11]. However, miniaturization of the quarter- wavelength transformer is limited by constraining the transmission line impedances.
In order to freely miniaturize the quarter-wavelength transformer, a compact high-low impedance transformer is proposed and analyzed by using the SIR approximation.
However, the SIR approximation may cause an erroneous prediction of the center frequency [12]. In this paper, a simple
transmission line method is proposed to precisely predict the center frequency. The compact high-low impedance transformer, which is implemented with the microstrip high- low impedance transformer, can both greatly reduce the circuit size and maintains the bandwidth of the conventional quarter- wavelength transformer. In order to verify the simulation results, back-to-back transformers are fabricated and measured where the measurement results are in good agreement with the simulation results.
II. C ONVENTIONAL Q UARTER -W AVELENGTH T RANSFORMER
A. Design Rule
Fig. 1 shows the ideal transmission line (Tx) model of the conventional quarter-wavelength transformer where θ denotes the electrical length and Z
qdenotes the characteristic impedance of the quarter-wavelength transformer. By given the
This work was supported in part by Taiwan National Science Council under Grant NSC 97-2221-E-011-022.
l
R
LZ
0Z
inZ
qFig. 1 Ideal transmission line model of the conventional quarter-wavelength transformer.
2.0 2.5 3.0 3.5 4.0
60 50 40 30 20 10 0
RL (Ideal Tx-line) RL (Microstrip line) IL (Ideal Tx-line) IL (Microstrip line)
Frequency (GHz)
Re tu rn Lo ss ( dB)
6 5 4 3 2 1 0
Inser tion Loss (dB)
Fig. 2 Frequency responses of the return and insertion losses for the quarter- wavelength transformer with θ = 90°.
θ
load impedance R
Land the characteristic impedance Z
0as 143.22 and 50 Ω, respectively, the characteristic impedance Z
qcan then be calculated to be 84.62 Ω through the geometric mean of the load impedance R
Land the characteristic impedance Z
0[3]. The center frequency f
0of the quarter- wavelength transformer is chosen to be 3.0 GHz at which the electrical length θ is π/2 (90
°). By simulating Fig. 1 with the commercial software ADS, the frequency responses of the return and insertion losses for the ideal transformer are denoted by the ideal Tx-line as shown in Fig. 2.
B. Microstrip Quarter-Wavelength Transformer (θ = 90°) The ideal quarter-wavelength transformer shown in Fig. 1 is then implemented with the microstrip quarter-wavelength transformer shown in Fig. 3 where the substrate used to realize this circuit is FR4, having a thickness of 1.5 mm, a relative dielectric constant of 4.3, and a loss tangent of 0.02. The dimensions of the microstrip quarter-wavelength transformer shown in Fig. 3 are summarized in Table I.
The microstrip quarter-wavelength transformer shown in Fig. 3 is then simulated by using the commercial software ADS and the frequency responses of the return and insertion losses are also shown in Fig. 2. As can be seen from Fig. 2, the return loss of the microstrip transformer is in good agreement with the return loss of the ideal transformer whereas the insertion loss of the microstrip transformer is approximately 1 dB larger the insertion loss of the ideal transformer due to the inclusion of the dielectric loss.
C. Back-to-Back Verification
Since the line width of W
Lis so thin that it is impossible to solder an SMA connector, two microstrip quarter-wavelength transformers are connected back-to-back instead. The back-to- back microstrip quarter-wavelength transformer is fabricated on the FR4 substrate as shown in Fig. 4 where the dimensions are described by Table I and the line length of L
back1is 2 mm.
This back-to-back transformer is then measured with Agilent E8363B PNA after the equipment is calibrated with the SOLT calibration kit. The frequency responses of the measured insertion and return losses are shown in Fig. 5 along with the
frequency responses of the simulated insertion and return losses. As can be seen from Fig. 5, the simulation and measurement results are in good agreement for both the insertion and return losses.
III. C OMPACT H IGH -L OW I MPEDANCE T RANSFORMER
A. Design Rule
The ideal transmission line model of the compact high-low impedance transformer is shown in Fig. 6 where the characteristic impedance Z
0is assumed to be smaller than the load impedance R
L. The smaller characteristic impedance Z
0is connected to a high impedance line Z
hwith electrical length θ
hwhile the larger load impedance R
Lis connected to a low impedance line Z
lwith electrical length θ
l. The total electrical length of the compact high-low impedance transformer is denoted as θ
T= θ
h+ θ
l.
By observing Fig. 6, the partial reflection coefficients Γ
ibetween adjacent lines can be described as follows
0 0
0 h h
Z Z
Z Z
,
1l h
l h
Z Z
Z Z
,
2L l
L l
R Z
R Z
. (1) By enforcing Γ
0= Γ
2in equation (1) [3], we have
0
h l L q
Z Z Z R Z
, (2) where Z
qis the characteristic impedance of the quarter- wavelength transformer.
Besides, the input impedance Z
in2could be expressed as W
0W
LW
qL
qFig. 3 Top view of the microstrip quarter-wavelength transformer.
L
back1= 2 mm L
q= 13.93 mm L
q= 13.93 mm
Fig. 4 Photograph of the back-to-back microstrip quarter-wavelength transformer.
TABLE I
D
IMENSIONS OF THEM
ICROSTRIPQ
UARTER-W
AVELENGTHT
RANSFORMER(unit: mm)
W
0W
qL
qW
L2.92 1.04 13.93 0.20
2.0 2.5 3.0 3.5 4.0
60 50 40 30 20 10 0
Measurement RL Simulation RL Measurement IL Simulation IL
Frequency (GHz)
Return Loss ( dB)
6 5 4 3 2 1 0
In sertion Loss (dB)
Fig. 5 Comparison between the simulation and measurement results for the
back-to-back microstrip quarter-wavelength transformer.
2
( tan tan ) ( tan tan )
( tan tan ) ( tan tan )
L l h h l l l l h h
in h
l h l h l L h l l h
R Z Z jZ Z Z
Z Z
Z Z Z jR Z Z
. (3)
In order to have zero reflection at the input port, the input impedance Z
in2should satisfy Z
in2= Z
0. By substituting equations (2) and (3) into Z
in2= Z
0and applying the criterion for the minimum electrical length [13], we have
4 2 2 2 2 4 2
0
tan ( ) (
0) tan ( ) 0
2 2
T T
l q L q l q L
Z Z Z R Z Z Z Z R
. (4)
Equation (2) and (4) can then be used to solve for various combinations of the high impedance Z
hand the low impedance Z
lwith various total electrical length θ
T. Given the characteristic impedance Z
0= 50 Ω, the load impedance R
L= 143.22 Ω, and the center frequency f
0= 3 GHz, various combinations of the high impedance Z
hand the low impedance Z
lare listed in Table II. As can be seen from Table II, when the high impedance Z
his much larger than the low impedance Z
l, the electrical length θ
Tof the compact high-low impedance transformer is greatly reduced, which in turn saves the circuit size. For example, by comparing the case of θ
T= 60° with that of θ
T= 90°, the electrical length has a 33% reduction.
By simulating Fig. 6 along with the electrical lengths and impedance values listed in Table II, the frequency responses of the return losses for various electrical lengths θ
Tare shown in Fig. 7. As can be seen from Fig. 7, the 15-dB bandwidth of the return loss is almost the same even though the total electrical length θ
Treduces greatly from 90° to 10°. Besides, the center frequency of the return loss stays at 3 GHz even though the electrical length θ
Treduces greatly. For comparison, the frequency response of the return loss using the SIR approximation [12] is simulated and shown in Fig. 8. As can be seen from Fig. 8, the center frequency of the return loss moves far away from 3 GHz as the electrical length θ
Treduces greatly.
B. Microstrip High-Low Impedance Transformer (θ
T= 60°) With θ
T= 60°, the ideal high-low impedance transformer shown in Fig. 6 is implemented with the microstrip high-low impedance transformer as shown in Fig. 9. The dimensions of the microstrip high-low impedance transformer with θ
T= 60°
are summarized in Table III. As compared with the dimensions of the microstrip quarter-wavelength transformer shown in Table I, the length of the microstrip high-low impedance
Fig. 6 The ideal transmission line model of the compact high-low impedance transformer.
2.0 2.5 3.0 3.5 4.0
60 50 40 30 20 10 0
Ret urn Los s (dB)
Freqency (GHz)
T = 90 T = 80 T = 70 T = 60 T = 50 T = 40 T = 30 T = 20 θT = 10
θ θ θ θ θ θ θ θ
°
°
°
°
°
°
°
°
°
Fig. 7 The frequency responses of the return losses for various electrical lengths θ
T.
2.0 2.5 3.0 3.5 4.0
Re tu rn L os s (dB)
Frequency (GHz)
T=90
T=80
T=70
T=60
T=50
T=40
T=30
T=20
T=10 0
20
40
60
80
100
120
Fig. 8 The frequency responses of the return losses using the SIR approximation.
TABLE II
Parameters of the Compact High-Low Impedance Transformer θ
THigh Impedance Z
h(Ω) Low Impedance Z
l(Ω)
90° 84.62 84.62
80° 93.18 76.85
70° 105.44 67.92
60° 123.14 58.15
50° 149.13 48.02
40° 188.91 37.91
30° 255.33 28.05
20° 387.35 18.49
10° 780.37 9.18
Fig. 9 Top view of the microstrip high-low impedance transformer.
TABLE III
D
IMENSIONS OF THEM
ICROSTRIPH
IGH-L
OWI
MPEDANCET
RANSFORMER(unit: mm)
W
0W
hL
hW
lL
lW
L2.92 0.35 4.59 2.25 4.34 0.20
transformer is L
h+ L
l= 8.93 mm, which is much smaller than the length L
q= 13.93 mm of the microstrip quarter-wavelength transformer.
The frequency responses of the return and insertion losses for the microstrip high-low impedance transformer are shown in Fig. 10 along with the frequency responses of the return and insertion losses for the ideal high-low impedance transformer.
As can be seen from Fig. 10, the return loss of the microstrip transformer is in good agreement with the return loss of the ideal transformer whereas the insertion loss of the microstrip transformer is approximately 1 dB larger the insertion loss of the ideal transformer due to the inclusion of the dielectric loss.
As compared with Fig. 5, the 15-dB bandwidth of the return loss is almost the same.
C. Back-to-Back Verification
In order to verify the simulation results of the microstrip high-low impedance transformer, two microstrip high-low impedance transformers are connected back-to-back and fabricated on the FR4 substrate as shown in Fig. 11 with the line length of L
back2of 18.3 mm. This back-to-back transformer is then measured with Agilent E8362B PNA after the equipment is calibrated with the SOLT calibration kit. The frequency responses of the measured insertion and return losses are shown in Fig. 12 along with the frequency responses of the simulated insertion and return losses. As can be seen from Fig.
12, both the simulation and measurement results are in good agreement, which verifies our design
IV. C ONCLUSIONS
In this paper, a simple transmission line method is proposed to precisely predict the center frequency of the compact high- low impedance transformer. It has been shown that the center frequency of the return loss stays at 3 GHz even though the electrical length θ
Treduces greatly. Besides, the compact high- low impedance transformer can yield 35.89% area-saving and maintains the 15-dB bandwidth of the return loss as compared with the conventional quarter-wavelength transformer. In order to verify the simulation results, back-to-back microstrip high- low impedance and quarter-wavelength transformers are
fabricated and measured where the measurement results are in good agreement with the simulation results.
A CKNOWLEDGMENT
The authors would like to thank Wireless Communications
& Applied Electromagnetic LAB, National Taiwan University of Science and Technology, for providing the simulation environment and the measurement instruments. This work was supported in part by Taiwan National Science Council under Grant NSC 97-2221-E-011-022.
R EFERENCES
[1] Y. Lou, C. H. Chan, and Q. Xue, “An in-line waveguide-to-microstrip transition using radial-shaped probe,” IEEE Microw. Wireless Compon.
Lett, vol. 18, no. 5, pp. 311–313, May 2008.
[2] C. H. Chang and K. L.Wong, “Quarter-wavelength printed loop antenna with an internal printed matching circuit for GSM/DCS/PCS/UMTS operation in the mobile phone,” IEEE Trans. Antennas Propag., vol. 57, no. 9, pp. 2541–2547, Sep. 2009.
[3] D. M. Pozar, Microwave Engineering, 3rd ed., New York: Wiley 2005.
[4] D. K. Cheng, Field and Wave Electromagnetics, 2nd ed., Addison- Wesley Publishing Company 1989.
[5] S. M. Wentworth, Fundamentals of Electromatnetics with Engineering Applications, John Wiley & Sons 2005.
[6] H. Suzuki, N. Sugiyama, T. Sato, K. Yamasawa, Y. Miura, Y. Miyake, M, Akie, and Y. Uehara, “A thin film spiral microstrip transmission-line using CoZrNb soft magnetic thin film for a quarter wavelength transformer,” IEEE Trans. Magnetics, vol. 41, no. 10, pp. 3574–3576, Oct. 2005.
2.0 2.5 3.0 3.5 4.0
60 50 40 30 20 10 0
RL (Ideal Tx-line) RL (Microstrip line) IL (Ideal Tx-line) IL (Microstrip line)
Frequency (GHz)
Return Loss ( dB)
6 5 4 3 2 1 0
In sertion Loss (dB)
Fig. 10 Frequency responses of the return and insertion losses for the microstrip high-low impedance transformer with θ
T= 60°.
L
back2= 18.3 mm
L
h+L
l= 8.93 mm L
h+L
l= 8.93 mm
Fig. 11 Photograph of the back-to-back microstrip high-low impedance transformer.
2.0 2.5 3.0 3.5 4.0
60 50 40 30 20 10 0
Measurement RL Simulation RL Measurement IL Simulation IL
Frequency (GHz)
Re tu rn L oss (d B)
6 5 4 3 2 1 0
Insertion Loss (dB)
Fig. 12 Comparison between the simulation and measurement results for the
back-to-back microstrip high-low impedance transformer.
[7] H. Y. D. Yang, V. K. Chekka, and H. Ma, “Slow-wave transmission line transformers/baluns,” 2010 IEEE MTT-S Int. Microw. Symp. Dig., pp.
1536–1539.
[8] M. Gil, J. Bonache, I. Gil, J. Garcıa-Garcıa, and F. Martın,
“Miniaturisation of planar microwave circuits by using resonant-type left-handed transmission lines,” IET Microw. Antennas Propag., vol. 1, no. 1, pp. 73–79, Feb. 2007.
[9] M.-G. Chen, C.-W. Tang, T.-B. Hou, and J.-W. Wu, “Adopting the broadside coupled line for the design of an impedance transformer,”
2011 IEEE MTT-S Int. Microw. Symp. Dig., pp. 1–4.
[10] A. F. Sheta, A. Mohra, S. F. Mahmoud, “A new class of miniature quadrature couplers for MIC and MMIC applications,” Microwave and Optical Technology Letters, vol. 34, no. 3, pp. 215–219, Aug. 2002.
[11] S. C. D. Roy, “Matching characteristics of the physically short linear impedance transformer,” in IEE Proc. Microw. Antennas Propag., vol.
148, no. 2, pp. 137–139, Apr. 2001.
[12] R. Y. Fang, C. F. Liu, and C. L. Wang, “Compact stepped-impedance resonator transformer,” in Proc. IEEE 20th Conf. Electr. Perform.
Electron. Packag. Syst., Oct. 2011, pp. 303–306.
[13] C.-F. Chen, T.-Y. Huang, and R.-B. Wu, “Compact microstrip cross- coupled bandpass filters using miniaturized stepped impedance resonators,” in Proc. Asia–Pacific Microw. Conf., vol. 1, pp 4–7, Dec.
2006.
Common-Mode Noise Suppression Using SMD Capacitor with Grounded Via
Che-Yu Lin, Bao-Ren Huang, and Chun-Long Wang
Department of Electronics Engineering, National Taiwan University of Science and Technology Taipei City, 106, Taiwan
[email protected], [email protected], and [email protected]
Abstract—In this paper, a bended differential transmission line using the SMD capacitor with grounded via is investigated to suppress the common-mode noise. The bended differential transmission line using the SMD capacitor with grounded via can reduce the mode conversion from –5.34 dB to –11.67 dB and the TDT common-mode noise from 0.075 V to 0.024 V as compared with the bended differential transmission line using the parallel- plate capacitor. Besides, the circuit size is greatly reduced. In order to verify the simulation results, measurement is done in the frequency and time domains where the measurement results are in good agreement with the simulation results.
Keywords- Common-mode noise, SMD capacitor, grounded via, differential transmission line.
I. I NTRODUCTION
Differential signaling has been widely used in PCB layouts since it has low noise generation and high common-mode noise immunity [1]–[6]. However, in practical layouts, the differential transmission line may need to bend due to limited PCB space, which in turn results in an asymmetric structure.
The asymmetric structure will then induce the common-mode noise when the differential signal passes through the asymmetric structure, leading to serious EMI and SI problems.
To eliminate the common-mode noise, a common-mode suppression filter using periodically dumbbell-shaped defected ground structure (DGS) is proposed [7]. The periodically dumbbell-shaped DGS can filter out the common-mode noise without disturbing the differential-mode signal. However, the size is so big that it will consume the available area in the ground plane. In order to reduce the circuit size, another common-mode suppression filter using two U-shaped and one H-shaped DGS is proposed [8]. Although the circuit size is reduced, it still occupies about 0.44 λ
g× 0.44 λ
g. In order to further reduce the circuit size, the multilayered LTCC process is adopted to implement the common-mode filter [9]. Although the size of the filter can be reduced to 0.16 λ
g× 0.26 λ
g, ground bounce caused by the vias penetrating the multilayered PCB will be introduced.
Other than the common-mode suppression filters mentioned previously, a bended differential transmission line using the compensation capacitance is proposed [10]. However, its performance may be limited when it is implemented with the parallel-plate capacitor. Besides, a wideband common- mode suppression filter using the tightly coupled microstrip
line is proposed [11]. The tightly coupled microstrip line can greatly reduce the common-mode noise; however, the differential-mode transmission is slightly degraded at the low frequency range. Moreover, a bended differential transmission line using the compensation inductance is proposed [12].
Although this structure could also greatly suppress the common-mode noise, it occupies 0.10 λ
g× 0.10 λ
garea, which may be impractical in the industrial.
In this paper, a bended differential transmission line using the SMD capacitor with grounded via is investigated. Both the common-mode noise and the circuit size are greatly reduced. In order to verify the simulation results, measurement is done both in the frequency and time domains where the measurement results are in good agreement with the simulation results.
II. B ENDED D IFFERENTIAL T RANSMISSION L INE U SING P ARALLEL -P LATE C APACITOR
A. Differential-to-Common Mode Conversion
The bended differential transmission line using the parallel- plate capacitor is shown in Fig. 1(a) along with the cross- sectional view shown in Fig. 1(b) [10]. The bended differential transmission line using the parallel-plate capacitor consists of three parts where the first and last parts denote the coupled transmission (Tx) lines and the middle part denotes the parallel-plate capacitor. The dimensions for the bended differential transmission line using the parallel-plate capacitor shown in Fig. 1 are w=1.75 mm, s=0.75 mm, L=25 mm, and X=6.05 mm. The substrate used is FR4 with a relative dielectric
This work was supported in part under the Grand NSC 99-2221-E-011-061 by National Science Council, Taiwan.
(a) (b)
Fig. 1. Bended differential transmission line using the parallel-plate capacitor.
(a) Top view. (b) Cross-sectional view.
0 1 2 3 4 5 6 -50
-40 -30 -20 -10 0
Diff er ential- to-Common M o d e Con v ersion (dB )
Frequency (GHz) Simulation Measurement
Fig. 2. Comparison between the simulated and measured mode conversions for the bended differential transmission line using the parallel-plate capacitor.
constant of ε
r=4.3 and a thickness of h=1.5 mm. Given the dimensions, Fig. 1 is simulated by Momentum to extract the four-port S parameters and then the four-port S parameters are converted to the mixed-mode S parameters [13]. The simulated frequency response of the differential-to-common mode conversion is smaller than –5.34 dB from DC to 6 GHz as shown in Fig. 2.
In order to verify the simulation result, the bended differential transmission line using the parallel-plate capacitor is fabricated on FR4 as shown Fig. 3. This circuit is then measured with Agilent/E5071B VNA after this equipment is calibrated with the N4431A electronic calibration kit where the measurement result is also shown in Fig. 2. As can be seen from Fig. 2, the measurement result is in good agreement with
the simulation result, and the measured differential-to-common mode conversion is smaller than –6.58 dB from DC to 6 GHz.
B. Time-Domain-Through Common-Mode Noise
The bended differential transmission line using the parallel-plate capacitor shown in Fig. 2 is now simulated by using ADS where the driving source is a differential step function having a ±0.5 volt amplitude with a rise time of 40 ps.
The driving source is acquired from the waveform generated by the time-domain reflectometry (TDR) TEK/CSA8000 after the waveform has passed through two cables. It is noted that this driving source will include the cable loss, which will closely approximate the real source used in the measurement.
The simulated TDT common-mode noise for the bended differential transmission line using the parallel-plate capacitor is shown in Fig. 4 where the magnitude of the TDT common- mode noise is 0.075 V.
In order to verify the simulation result, Fig. 2 is measured in the time domain with TEK/CSA8000 where the measurement result is also shown in Fig. 4. As can be seen from Fig. 4, the simulation and measurement results are in good agreement where the magnitude of the measured TDT common-mode noise is 0.070 V.
III. B ENDED D IFFERENTIAL T RANSMISSION L INE U SING L- S HAPED P AD
In order to reduce the circuit size, a bended differential transmission line using the L-shaped pad is proposed as shown in Fig. 5. The dimensions shown in Fig. 5 are L
p=6mm, W
p=1mm, S
p=0.2mm, and R=0.3 mm. As compared with the bended differential transmission line using the parallel-plate capacitor shown in Fig. 1, the circuit size is greatly reduced since a small L-shaped pad is used to replace the large parallel- plate capacitor. The bended differential transmission line using the L-shaped pad shown in Fig. 5 is now simulated in the time domain by using ADS where the driving source is a differential step function having a ±0.5 volt amplitude with a rise time of
Fig. 3. Photograph of the fabricated circuit for the bended differential transmission line using the parallel-plate capacitor.
0.0 0.2 0.4 0.6 0.8 1.0
-0.04 -0.02 0.00 0.02 0.04 0.06 0.08 0.10
T D T C o mmon Mode N o ise ( V )
Time (ns)
Simulation Measurement
Fig. 4. Comparison between the simulated and measured TDT common-mode
noise for the bended differential transmission line using the parallel-plate
capacitor.
0 1 2 3 4 5 6 -60
-50 -40 -30 -20 -10 0
D if fe re nti a l- to- C om mon Mode C o nve rs ion ( d B)
Frequency (GHz)
Right-angle bendCompensation capactiance Cc =1.25 pF Compensation capactiance Cc =1.00 pF Compensation capactiance Cc =0.75 pF Compensation capactiance Cc =0.50 pF Compensation capactiance Cc =0.25 pF
