Circuit design

The rat race ring is a widely used 180^o hybrid coupler. Its drawbacks are its large size and limited bandwidth. The limiting factor in the hybrid ring coupler is the three-quarter wavelength section, which restricts the useful frequency range for the 180°

hybrid to f0 ± 0.23f0 where f0 is the center frequency in the band of interest . Numerous publications have been produced to improve the performance of this type of coupler.

Several design techniques have been proposed to enhance the useful bandwidth and to reduce its size. In [12] one quarter wavelength section with phase reversal was used to replace the three-quarter wavelength line. This phase-reversal section was realized with parallel coupled lines with two diagonal grounded ends (show in figure 4.2-1). Such a modified coupler achieved more than one-octave bandwidth. Although the bandwidth increases considerably, the even-mode impedance required for the coupled section would be too large to be realized, which is difficult to fabricate by using conventional chemical etching technique.

Figure 4.2-1 March's version of rat-race coupler

The conventional rat-race hybrids are inherently narrowband structures. This bandwidth limitation was attributed to the narrowband phase inverter within the quarter-wavelength and

3/4–wavelength line section. Many efforts have been made to make the bandwidth of rat-race ring coupler larger [13][14]. Most of them pay their attention by making a broadband phase inverter.

The finite- ground-plane CPW (FCPW) is a good candidate to realize this ideal phase inverter, because the “hot” lines as well as the ground planes are located on the upper surface of the carrier material. This enables parallel implementation of active and passive lumped elements into the circuit without any via hole structure, which results in a significant simplification in manufacturing process.  

Figure 4.2-2 Twist between signal and ground path

Even if an ideal phase inverter is used, the conventional rat-race ring coupler with 70.7Ω ring impedance and Butterworth-type response is still limited to of about 74%

fractional bandwidth for return loss better than 15 dB. The ring coupler contain an ideal phase inverter can show a Chebyshev-type response of order two. For 12 dB return loss, the ring coupler with 55Ω ring impedance has 100% fractional bandwidth.

Because of the crossover at the ring arm, the signal line of the rat-race ring is DC grounded. We must use RF virtual ground to terminate two diodes and the IF signals are picked up from these RF virtue ground. The design parameters and circuit dimensions are list in Table 4.2-1.

Table 4.2 Design parameters and circuit dimensions of the rat-race ring

This single-balanced mixer is fabricated on 15mil Al2O3 substrate with center frequency 4GHz and 100% fractional bandwidth. The RF virtual ground can be formed by an FCPW open stub or shunt a capacitor. Here we used a 2pF capacitor at the IF port.

The diodes used in the mixer are Metelics silicon schottky diodes (MSS30,242-B20). In view of the layout of the IQ mixer, we squeeze the single-balanced mixer. The effect of the bond wires at crossover can be compensated by shortening the corresponding ring arms about 5 mils.

4.3 Measurement

There are two ways to excite the single-balanced mixer. One is to excite Local source at sum port and the other is to excite it at difference port. The measurement results are shown in the following figures respectively.

Figure 4.3-1 Conversion loss versus RF frequency. ( Local at difference port )

Figure 4.3-2 LO to RF isolation and RF to IF isolation ( Local at difference port )

Figure 4.3-3 Conversion loss versus IF frequency for fixed LO frequency.

( Local at difference port )

Figure 4.3-4 Conversion loss versus LO power

Figure 4.3-5 P1dB ( Local at difference port )

Figure 4.3-6 Conversion loss versus RF frequency. ( Local at sum port )

Figure 4.3-7 LO to RF isolation and RF to IF isolation ( Local at sum port )

Figure 4.3-8 Conversion loss versus IF frequency for fixed LO frequency.

( Local at sum port )

The conversion loss is around 8dB over the desired RF frequency band and it’s better when RF signal is excited at difference port. In this case, RF signal is virtual ground at IF port that causes the voltage across the diodes would be larger, hence the conversion loss would be smaller.

The RF to LO isolation is provided by the ring because RF port is also the virtual ground of the LO signal and vice versa. From figure 4.3-2, the RF to IF isolation is poor compared with figure 4.3-6. Because RF and IF are both placed at sum port, the ring provides no inherent isolation. The isolation here is only provided by a 2pF shunt capacitor. One way to solve this problem is to use a sharp low pass filter rather than just one order at IF port. From figure 4.3-6, the RF to IF isolation is better when RF is placed at difference port because the IF port is the virtual ground of the RF signal and the effect of the capacitor (RF-short circuit) The photograph of the circuit is shown in figure 4.3-8.

  Figure 4.3-9 photograph of the proposed single-balanced mixer

Chapter 5 I/Q mixer

In this chapter, three device ( coupler, power divider, single-balanced mixer) are constituted to form the I/Q mixer. The whole circuit is fabricated in a one inch by one inch Al2O3 substrate. The measurement result will display in the following section. The photograph of the circuit is shown in figure 5-1.

Figure 5-1 The photograph of the I/Q mixer  

5.1 Measurement

Figure 5.1-1 shows the conversion loss versus RF frequency. The conversion loss at the desired frequency band is around 12dB. The CH1 and CH2 are the outputs of the I/Q mixer which is shown in figure 5-1.

Figure 5.1-1 Conversion loss versus RF frequency ( LO (11dBm) is excited from 90 degree hybrid and RF (0dBm) is excited from the power divider )

Figure 5.1-2 shows the conversion loss versus RF frequency The conversion at the desired frequency band is around 12dB too.

  Figure 5.1-2 Conversion loss versus RF frequency ( LO (11dBm) is excited from the

power divider hybrid and RF (0dBm) is excited from 90 degree hybrid )  

  Figure 5.1-3 Conversion loss versus LO power ( LO from 90o hybrid ; RF from power divider ;IF=5 MHz ; RF<LO )

  There are two methods to check that the two outputs of the I/Q mixer have 90 degree phase difference. First, we can just excite the desired RF and LO signal in a fixed frequency and see the result. If LO is 2GHz and RF is 1995MHz. Two outputs of the I/Q mixer is shown in figure 5.1-4.

Figure 5.1-4 LO = 2GHz; RF = 1995 MHz ; IF = 5 MHz

As can be seen the two outputs has a phase difference 90 degree. The following figures show the results with different LO and RF frequency.

Figure 5.1-5 LO = 3GHz; RF = 2995 MHz ; IF = 5 MHz

Figure 5.1-6 LO = 4GHz; RF = 3995 MHz ; IF = 5 MHz

Figure 5.1-7 LO = 5GHz; RF = 4995 MHz ; IF = 5 MHz

Figure 5.1-8 LO = 6GHz; RF = 5995 MHz ; IF = 5 MHz

The second method can be shown in figure 5.1-9.

Figure 5.1-9 Second method to check the phase difference between two outputs of I/Q mixer

The RF and LO of the I/Q mixer is excited by a signal which its frequency change with time. The two outputs will produce DC voltage (cosθ , sinθ) because RF frequency is equal to LO frequency. The delay line in the upper path is essential for this method, because it introduce a phase delay. The phase delay is changed with frequency of the input signal because a same delay line saw different electric length at different frequency. Hence the DC outputs will also change with time. Figure 5.1-10 show the measurement result of second method.

Figure 5.1-10 Measurement result of second method.

The difference between the first method and the second method is that the first method can only see the phase difference and amplitude balance at some fixed frequency. The second method can see the results over all frequency components.

Chapter 6 Conclusion

This thesis has demonstrated a minimized wideband I/Q mixer on a one inch by one inch Al O substrate with a dielectric constant of 9.8.

In chapter 2, miniaturized wideband quadrature hybrid coupler has been realized by using three-section cascaded CPW coupler structure. The return loss would be better if the meander ground in the center section is used. The way to tune the meander ground is to let the input impedance around 50Ω, because the deep in the input impedance diagram can change if the dimension of the meander is changed. In consider of the area limitation, the length of the coupler is shrinked and the center frequency is shifted to higher frequency.

In chapter 3, a wideband wilkinson power divider is fabricated. Because the circuit is in CPW based, the coupling between two paths can be reduce by insert a ground strip between them. Hence, the circuit size is reduced.

In chapter 4, a single-balanced mixer is fabricated. The 180 degree hybrid in the mixer is chosen as a rat race ring coupler. To realized a wideband rat race ring, we can replace the 3/4 wavelength transmission line by a 1/4 wavelength transmission and a phase inverter which not only widen the bandwidth but also reduce the circuit size. To further reduce the circuit size, the diodes is placed inside the ring and the RF virtual ground is form by a shunt 2pF capacitor.

In chapter 5, the circuits fabricated in previous chapter are combined to form an I/Q mixer. There are two ways to check the phase and magnitude difference between two outputs of I/Q mixer. The measurement result is shown in figures and the specification is achieved.

References 

[1]E. G. Cristal and L. Young, “Theory and tables of optimum symmetrical TEM-Mode coupled transmission line directional couplers,” IEEE Trans. Microwave Theory Tech., vol. MTT-13, no. 5, pp. 544–558, Sep. 1965

[2]J. S. Izadian, “A new 6–18 GHz, 3dB multi-section hybrid coupler using asymmetric broadside, and edge coupled lines,” in IEEE MTT-S Int. Microwave Symp. Dig., 1989, pp. 243–246.

[3] G. Kemp, J. Hobdell, and J. W. Biggin, “Ultra-wideband quadrature coupler,”

Electron. Lett., vol. 19, no. 6, pp. 197–198, 1983.

[4] S. Uysal and H. Aghvami, “Synthesis, design, and construction of ultra-wide-band nonuniform quadrature directional couplers in inhomogeneous media,” IEEE Trans.

Microwave Theory Tech., vol. 37, pp. 969–975, June 1989.

[5] “Improved wideband 3dB nonuniform directional coupler,” Electron. Lett., vol. 25, no. 8, pp. 541–542, Apr. 1989.

[6] Duncan K.Y.Lasu, Steve P.March, Lionel E.Davis and Robin Sloam, “Simplified design technique for high performance microstrip multi-section couplers”, IEEE Trans.

Microwave Theory Tech., Vol.46, No.12, PP.2507-2513, Dec.1998.

[7] T. Y. Chang, C. L. Liao, and C. H. Chen, “Novel uniplanar tandem couplers,” in Proc. 32nd Eur. Microwave Conf., 2002, pp. 127–130.

[8] R. M. Osmani, ‘Synthesis of Lange couplers,’ IEEE Trans. Microwave Theory and Techn.

Vol. 29, Feb. 1981, pp

[9] Wilkinson, E.J., “An N-way hybrid power divider,” IRE Trans. Microw. Theory Technol., 1960, MTT-8, pp. 116–118

[10] S. B. Cohn, “A class of broadband three-port TEM-mode hybrids, ”. IEEE Tram.Microwave Theory Tech., vol. MTT-16, Feb. 1968, PP. 110-116

[11] Walker, J.L.B. ”Improvements to the Design of the 180 Rat Race Coupler and its 　 Application to the design of Balanced Mixers with High LO to RF Isolation”, IEEE MMT-S Digest, 1997, Vol. II, pp 747-750

[12]S. March, “A wide band stripline hybrid ring,” IEEE Trans. Microwave Theory Tech., vol. MTT-16, p. 361, June 1968.

[13]T. Wang Z. OU and K. Wu, “Experimental study of wideband uniplanar phase inverters for MICs” in IEEE MTT-S Int. Microwave Symp. Dig., 1997, pp777-780

[14]T. Wang and K. Wu, “Size-reduction and band-broadening design technique of uniplanar hybrid ring coupler using phase inverter for M(H)MIC’s,” IEEE Trans.

Microw. Theory Tech., vol. 47, pp. 198-206, Feb. 1999.