For the requirement of compact antennas and decoupling techniques, a series of miniaturized antennas and decoupling methods has been developed. The operational principles and theories of these designs have been studied and demonstrated by using equivalent circuits. The equivalent circuit has also been proved of a useful method for compact antenna design.
In miniaturized antenna design, there are four kinds of antennas being proposed.
To further understand the miniaturization methods, two different concepts were considered, the concept involved wavelengths for resonance and circuit approach instead of wavelengths for resonance. An important task of this dissertation is to prove that circuit approach method can bring compact and efficient antennas in an efficient way. About different concepts, both lead to successful designs.
The first proposed antenna by spiraling the conventional inverted-F antenna has 50%-size reduction and maintains the radiation efficiency. Also, the dual-band mechanism, which utilizes two resonant modes in one resonator, has been explained by using the equivalent circuit. This design successfully achieves the miniaturization and dual-band operation simultaneously. The antenna size of 9.5 mm by 6.5 mm for 2.4/5.2GHz band that is compact for most applications. The second proposed antenna by using dielectric load, with relative permittivity of 40, on QHA has great size reduction to 2.7% of air-loaded one. The related equivalent circuit has also explained impedance feature of self-phasing circular polarized QHA and the observed impedance curve can help to tune the circular polarization state. However, the small antenna size causes small radiation resistance that shrinks the impedance bandwidth.
The matching structure was proposed for impedance matching. For narrow bandwidth operation of GPS, the proposed small QHA is sufficient.
The third proposed antenna adopts the circuit approach design. The resonance is achieved by circuit synthesis based on the concept of cascaded right/left handed transmission line with opposite phase delay. The antenna layout is formed by using printed elements corresponding to the equivalent lumped circuit of transmission line.
In fact, this method can be considered as a sub-category of the general π and T models.
From experiments, it is proved that the resonant frequency can be synthesized by the circuit design without being affected by ground size. Also, the radiation mechanism was discussed. The most radiation from slot radiator has been confirmed. The circuit approach design has many design parameters thus flexible. The proposed type for 2.4GHz is finished inside 11.5 mm by 11.5 mm that is slightly larger than spiraled inverted-F antenna because of the size of printed capacitors. However, it proved the design concept of circuit approach and it can be further reduced by changing design parameters. Based on the short slot radiator that is proved of efficient radiation, the fourth antenna by using SMD elements on the slot has smaller size than using printed element for the entire layout. The design still starts with the equivalent circuit. The radiation resistor is considered in this design. The developed circuit structure of this antenna is simpler than the third antenna thus saving layout elements. The antenna for 2.4GHz band only with a short slot radiator (9 mm by 1.5 mm) and two SMD capacitors is more compact than previous designs. From the experiments, the equivalent circuit has shown its accuracy for antenna design. The impedance bandwidth is sufficient for IEEE802.11b/g. The radiation efficiency is fairly good according to the measured radiation patterns and gains. It is worth to notice that the frequency tunable type is developed from the same configuration. The lower frequency was achieved in this type using same slot radiator. Although the tuning range is determined on varactors, the realized antenna has measured tuning range from 1.7GHz to 2.8GHz. It means the proposed antenna configuration is very flexible and compact. All proposed miniaturized antennas have revealed their equivalent circuit model. Developing different kinds of miniaturization methods helps to understand the features of the miniaturized antenna more deeply.
The proposed miniaturized decoupling structures also follow the concept of circuit approach. Based on the LC parallel resonator, the ground edge current choke (GECC) has been developed using printed capacitor and inductor. Its function of blocking ground edge current is proved from the experiments of two presented applications. The decoupling function in 5GHz has been evaluated successfully. The simulated current distribution shows that it prevents the nearby antenna from the coupling current induced by the driven antenna. The GECC for 5GHz band is 3.5 mm
reduction and more flexible use, the decoupling circuit is developed. It can enhance the port isolation for any two nearby antennas that solves the efficiency problem due to the coupling. The circuit structure is compact that consists of two segments of transmission line and a lumped element. The formulas for circuit design have also been developed. In a series of experiments, the circuit structure has been proven that the decoupling function works and the formulas can design the circuit. The proposed two decoupling methods are compact. Their flexible design can be used in most applications.
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Appendix A Decoupling Circuit Network
In this section, a miniaturized structure to achieve high port isolation between two antennas is proposed. The decoupling structure is accomplished on the circuit board. Formulas for decoupling circuit construction are developed. The required parameters of the structure are derived based on the measured or simulated coupling coefficient between antennas. The circuit-approach decoupling can achieve good isolation between two antenna input ports. However, the driving antenna can still induce the current on the other antenna although the power will not deliver to its port terminal. The driven current on antenna has also be studied in this section.
A1. Operational Principle
It is usually easy to design the input impedances but hard to reduce the coupling between two closely spaced antennas. In this study, a dual antenna system (Ant.1 and Ant. 2 in Figure A.1) with good input impedance matching but poor port isolation is first assumed. For simplicity, the antennas are symmetrical to each other and have input impedance of Z0 (= 50Ω). A four-port decoupling network is proposed, with two output ports connected to the antennas, for reducing the coupling between the two resultant new input ports. Each input port is in turn connected to a matching network for improving the input impedance. Figure A.1 shows the function blocks of the decoupling structure.
The decoupling network consists of two transmission lines with characteristic impedance Z0 and electrical lengthθ and a shunt reactive component with admittance
jB. Let the scattering matrix of the coupled antennas be denoted as [S
A] at the reference plane t1. After connecting the transmission lines, the new scattering matrix at the reference plane t2 is expressed as [SA’]. [SB] is the scattering matrix at the reference plane t3 after the addition of the shunt lumped element. Finally, the total scattering matrix at the reference plane t4, including the antennas, the decoupling network, and the matching networks, is indicated as [S].jB lines, a shunt reactive component, and two impedance matching networks.
Since the coupled antennas are assumed with good input matching, the diagonal coefficient experiences an extra phase delay of 2
θ. Thus, the scattering matrix at t
2can be described as
[ ]
Once this scattering matrix is known, the corresponding admittance matrix [Y
A’] can be easily derived [21].
As shown in Figure A.1, the two-port network seen at t
2is in shunt with a
reactive element of susceptance B , and thus the resultant new two-port network, i.e.,
that with ports at t
3, should have an admittance matrix [Y
B] equal to
component,
through the following formulas:
21
coefficient S
21Bat t
3should be zero, which means that, from (A-7), the Both of these values are physically realizable, since both positive and negative values of
Bare possible (positive
Bimplies a capacitor and negative implies an inductor).
However, a shunt capacitor is preferred, since a negative B corresponds to a negative θ (
φis negative due to close spacing between antennas), which means longer transmission lines (longer than half wavelength) are required.
The solution of (A-9) is quite obvious, which shows that the coupling coefficient
after the introduction of the transmission lines should possess a phase (−2θ φ+ ) equal to m 90o. As revealed in (5), when this condition holds, the trans-admittance Y21A’
at t2 would be pure imaginary, and can thus be cancelled by the reactive component jB.
The transmission lines have the functions of not only delay lines of connecting the antenna input ports to the decoupling network, but also transferring the complex trans-admittance at the coupled antennas to a pure imaginary one. The solutions of (A-9) and (A-10) can be cast into (A-6) to get the input admittances Y11B
and Y22B
A2. Derivation of Antenna Driving Currents
jB corresponding even-mode circuit, and (c) the odd-mode circuit.
Due to the close coupling between the two antennas and the introduction of the decoupling network, both the antennas will be excited even if the input power is only fed to one port. The antennas form a two-element array, with the radiation pattern determined by the excited currents on the antennas. Consider now that port 1 is fed by a current source and port 2 is terminated by Z0. Let the input current, after passing the impedance matching network, at the reference plane t3 (or point B1 of
Figure A.2(a)) be denoted as I . Note that since at point B+in 1 the impedance is not matched, this input current would cause a returning current propagating back to the impedance matching network. In order to estimate the radiation pattern, the resultant driving currents IA1 and IA2 at the input points (points A1 and A2) of the two coupled antennas are to be derived through the even and odd modes analysis. To this end, the two-port coupled antennas are first modeled by a T network. Then, by setting the connecting points at the symmetric plane as open-circuits for the even mode and short-circuits for the odd mode, the even-mode and odd-mode circuit schematics can be obtained as shown in Figure A.2(b) and (c), respectively. The even-mode (ZeL) corresponding reflection coefficients at the antenna input points are
e Similarly, the odd-mode driving current IoA is expressed as
in L
where which can be easily derived by modeling the two-port network shown in Figure A.2(a) as a π network, and and using the relationships of (A-9), (A-10), and (A-18), the ratio of driving currents at the antenna input points can be derived as a function of the coupling coefficient:
A1 A A
A3. Experimental Results of Single-band Solution
Two examples are tackled in this study. One is a dual-antenna system with two closely spaced parallel printed monopole antennas, and the other is with two miniaturized printed antennas. The antennas, operating at the frequency of 2.45 GHz, were all implemented on the FR4 substrates with the dielectric constant of 4.5, the loss tangent of 0.02, and the thickness of 0.8 mm. Both the EM simulator HFSS and the circuit simulator AWR Microwave Office were used for the simulation. The former handles the full-wave simulation for the antenna structure, with the results cast to the latter, if needed, for the following lumped-element related simulation.
Via hole Lumped element
Ant.1 Ant.2
Port1 Port2 La
W L
S
Via hole Lumped element
Ant.1 Ant.2
Port1 Port2 La
W L
S
Figure A.3 The configuration of the two closely spaced printed monopole antennas.
The first example is two parallel printed monopole antennas with length La and spacing S = 8.5 mm (0.069
λ
0 at 2.45GHz), as shown in Figure A.3. The antennas are fed by two 50Ω
microstrip lines of width 1.5 mm on a substrate with ground size ofL × W = 45 mm × 22 mm, which is suitable for a general USB dongle. By using the
full-wave simulator HFSS, the antenna dimensions are first designed to be with lengthL
a= 22.5 mm and metal strip width of 1.5 mm for good input impedance matching.Figure A.4(a) plots the results of simulated S-parameters from 2 GHz to 3 GHz for the two coupled antennas in the complex plane. It is seen that at this stage, the reflection coefficient S11 (and S22) is close to the origin of the coordinate around the center frequency, but the coupling coefficient S21 is not. This means that the two antennas exhibit input impedances close to 50
Ω
while have poor isolation between them. The coupling coefficient S21 at 2.45GHz is with an amplitudeα = 0.77 and a phase φ = -25o. Figure A.4(b) and (c) depict, respectively, the return loss (1/S11) and the isolation (1/S21) as functions of the frequency. Both the simulation and measurement results are shown, which show good agreement with each other. The measured 10-dB return-loss bandwidth is 15.1% from 2.27 GHz to 2.64 GHz and the worst isolation in the in-band (from 2.4 GHz to about 2.5 GHz) is only 3 dB.Re Im
-1
-1 1
1 Sweep 2~3GHz
S11 S21
2.45GHz
S11 S21
2.45GHz
(a)
2 2.2 2.4 2.6 2.8 3
Frequency (GHz)
Frequency (GHz)