Chapter 2 Planar Ultra-wideband Antennas
2.2 Planar Ultra-wideband Antenna with a New Band-Notch Structure
2.2.2 Antenna Design
In this section, the antenna covering the full UWB band (3.1-10.6 GHz) is first described. Then, the new band-notched structure, which is equivalent to a parallel LC circuit, is investigated. The effects with respect to the geometric parameters of the proposed antenna on impedance bandwidth and radiation pattern are discussed. The proposed antenna structure is simulated using the HFSS.
A. Full band UWB Antenna Design
At low frequencies, the current is mainly distributed over the radiation patch and the ground plane, which is similar to the current of a printed finite-ground monopole antenna. Thus, increasing the patch length La is equal to increasing equivalent current
length and decreasing the resonant frequency. Figure 2.10 shows the simulated return losses for La varied from 7 to 13 mm. It can be seen that the edge of low frequency decrease as La increase. When La varies from 7 to 13 mm, the low frequency edge moves from 3.25 to 2.75 GHz.
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10 0
La = 7 mm La= 9 mm La= 11 mm La= 13 mm
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10 0
La = 7 mm La= 9 mm La= 11 mm La= 13 mm
Figure 2.10 Simulated return losses for the proposed antenna with various patch length La. Ws = Ls = Wc = We = r = 0 mm. Other geometric parameters are the same as given in Figure 2.9.
The ground plane of the proposed antenna is also a part of the antenna. The current distribution on the ground plane affects the characteristics of the antenna. The monopole antenna as well as the ground plane forms an equivalent dipole antenna.
Figure 2.11 shows the effects of varying the ground plane length Lg (Lg = 7.4, 9.4, 11.4, and 13.4 mm) on the simulated return losses, with La = 13 mm. In Figure 2.11, the edge of low frequency decreases as Lg increases, the behavior is similar to changing La. When Lg varies from 7.4 to 13.4 mm, the edge of low frequency moves from 3.5 to 2.75 GHz.
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10 0
Lg= 7.4 mm Lg= 9.4 mm Lg= 11.4 mm Lg= 13.4 mm
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10 0
Lg= 7.4 mm Lg= 9.4 mm Lg= 11.4 mm Lg= 13.4 mm
Figure 2.11 Simulated return losses for the proposed antenna with various ground plane length Lg. Other geometric parameters are the same as given in Figure 2.9.
The gap between the radiation patch and the ground plane has an important effect on the impedance matching of the proposed antenna, as shown in Figure 2.12. When the gap g is increasing from 0 to 0.6 mm, the impedance matching at low frequencies can be greatly improved, at the expense of little deterioration in high frequency matching.
By comparing Figure 2.10 to Figure 2.12, it is found that La, Lg, and g are principally relevant to the low frequency characteristics, but not the high frequency performance. The reason is that in the low frequency band, the proposed antenna acts like as a printed monopole (or dipole) antenna, while in the high frequency band, the antenna behavior is like a slot antenna. Hence, properly designing the shape of the two bevels between the patch and ground plane will enhance the slot mode radiation and improve the impedance matching in high frequency band.
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10 0
g = 0 mm g = 0.2 mm g = 0.4 mm g = 0.6 mm
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10 0
g = 0 mm g = 0.2 mm g = 0.4 mm g = 0.6 mm
Figure 2.12 Simulated return losses for the proposed antenna with various gap g. Other geometric parameters are the same as given in Figure 2.9.
Figure 2.13 and Figure 2.14 show the simulated return losses for various bevel sizes of the ground plane. It is clearly seen that changing Ws or Ls is an efficient way to improving the input impedance matching, especially at the high frequency. For the case of the bevel size Ws = Ls = 0 mm, which means no bevel on the ground plane, the bandwidth is not sufficient. Properly choose Ws and Ls, a widest bandwidth can be obtained. From the simulated results in Figure 2.13 and Figure 2.14, it occurs when
W
s = 7 mm and Ls = 3.7 mm.Frequency (GHz)
Figure 2.13 Simulated return losses for the proposed antenna of various bevel length Ls with a fixed value of Ws = 7mm. Other geometric parameters are the same as given in Figure 2.9.
Frequency (GHz)
Figure 2.14 Simulated return losses for the proposed antenna of various bevel width Ws with a fixed value of Ls = 3.7mm. Other geometric parameters are the same as given in Figure 2.9.
Additionally, two semicircle slots cut in the bottom side of the ground plane may further improve the antenna performance. The effects of different radii and positions of the semicircle slots were investigated. The simulated return losses for various sizes and positions of the semicircle slot are shown in Figure 2.15 and Figure 2.16. It can be seen in Figure 2.15 that the return loss curves have similar shapes for the three different slot radii (r = 2.0, 2.5, and 3.0 mm) at low frequencies, but the high frequency impedance matching changes significantly with the variation of r. In Figure 2.16, when We becomes larger (i.e., becoming farther from the side edge of the ground plane), the high frequency matching is slightly improved.
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10 0
r = 0 mm r = 2.0 mm r = 2.5 mm r = 3.0 mm
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10 0
r = 0 mm r = 2.0 mm r = 2.5 mm r = 3.0 mm
Figure 2.15 Simulated return losses for the proposed antenna of various slot radii r with a fixed value of We = 3 mm. Other geometric parameters are the same as given in Figure 2.9.
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10 0
without slot We= 1 mm We= 2 mm We= 3 mm
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10 0
without slot We= 1 mm We= 2 mm We= 3 mm
Figure 2.16 Simulated return losses for the proposed antenna of various distance We with a fixed value of r = 3 mm. Other geometric parameters are the same as given in Figure 2.9.
The radiation pattern in low frequency band is omnidirectional, but it usually deteriorates in the high frequency region. It is because that at the high frequencies, the magnetic currents mainly distributed over the slots between the radiation patch and the ground plane. The waves travel through the slots cause directional radiation patterns in the horizontal plane (i.e., the xz-plane). By introducing these two semicircle slots, the transverse currents on the ground plane near the bottom side diminish. Thus the waves radiated from the slot propagate in a more omnidirectional way. Also, since the currents on the ground plane at high frequencies are rectified with the insertion of the two semicircle slots, more power is fed into the slots between the patch and the ground plane. As the results, the return-loss bandwidth of the antenna is broadened, and the gains in high frequency band become larger. Figure 2.17 shows the comparison of the simulated 3-D radiation patterns with and without the semicircle slots at 9 GHz. In Figure 2.17, the radiation pattern with semicircle slots is more omnidirectional than that without slots in the horizontal plane (i.e., the xz-plane).
x y
x y
z z
without semicircle slots with semicircle slots x
y
x y
z z
without semicircle slots with semicircle slots
Figure 2.17 Simulated 3-D radiation patterns with and without semicircle slots in the ground plane of the proposed antenna at 9 GHz.
B. UWB Antenna with Band-Notched Function Design
The frequency range for UWB systems approved by the FCC is between 3.1 GHz to 10.6 GHz. It might cause interference to the existing wireless communication systems, for example the WLAN operating in 5.15-5.85 GHz. Therefore, the UWB antenna with a band-notched characteristic is required. To obtain the band-notched function, there are various methods to achieve it as described in Chapter 1. The conventional methods are cutting a slot on the patch [46]-[54], inserting a slit on the patch [55]-[57], embedding a quarter-wavelength tuning stub within a large slot on the patch [58], or using the SRR structure on the patch [59]-[60]. Another way is putting parasitic elements near the printed monopole as filters to reject the limited band [61]-[63] or introducing a parasitic open-circuit element, rather than modifying the structure of the antenna’s tuning stub [64]. Changing the feed structure is also a method to achieve the band-notched response such as using the lumped and distributed inductors and capacitors integrated on the top side of the substrate in front of the feed port [65] and inserting two quarter-wavelength tuning stubs or a resonance cell into the proposed feeding [66]-[67].
In our design, the concept of the parallel LC circuit is applied. At resonant frequency, the parallel LC circuit will cause high input impedance that leads to the desired high attenuation and impedance mismatching near the notch frequency. In this section, a pair of T-shaped stubs is embedded inside an elliptical slot cut in the radiation patch to form the parallel LC circuit. The elliptical slot and the T-shaped stubs are equivalent to an inductor and a capacitor, respectively. By adjusting the inductor and capacitor values, the suitable notch frequency and bandwidth can be achieved.
Figure 2.18 shows the simulated return losses for various axial ratios (AR) of the elliptical slot with the minor axis fixed at 2.3 mm. It is seen that, increasing the axial ratio, which is similar to increasing the inductor value of the parallel LC circuit, has the effects of adjusting the center notch frequency as well as increasing the notch bandwidth. When AR varies from 1.8 to 2.2 mm, the center notch frequency varies from 6 to 5 GHz. On the other hand, as Wc increases, the rejection-band region moves toward lower frequency with a narrower notch bandwidth. It is similar to increasing the capacitor value of a parallel LC circuit. The simulated return losses for various Wc
are shown in Figure 2.19. When Wc
varies from 2.6 to 5.6 mm, the center notch
frequency varies from 6.5 to 4.75 GHz. Thus, the notch frequency can be adjusted by selecting the suitable Wc and AR.0
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10
without notch AR = 1.8 AR = 2.0 AR = 2.2 0
Frequency (GHz)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
50 40 30 20 10
without notch AR = 1.8 AR = 2.0 AR = 2.2
Figure 2.18 Simulated return losses for the proposed antenna of various axial ratio (AR), the minor axis is 2.3 mm with a fixed value of Wc = 3.6 mm. Other geometric parameters are the same as given in Figure 2.9.
Frequency (GH z)
2 3 4 5 6 7 8 9 10 11
Return Loss (dB)
-50 -40 -30 -20 -10 0
without notch (simulated) W c = 2.6 mm (simulated) W c = 3.6 mm (simulated) W c = 5.6 mm (simulated) without notch (measured) W c = 3.6 mm (measured)
Figure 2.19 Simulated and measured return losses of the proposed antenna for various T-shaped stub width Wc with a fixed value of AR = 2. Other geometric parameters are the same as given in Figure 2.9.
In the case of Wc = 3.6 mm and AR = 2.2, the resistance and reactance of the proposed antenna at 5.5 GHz are 112Ω and 146Ω, respectively. This high input impedance causes impedance mismatching at the notch frequency, and the band-notched function covering the WLAN frequencies is obtained. Based on the analysis described above, the optimized design value of each physical dimension of the proposed antenna is determined and as shown in Figure 2.9. The simulated current distribution at 5.5 GHz is shown in Figure 2.20. It reveals that the currents mainly concentrate over the area of the two T-shaped stubs inside the elliptical slot cut in the radiation patch.
Figure 2.20 Simulated current distribution of the proposed antenna at 5.5 GHz.
2.2.3 Experiment Results
The measured return losses of the proposed antenna with and without the band-notched function are also shown in Figure 2.19. Without band-notched function, the antenna bandwidth (2:1 VSWR, or about 9.5 dB return loss) covers the range of 3.1-10.6 GHz assigned for the UWB application. Whereas with the band-notched function, the bandwidth is from 2.95 GHz to more than 11 GHz, and the antenna has a
rejection frequency band of 5 to 6 GHz, where the wireless LAN service is allocated, when inserting the equivalent parallel LC circuit into the radiation patch. An immediate sharp increase in VSWR is observed at the notch frequency.
Figure 2.21(a) to (d) show the measured radiation patterns at 3, 5.5, 6 and 9 GHz, respectively. It can be seen that the patterns of the proposed antenna at frequencies out of the notched band present omindirectional and stable radiation characteristics in the
xz-plane (H-plane) over the operating frequency range, which are similar to that of the
typical monopole antenna. The patterns measured at 5.5 GHz demonstrates that the antenna has much lower gains in the notched band than at other frequencies (3, 6 and 9 GHz), as shown in Figure 2.21(b).
-35 -30 -25 -20 -15 -10 -5 0 5
-35 -30 -25 -20 -15 -10 -5 0 5
Figure 2.21 Measured radiation patterns at (a) 3 GHz, (b) 5.5 GHz, (c) 6 GHz, and (d) 9 GHz. (solid line:
Eθ dashed line: Eψ)
The measured antenna gains from 3 to 10 GHz of the realized antenna are shown in Figure 2.22. The figure indicates that, the proposed antenna has good gain flatness except for in the notched band. The measured antenna gain variations are less than 4 dB throughout the desired UWB frequency band, and a sharp gain drop of about 10 dB occurs at 5.5 GHz.
The reduction in gain at the notch frequency is significantly greater than the reduction of power fed into the antenna caused by the return loss. This phenomenon can be investigated by examining the radiation efficiency. In Figure 2.22, the
simulated radiation efficiency, which excludes the impedance mismatching effect, at 5.5 GHz is only about 21%. It is because that most currents are trapped in a small region of the equivalent parallel LC circuit at this frequency, as shown in Figure 2.20, the resultant radiation fields cancel out, and thus the antenna does not radiate efficiently.
Figure 2.23 shows the photograph of the finished antenna.
Freq u en cy (G H z)
2 3 4 5 6 7 8 9 1 0 1 1
Gain (dBi)
-4 5 -4 0 -3 5 -3 0 -2 5 -2 0 -1 5 -1 0 -5 0 5
Radiation Efficiency (%)
0 1 0 2 0 3 0 4 0 5 0 6 0 7 0 8 0 9 0 1 0 0
S im u lated G ain M easu red G ain S im u lated R a d iatio n E fficien cy
Figure 2.22 Gains and radiation efficiency of the proposed antenna with band-notched function.
(a) (b)
Figure 2.23 Photograph of the proposed antenna. (a) Full band design and (b) with band-notch function design.
2.2.4 Summary
A compact microstrip-fed planar UWB antenna with the band-notched characteristic at around 5.5 GHz has been proposed and implemented. The total antenna size is 24 mm × 35 mm × 0.8 mm. Several design parameters have been investigated for the optimal design. By using two bevels on the upper side of the ground plane, the impedance matching in high frequency band can be improved. Moreover, adding two semicircle slots in the bottom side of the ground plane improve not only the input matching, but also the radiation characteristics at high frequencies. A pair of T-shaped stubs inside an elliptical slot, which is an equivalent parallel LC circuit, is realized to obtain the band-notched function. The center notch frequency and desired notch bandwidth are achieved by the properly designed equivalent capacitor and inductor values (i.e., Wc and AR of the elliptical slot). It is seen from the measured results that the proposed antenna has omnidirectional radiation patterns and a rather flat gain variation over the full UWB band expect for in the notched band. Therefore, the proposed antenna is suitable for the UWB communication applications and at the same time prevents interference with the WLAN systems.
Chapter 3
A Simple Monopole-like Printed Ultra-wideband Antenna with a Quasi-Transmission Line Section
In this chapter, we propose a simple and compact monopole-like printed ultra-wideband antenna. The antenna is composed of a monopole section and a quasi-transmission line section. The input signal from the feed line first passes through the line section then enters the monopole. The quasi-transmission line section provides different functions as the operating frequency changes. It serves not only as an impedance matching circuit but also a main radiator, which leads to the ultra-wideband performance of the antenna. The resonance mechanisms across the full band are described, followed by a thorough study of the antenna’s geometrical parameters. The experimental results show good agreement with the simulation. The measured 10-dB return loss bandwidth is 93.2% from 3.57 to 9.80 GHz, with a total antenna size of 20 × 40 mm2.
3.1 Antenna Configuration
Figure 3.1 (a) shows the geometry of the proposed antenna. It consists of a vertical monopole section and a short horizontal quasi-transmission line section. The quasi-transmission line section is formed by a parallel metal wire and the ground plane with a small gap of G. The metal wire has a length of Lt and a width of wt. The height and width of the monopole section are denoted as H and w, respectively, and the size of the ground plane is W × L. A 50 Ω microstrip line of 1.5 mm width is connected to the antenna as the feed line. The antenna is implemented on two sides of a FR4 substrate, whose dielectric constant is 4.4, loss tangent 0.02, and thickness 0.8 mm. Figure 3.1(b)
is the geometry of a conventional printed monopole antenna for comparison.
G
L
W Lt H
x y
z
Feed line w
wt
G
L
W Lt H
x y
z x
y
z
Feed line w
wt
(a)
H
L
x W y
z
Feed line H
L
x W y
z x
y
z
Feed line
(b)
Figure 3.1 Geometries of (a) the proposed antenna and (b) a conventional monopole antenna.
3.2 Resonances of the Antenna
The resonance phenomenon of an antenna depends on the antenna configuration.
For a wideband antenna, like a printed fat monopole antenna, it may not have only single but multiple resonances. In this section, the resonance mechanisms of the proposed antenna are described according to the simulated current distributions on the antenna structure. They can also be checked from the parameter study in Section IV.
The antenna is simulated using the Ansoft High Frequency Structure Simulator (HFSS), which is a commercial 3-D full-wave electromagnetic simulation software.
Figure 3.2 shows the simulated return loss of the proposed antenna with the monopole of H = 12.4 mm, w = 2 mm and the quasi-transmission line of Lt = 4 mm, wt
= 1.5 mm, and G = 0.6 mm. The ground size W × L equals 20 mm × 27 mm. It is seen from the figure that, a total impedance bandwidth, determined by a 10-dB return loss, is about 98.6 % from 3.33 to 9.80 GHz. There are four resonances, with resonant frequencies at 3.66, 4.83, 6.46, and 8.43 GHz, sustaining the full band.
1 3 5 7 9 11
Frequency (GHz) 30
20 10 0
Return Loss (dB)
1 3 5 7 9 11
Frequency (GHz) 30
20 10 0
Return Loss (dB)
Figure 3.2 Simulated return loss of the proposed antenna with H = 12.4 mm, w = 2 mm, Lt = 4 mm, wt = 1.5 mm, and G = 0.6 mm. The ground size W × L = 20 mm × 27 mm.
To understand the resonance mechanisms of the antenna, the current distributions at the four resonant frequencies are examined and shown in Figure 3.3 (a)-(d), respectively.
At the first resonant frequency (Figure 3.3 (a)), the current on the vertical monopole section is weak as compared to that on the quasi-transmission line section. However, as will be shown later, the quasi-transmission line section at this frequency behaves as a series inductor, thus not corresponding to the antenna radiation. Although not obvious in the figure, it is actually the currents on the two sides of the ground plane contributing to the antenna radiation. It means that the ground plane is the main radiator at the present resonant frequency. This can be doubly checked in Section 3.4 when one changes the ground size to see the variation of the resonant frequency.
At the second resonant frequency (Figure 3.3 (b)), the vertical monopole section has the strongest current as compared to other portions of the proposed antenna. And the current vanishes at the open end and becomes larger when moving toward the connection point of the monopole and the quasi-transmission line section, which is a current distribution similar to that on a typical quarter-wavelength monopole antenna (Figure 3.1 (b)). It is thus evident that at the second resonant frequency of 4.83 GHz, the monopole section is the main radiator of the proposed antenna and functions as a quarter-wavelength monopole antenna.
As for the third resonant frequency (6.46 GHz), as shown in Figure 3.3 (c), the current is mainly distributed on the quasi-transmission line section, surrounding the transmission line gap G. The current vanishes near the right end of the line section and is a maximum at the left end. This current distribution is like that of a quarter-wavelength open slit antenna. Thus, the quasi-transmission line section plays a role as a resonant slit antenna, which is the key contributor of the antenna radiation at the third resonant frequency.
Finally, for the fourth resonant frequency of 8.43 GHz, the proposed antenna has a current distribution as shown in Figure 3.3 (d). The vertical monopole section has a strong current with a maximum at the midpoint and nulls at both ends. It is clear that a half-wavelength resonance is formed on this vertical monopole section. The monopole section is again the main radiator of the proposed antenna at this resonant frequency, and behaves as a half-wavelength monopole antenna.
(c) (d) (a) (b)
(c) (d) (a) (b)
Figure 3.3 Simulated current distributions of the proposed antenna at (a) 3.66, (b) 4.83, (c) 6.46, and (d) 8.43 GHz.
3.3 Effect of the Quasi-Transmission Line Section
As shown in Figure 3.1, the quasi-transmission line section is the only geometrical difference of the proposed ultra-wideband antenna from a conventional narrow-banded monopole antenna. It is thus interesting to investigate its position on the antenna’s wideband performance.
Figure 3.4 compares the input impedances of the proposed antenna (solid lines) and the convention monopole antenna (dashed lines). It is seen that the impedance
Figure 3.4 compares the input impedances of the proposed antenna (solid lines) and the convention monopole antenna (dashed lines). It is seen that the impedance