Chapter 1 Introduction
1.3 Organization of This Thesis
The thesis is organized as follows. In chapter 2, we describe the MIMO-OFDM system model, explain the MIMO-OFDM concept and discuss the synchronization issues and the effects of synchronization errors. In chapter 3, we introduce the background and the physical layer concept of IEEE 802.11n. The proposed synchronization scheme is described in chapter 4. The synchronization scheme is simulated by the Matlab program and the results are shown in chapter 5. Finally, chapter 6 gives the conclusions and future work.
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Chapter 2
Fundamentals of MIMO-OFDM Systems
2.1 MIMO-OFDM Basics [7]
The basic principle of OFDM is to divide the high-rate data stream into many low rate streams that each is transmitted simultaneously over its own subcarrier orthogonal to all the others. Due to narrowband property, they experience mostly flat fading, which makes channel equalization very simple. In order to eliminate intersymbol interference (ISI) and intercarrier interference (ICI) as much as possible, it is a good idea to add a trailing portion of each symbol to the head of itself, which is called cyclic prefix extension as shown in Figure 2.1. The guard interval is chosen larger than the expected
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delay spread, such that multi-path components from one symbol will not interfere with its succeeding symbol. After a signal passes through the time-dispersive channel, orthogonality of its subcarrier components can be maintained by the introduction of cyclic prefix.
Figure 2.1 Guard interval and cyclic prefix structure of an OFDM symbol.
For the reasons mentioned above, OFDM is a leading modulation technique for wireless communications. Combining it with MIMO transmission systems increases the achievable throughput over wireless media significantly.
2.1.1 Baseband MIMO-OFDM Model
A MIMO OFDM system with NT transmitter and NR receiver antennas is considered as an NT × NR MIMO-OFDM system. Figure 2.2 shows the baseband structure of an NT × NR MIMO-OFDM system. The OFDM symbol transmitted by the pth transmit antenna is given by
( )
S( )
k e n Nn N
s N
k
N j nk p
p = ∑− ≤ ≤
=
0 1 1 ,
0
2π
(2.1) where Sp(k) denotes the transmitted data from the pth transmitter antenna, 1≦p≦NT,
Cyclic prefix
Useful symbol period Guard
interval
6
and N denotes the number of subcarriers. The received signal on the qth receiver antenna is
where hp,q(n) denotes the multi-path Rayleigh fading channel between the pth transmitter and the qth receiver antenna, and vq(n) denotes the complex additive white Gaussian noise with variance N0 at the qth receiver antenna. The received signal is demodulated with FFT as
( ) ( )
Figure 2.2 Block diagram of an NT×NR MIMO-OFDM system.
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2.1.2 Channel Estimation of MIMO OFDM Systems [38]
Transmitter diversity is an effective technique for combating fading in mobile wireless communications, especially when receiver diversity is expensive or impractical. For OFDM systems with transmitter diversity using space-time coding, two or more different signals are transmitted from different antennas simultaneously.
The received signal is the superposition of these signals, usually with equal power.
The transmitter antennas simultaneously transmit OFDM signals Sp(k). Hence, discrete Fourier transform (DFT) of the received signal at each receiver antenna is the superposition of distorted transmitted signals, which can be expressed as
∑=
The frequency response at the kth tone corresponding to the pth transmitter antenna can be expressed as
( ) ( )
, exp( 2 / )dispersion of the wireless channels.
According to [38], we can use the known long training sequences to find the temporal estimation of the channel parameters by minimizing the following MSE cost function.
Further details can be found in [38, 39]. We will use the channel parameter estimation approaches to decode the space-time block coded data.
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2.1.3 Space-Time Block Coding [40, 41]
Alamouti proposed a code scheme with full diversity that is originally designed for systems with two transmitter antennas and flat-fading channel [40]. Two consecutive symbols are processed in the transmitter in such a way that signals on the antennas are orthogonal. Therefore, they can be combined easily in the receiver. The scheme is further extended to four antennas [41]. Alamouti’s scheme is discussed as follows.
Table 2.1 The symbol placement of Alamouti’s scheme.
Antenna 0 Antenna 1
Time t S0 S1
Time t+T -S1* S0*
According to signal format listed in Table 2.1, the channel is assumed flat and constant across two consecutive symbols. That is,
( ) ( )
where T is symbol duration. The received signal can be expressed as
( )
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where r0 and r1 are received signals at time t and t+T, respectively, while v0 and v1 are complex random variables representing receiver noise and interference, respectively.
The combiner builds the following two combined signals.
*
Unfortunately, the flat fading channel is usually hard to achieve. OFDM systems separate the total frequency-selective fading channel into lots of sub-channels, which are often flat fading. Therefore, OFDM systems can achieve the condition. Thus, for each sub-channels, the channel condition matches the constraint placed by Alamouti.
In both WWiSE and TGnSync IEEE 802.11n proposals, the number of data streams is supported up to four. In order to achieve the optional 4×3 MIMO OFDM configuration, we extend the scheme to four antennas proposed in [41].
When the number of data streams is three and four, we use the following space-time block codes, H3 and H4, to encode the data. The decoders for H3 and H4 are derived in the Appendix of [41].
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2.2 Synchronization Issues of MIMO OFDM
OFDM is an efficient method of fast data transmission over frequency selective fading channels. Due to the division of the data stream into many sub-streams transmitted at a lower data rate on different sub-carriers and the application of a guard period preceding the data pulse, ISI (inter-symbol interference) is almost avoided. In most of the OFDM systems, the guard period is filled with the samples from the end of the symbol making the requirements for timing recovery easier.
Although OFDM is a spectrally efficient method of digital modulation, it has some serious drawbacks. The major one is that it is highly sensitive to synchronization errors.
In MIMO-OFDM systems, the effect of synchronization errors is also a serious problem. Therefore, MIMO-OFDM systems also require synchronization in both the
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time and frequency domains.
Synchronization of an OFDM signal requires detecting the beginning of a frame, finding the symbol boundary, the carrier frequency offset, and the sampling clock offset.
Here, we will discuss the effects of the carrier frequency offset, the symbol timing offset, and the sampling clock offset to an OFDM based system.
2.2.1 Effect of the Carrier Frequency Offset
Frequency offset is caused by inaccuracies and thermal changes of the oscillators used in the transmitter and receiver as well as by the terminal mobility (Doppler spread).
In the IEEE 802.11n the frequency offset f∆ in comparison to the subcarrier spacing of the OFDM signal is small.
Before discussing the effect of carrier frequency offset, we define the normalized frequency offset η as
spacing
f
∆f
η= (2.13)
s spacing
f = NT1
where fspacing is the subcarrier spacing of the OFDM system.
Generally, the fractional part of η causes attenuation and introduces ICI (Inter-Carrier-Interference), and the integral part of η introduces the effect that the received symbols would appear at incorrect output bins of the FFT demodulator. To prevent such cases, the frequency offset estimation and correction algorithm must be established.
The sampled signal with the sampling period Ts is [8]
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Thus, the received signal sample on the k-th subchannel is
( ) ( )
The first term shows the received Sˆ
( )
k amplitude degradation and phase shift due to the frequency offset. The second term I ′( )
k is the ICI caused by the frequency offset.If degradation D in SNR caused by a frequency offset that is small relative to the subcarrier spacing, then it can be approximated as [9]
0
It can be clearly seen that the effects introduced by frequency offset ∆f are twofold.
First, the frequency offset causes attenuation and phase rotation of the desired data signalS ′
( )
k . The second form of the influence of the frequency offset is the intercarrier interference expressed by the second term. It cannot be easily compensated by the set of the complex equalizer. These two effects vindicate that the compensation of the frequency offset is an important task.13
2.2.2 Effect of the Symbol Timing Offset
In symbol timing synchronization, inter channel interference (ICI) and inter symbol interference (ISI) occur when the FFT window wrongly covers an area that partially contains the symbol and cyclic prefix of the following symbol, as shown by case 3 in Figure 2.3.
Case 1 in Figure 2.3 shows the perfect FFT window position in AWGN channels [7]. In this case, the orthogonality of the sub-carriers is maintained, and there is no phase rotation in the frequency domain after the FFT block. However, it is not the case for a transmission in multi-path fading channels [10][11]. The symbol timing should be estimated in between the last delay path and the end of guard interval to avoid the ISI.
Thus, the ISI is perfectly removed only when the first and last delay paths are accurately estimated.
When the cyclic prefix is longer than the channel delay spread, there is a certain range in the prefix itself that is not affected by the previous symbol. As long as the FFT window starts within this range (Case 2 in Figure 2.3), the symbol timing error just results in a circular shift in the time domain and therefore in a phase rotation in the frequency domain after the FFT block. In this case, the orthogonality of the sub-carriers is maintained.
In Case 3, the FFT will cover partially not only the target samples belonging to symbol n, but also part of the cyclic prefix of the next symbol n+1. Thus part of the information is lost and the prefix of symbol n+1 will cause irrecoverable ISI. The effect of this unwanted ISI is that it will destroy the orthogonality between the subcarriers of current symbol. As a result, one can see spreading effects of the transmitted constellation points which can be modeled as additional noise.
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Symbol n Symbol n+1
Symbol n-1 Symbol n GI
Figure 2.3 Scenario of boundary misplacement of a received OFDM symbol.
For the reasons mentioned above, the FFT window must be selected within the appropriate range. Thus, an MIMO-OFDM system needs to do timing synchronization and avoids the effect of timing errors.
2.2.3 Effect of Sampling Clock Offset
Sampling clock errors introduce the clock phase error and the clock frequency error. Clock phase error effects are similar to symbol timing errors and can be treated in the same way. The constant fractional error can be compensated by a simple rotation or interpolation techniques. However, the time varying offset results in phase changes and ICI, because the sub-carriers are not orthogonal any more.
If we define the sampling clock frequency offset as
s
where Ts and Ts` are the transmitter sampling period and the receiver sampling period respectively. The received signal can be expressed as [12,13]
( )
n k15
is the noise caused by the clock frequency offset with variance [13]
[ ] 2 ( )2
) 3 , (
Var π β
β n k k
N ≈ (2.19)
The degradation grows at a rate proportional to the square of the product of the clock offset β and the subcarrier index k. The phase rotation increases with symbol index n. The clock offset β is small in most practical case, sinc(πβk)≈1. Thus, the effect of sampling clock offset can be approximated as an additional phase rotation. Under wireless communication conditions Nβ(n,k) can be neglected (βk << 1), but the phase rotation increases with symbol index and cannot be neglected.
Table 2.2 lists the parameters of several wireless communication standards about the effect of sampling clock offset. In DAB system, the number of symbols in a frame is fixed as 76 and the modulation scheme is DQPSK. Therefore, the sampling clock offset is not a serious problem in this system. However, when the frame size is not fixed or the number of subcarriers is large, we suggest that the sampling clock synchronization is needed, especially for 8K-mode DVB-T.
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Table 2.2 Parameter comparison among various OFDM standards.
Fs : the sampling frequency. Nsubcarrier : the number of subcarriers.
Fs N subcarrier Modulation
802.11n 20MHz 64
128
BQPSK QPSK 16QAM 64QAM
DAB 2.048MHz
2048 512 256 1024
DQPSK
DVB
9.14MHz 8MHz 6.8MHz
2048 8192
QPSK 16QAM 64QAM
802.16a 22.857MHz 2048
QPSK 16QAM 64QAM
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Chapter 3
Introduction to IEEE 802.11n
3.1 Overview of IEEE 802.11 Standard
The original version of the standard IEEE 802.11 [5] released in 1997 specifies two raw data rates of 1 and 2 megabits per second (Mbps) to be transmitted via infrared (IR) signals or in the Industrial Scientific Medical frequency band at 2.4 GHz. IR remains a part of the standard but has no actual implementations.
In the IEEE 802.11 family, the most popular techniques are those defined by the a, b, and g amendments to the original standard; security was originally included, and was later enhanced via the 802.11i amendment. Other standards in the family (c–f, h–j, n)
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are service enhancement and extensions, or corrections to previous specifications.
802.11b [15] was the first widely accepted wireless networking standard, followed by 802.11a [6] and 802.11g [16].
802.11a standard uses the 5 GHz band, while 802.11b and 802.11g standards use the unlicensed 2.4 GHz band. Operating in an unregulated frequency band, 802.11b and 802.11g equipments will result in interference from microwave ovens, cordless phones, and other appliances using the same 2.4 GHz band.
The IEEE 802.11 standard was adopted in 1997. Since then, several extensions to the standard have been developed, and more are emerging. The complete family of the current and emerging 802.11 standards are listed in Table 3.1.
3.2 History and Current Status of IEEE 802.11n
The 802.11n is the next generation extension of the physical layer. It is expected that 802.11n will support throughput rates (useful data rates) of over 100 Mbps. The standard is still in the earlier development phase. Among the proposed approaches to provide such high data rates are smart antenna technology, enhanced modulation, and increased channel bandwidth (using both 2.4 and 5GHz bands).
IEEE 802.11n standard process has three stages: the stage 1 is the preparation stage from Jan. 2002 to Sep. 2002; the stage 2 is the IEEE 802.11 High Throughput Study Group (HTSG) from Sep. 2002 to Sep. 2003; the stage 3 is the IEEE 802.11n Task Group from Sep. 2003 to Sep. 2005 (expected).
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Table 3.1 Summary of IEEE 802.11 standards.
Standard Description Status
802.11 The original 1 Mbps and 2 Mbps 2.4 GHz RF and IR standard
Released in 1997
802.11a 54Mbps,5GHz Ratified in1999, shipping products in 2001
802.11b Enhancements to 802.11 to support 5.5 and 11 Mbps, 2.4GHz
Ratified in1999
802.11c Access point bridging Completed
802.11d Regulatory extensions Completed
802.11e Quality of Service Completed 802.11f Inter Access Point roaming Completed
802.11g 54Mbps,2.4GHz
(backwards compatible with b)
Completed in 2003
802.11h Transmit power control, Dynamic frequency selection
Completed
802.11i Enhanced security Completed
802.11j Japanese regulatory extensions Completed 802.11k Radio resource measurement Ongoing
802.11m Maintenance Ongoing
802.11n Higher throughput improvements (100+ Mbps)
Expected completion in 2006-2007
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In January 2004, IEEE announced that it had formed a new 802.11 Task Group (TGn) to develop a new amendment to the 802.11 standard for local-area wireless networks. The real data throughput rate will be at least 100 Mbps (which may require an even higher raw data rate at the physical layer), and should be up to 4–5 times faster than 802.11a or 802.11g, and perhaps 20 times faster than 802.11b. It is projected that 802.11n will also offer a better operating distance than current networks. There are two competing variants of the 802.11n standard: WWiSE (backed by companies including Broadcom) and TGn Sync (backed by Intel, Philips and others). The standardization process is expected to be completed by the end of 2006.
802.11n builds upon previous 802.11 standards by adding MIMO (multiple-input multiple-output). The additional transmitter and receiver antennas allow for increased data throughput through spatial multiplexing and increased range by exploiting the spatial diversity, perhaps through coding schemes like Alamouti coding.
TGnSync [4] and WWiSE [3] are the two major proposals for IEEE 802.11n Task Group.Both proposals emphasize compatibility with existing installed base, building on experience with interoperability in 802.11a/g and previous 802.11 amendments.
Therefore, in the next section, we review the physical layer of wireless LAN 802.11a systems based on OFDM technology.
3.3 Physical Layer of IEEE 802.11a
The 802.11a standard, introduced at the same time as 802.11b, is intended for the 5 GHz license-free UNII band and provides data rates up to 54 Mbps. The 5 GHz band has an advantage of large bandwidth allocated for the unlicensed operations. There are
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455 MHz available (5.15 – 5.35 MHz and 5.470 – 5.725 MHz) for use by WLAN systems in Europe. This allows 19 non-overlapping channels in the 5 GHz band versus 3 non-overlapping channels in the 2.4 GHz band.
802.11a is based on Orthogonal Frequency Division Multiplexing (OFDM) modulation, and allows to achieve higher data rates within about the same channel bandwidth as 802.11b.
The IEEE 802.11a operates at a sampling rate of 20M Hz and uses 64-point FFT.
The OFDM frame duration is 80 samples where 64 is for data while 16 is cyclic prefix.
Since the symbol rate on each subcarrier is slower than the original data rate, the OFDM technique is particularly efficient in time dispersive environments. The 802.11a OFDM signal consists of 52 carriers. Data are sent on 48 carriers simultaneously, with 4 carriers used as pilots to aid in channel estimation at the receiver. The main system parameters of IEEE 802.11a Wireless LAN standard are listed in Table 3.2.
Using different modulation QAM combined with punctures of convolutional encoder, variable data rate can be achieved with a minimum 6Mbps and maximum 54 Mbps. Table 3.3 shows supported data rates. Various data rates are provided by changing the redundancy in the error correction coding and by changing modulation scheme.
Adopted in 2003, the 802.11g extension enables 54 Mbps data rates, the same data rate as provided by the 802.11a standard, but now in the 2.4 GHz band. This is achieved by using the same data rates and modulation formats as used in the 802.11a standard.
Additionally, the 802.11g standard is backward compatible with the 802.11b standard, i.e. the 802.11b modulation formats and data rates are supported.
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Table 3.2 The parameters of IEEE 802.11a system.
Sampling rate 20MHz
Number of FFT points 64
Number of data subcarriers 48 Number of pilot subcarriers 4
Subcarrier spacing 0.3125 MHz (=20MHz/64)
OFDM symbol period 4μs (80 samples)
Cyclic prefix period 0.8μs (16 samples)
FFT symbol period 6.2μs (64 samples)
Modulation scheme BPSK,QPSK,16QAM,64QAM
Coding 1/2 convolutional, constraint length 7,
optional puncturing
Data rate 6, 9, 12, 18, 24, 36, 48, 54 Mbps
Short training sequence duration 8μs Long training sequence duration 8μs Long training symbol GI duration 1.6μs
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Table 3.3 Data rates and rate-dependent parameters of 802.11a standard.
In IEEE 802.11a standard, a frame is composed of three fields. Figure 3.1 shows the frame format of IEEE 802.11a. The preamble field is modulated with QPSK, and no interleaving and scrambling. Figure 3.2 shows the preamble format and the possible arrangement of the synchronization and channel estimation for the receiver.
In the preamble field, the preambles are composed of ten repeated short symbols and two repeated long symbols. Both the total durations of short training symbols and long training symbols are 8 µs. The SIGNAL field is modulated with BPSK, interleaving, but no scrambling. Because the SIGNAL contains the most important information of the packet, every synchronization and channel estimation must be done before decoding of the SIGNAL. In the DATA field, modulation and coding rate depend on the information carried by SIGNAL field, interleaving and scrambling are executed.
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Figure 3.1 The frame format of IEEE 802.11a standard.
Short Training Symbol Field
•Frame Timing Sync.
•Coarse CFO Sync.
•Symbol Timing Sync.
Long Training Symbol Field
•Fine CFO Sync.
•Channel Estimation.
Figure 3.2 The training symbol structure of 802.11a.
Pad One OFDM Symbol
DATA
Variable Number of OFDM Sy mbols
Pad One OFDM Symbol
DATA
Variable Number of OFDM Sy mbols PLCP Preamble
12 Symbols
SIG NAL One OFDM Symbol
DATA
Variable Number of OFDM Sy mbols
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3.4 Preamble Format of IEEE 802.11n
For the purpose of compatibility with the 802.11 legacy devices, the legacy part of preamble format in WWiSE and TGnSync proposals is the same as in 802.11a. If the legacy preambles are transmitted from multiple antennas, the mapping of this single spatial stream to multiple antennas has to be done such that beamforming in far-field is
For the purpose of compatibility with the 802.11 legacy devices, the legacy part of preamble format in WWiSE and TGnSync proposals is the same as in 802.11a. If the legacy preambles are transmitted from multiple antennas, the mapping of this single spatial stream to multiple antennas has to be done such that beamforming in far-field is