All pads size on RFIC were 100 μm×100 μm. Set the size of pads on PCB were 200 μm×200 μm, and the distance from RFIC pad to PCB pad was 500 μm.
The diameter of bond wire was 0.7mil (0.01778 μm); the specification was defined by IST. The simulation setup of wire bonding was designed by AnsoftTM HFSS. In the chip package, there were two types of via hole, one via holes were next on the chip (Fig. 3.10 (a)), and the other were bored of being in the bottom of the chip (Fig. 3.10 (b)).
(a)
(b)
Fig. 3.10(a) via holes was next on the chip in the chip package, and (b) via holes was in the bottom of the chip.
While via holes were in the bottom of the chip package with a better frequency response that was shown in Fig. 3.11 (b), but then more easily lead to mount chip failure by its uneven surface. On the other hand, Figure 3.11 (a) illustrated the via-holes were next to the chip on a package more easily mount the circuit board, but its relatively poor frequency response.
(a)
(b)
package. The first stage of the matched network design was made a L-section to match the output impedance of wire-bonding to 50Ω, and the second stage of the matched network design another L-section to match the input impedance of the external driving amplifier to 50Ω.
Fig. 3.12 The matched concept of a CMOS transceiver (RFIC) package for the transmitter port matching.
In the first stage, the return loss of the transmission port of the RFIC was measured from the probing test by cascadeTM M150 and the S-parameters of the bond wire were obtained from the linear simulation by AnsoftTM HFSS. The L-section matched network design for the output impedance of wire-bonding was made by AgilentTM ADS2006, and then the network would be matched to 50Ω. Figure 3.13 displayed the structure and result of the simulation. It was combined with the L-section matched network and SP files (S-parameters) of the bond wire and TX output of RFIC. The purpose of this simulation was built the physical layout for the L-section matched network and matched the impedance of a signal path to 50Ω. Therefore, the smith chart would be applied to the matched network design, the lengths and widths of the L-section were adjusted to leading the curve of S11 (the return loss of the path) to 50Ω at the operating frequency (10.525GHz). In the second stage, the S-parameters of the external driving amplifier were obtained from the HittiteTM datasheet. The L-section
matched network design for the input impedance of the external driving amplifier was also made by AgilentTM ADS2006. Figure 3.14 displayed the structure and result of the simulation.
Fig.3.13 The structure and result of the output impedance matching were illustrated for the wire-bonding.
Fig.3.14 The structure and result of the input impedance matching were illustrated for the external driving amplifier.
design would be used to compensate the mismatched impedance. First the prototype model was made from ideal components of AgilentTM ADS2006, the schematic and the response of π-pad design was illustrated in Fig. 3.15.
Fig. 3.15 The schematic and response displayed of the ideal π-pad design.
Nevertheless, the physical transmission line would be considered in the π-pad design. The layout of transmission lines designed according to the size of physical resistors of π-pad and the operating frequency. The layout, schematic, and response of π-pad design were shown in Fig.3.16, but the response result was poor in operating frequency. The resistance values of π-pad would be recalculated with pad effect. After re-calculation, the resistance of 18Ω was changed to 30Ω and 300Ω was changed to 540Ω, then the better response result was obtained and displayed in Fig. 3.17.
Fig.3.16 The layout, schematic, and response of π-pad design with pad effect of the transmission line.
Fig. 3.17 Better response results after recalculated the resistors of π-pad design.
After the matching process, the package job would be started, the pin assignment of RFIC should be defined (Fig. 3.18), then the RF pads would be matched to the proper impedance that was like the above matching process and DC pads could be connected directly with the ESD circuits. Figure 3.19 demonstrated the physical layout pads definition of the CMOS transceiver (RFIC).
Fig. 3.18 Pin assignment defined for the CMOS transceiver (RFIC).
Fig 3.19 Physical layout defined for the CMOS transceiver (RFIC).
Additionally, the CMOS RF transceiver IC was wire-bonded (golden wires of 0.7 mils in diameter) directly on the RO4003TM substrate separately for evaluation purpose. As shown in Fig. 3.20 the CMOS chip was silver glued on the substrate before bonding and before the glow top epoxy process was performed. The wire inductance were compensated for by the additional matching circuits on the board to ensure the RF signal integrity or was used as part of the RF choke for DC bias. The matching circuit was made from well-known transmission line L-matching circuits with a single stub. A
parallel-coupled filter was added on the board between the L-matching circuit and the antenna arrays.
Fig. 3.20 Photograph of FMCW CMOS die attached to board with aluminum bonding wires.
CHAPTER 4
Dual Leaky-wave Antenna Arrays Structure Design with High Isolation and High Gain
The traditional csc2θ type antennas were designed by technology of the aperture or the reflector antenna [20], [41] and applied both airborne and surveillance radar for military utilization. However, those operations of military possess the greater power consumption, large scaling volume, more complicated structure, and farther distance measurement. Especially, the characteristics of csc2θ type antennas were the special pattern compensates for the space loss of 1/R4 that can perform suitably the range measurement of multiple-lane for TMS application. Furthermore, the short range detection and low power depletion were required for this application, and then the FMCW detected method was considered to achieve the new system.
4.1 The Design Method of the Leaky-wave Antenna Arrays
The csc2θ pattern was our special design that increases the length of the antenna, which the characteristics of the csc2θ pattern became increasingly apparent. The characteristic curve was showing a range between the two dashed-lines in Fig. 4 of [42]. The theory of a csc2θ pattern of leaky-wave antenna has already been derived and proved in [42]. For the article how to use one element of the leaky-wave antenna to design a csc2θ pattern for the method illustrated, the variable length of the antenna metal (Fig. 4.1) adjust gradually to prove that the csc2θ pattern of the design concept by the numerical solution. The return loss of one element of the leaky-wave antenna at the operating frequency
was shown in Fig. 4.2.
Fig. 4.1 Tuning the metal length of one element of the leaky-wave antenna array.
9 10 11 12
Frequency (GHz) LWA_RO4003
-30 -20 -10 0
Return Loss (dB)
11.325 GHz -10.006 dB 9.7509 GHz
-10.168 dB
DB(|S[1,1]|)
LWA_78_150_20_1645_matching
Fig.4.2 Return loss of one element of the leaky-wave antenna.
4.2 The Integration and Realization of Leaky-wave Antenna Arrays First design a pair differential port to feed one element of the leaky-wave antenna array. Figure 4.3 shown the structure (balun) of a differential input of one element of antenna array and achieved a phase difference of 180° between those ports. Secondly, the differential feeding network has been realized and then the equal power divider is designed to combine each element of the leaky-wave antenna array (Fig. 4.4). After feeding design, the full wave simulation of the whole feeding network of the leaky wave array was executed by IE3DTM, and Fig. 4.5 shows the structure for the feeding network with an 8-element array.
Fig. 4.3 Feeding ports of one element of the leaky-wave antenna array
Fig. 4.4 Feeding network and power divider of the leaky-wave antenna array
Fig. 4.5 Whole feeding network of the 8-element leaky-wave antenna array
The proposed antenna was designed to operate in the first high-order leaky mode (EH1) of a microstrip excited differentially at the X-band [43], [44]. And the derivation and explanation of the leaky-wave antenna theories are found in [43]-[48]. The basic structure of a leaky-wave antenna comprises a feeding network and a substance of antenna. Additionally, the leaky-wave antenna design is used as the differential feeding structure to excite the leaky mode. As shown in Fig. 4.6, this antenna array may consist of a set of equal power dividers of a feeding network [43]. And eight matching baluns for the differential input of each element of antenna [44], [46], and microstrip leaky-wave antennas of length (L) 150.0mm, width (W) 7.80mm, spacing (S) 6.40mm, and thickness (h) 0.508 mm in the designed array. The designed antennas are fabricated on RO4003TM substrates that have a relative dielectric constant (εr) of 3.5. The baluns were realized by a microstrip circuit with a phase difference of 180° between two differential ports and the equal power dividers were designed to combine all elements of the leaky-wave antenna array. Figure 4.7 shows the results of simulation of the feeding network with an 8-element
Fig. 4.6 The Leaky-wave antenna array, including (a) a set of equally power dividers of feeding network, (b) 8 matching baluns for the differential input of each element of antenna, and (c) Microstrip leaky-wave antennas with length of L, width of W, and spacing of S on the top side of the substrate.
Fig. 4.7 Performance of the feeding network for the 8-element array
According to depending on the relationship between the propagation constants of the guided mode, the higher-order modes of the microstrip line can be divided into the following four frequency regions [46]:
1) β >βS , α = 0, bound mode region;
2) βS>β >k0 , small α, surface wave leakage region;
3) k0 >β, smallα, surface wave and space wave leakage region;
4) k0 >β, largeα, cutoff region.
with β is the phase constant and α is the attenuation constant of higher-order mode of the microstrip line, βS is the propagation constant of the surface wave mode of the surrounding structure, and k0 is the wave number in the air. The original concept of this proposed antenna array is from the design of [43].
Moreover, the dispersion characteristics of the leaky-wave antenna between the operating frequencies derive an inequality as β < ko, with small α. β is the phase constant and α is the attenuation constant of the microstrip line higher-order mode, and ko is the wavenumber in the air. In [46], the inequality represented the higher-order modes of microstrip line can be entered the surface wave and space wave leakage region.
A simple procedure of the leaky-mode antenna array was designed according to using one of two approaches, namely, E-plane and H-plane techniques. In an E-plane antenna array, the number of antenna elements controls beam width of the E-plane pattern, and coarsely tunes gain. Table 4.1 shows the beam width and gain of an E-plan antenna. In an H-plane antenna array, the length of each antenna element regulates the beam width of the H-plane pattern, and fine tunes gain. Table 4.2 shows the beam width and gain of an H-plan antenna.
Table 4.1: E-Plane Beam-width and Gain vs. the Number of Antenna Elements
Table 4.2: H-Plane Beam-width and Gain vs. the Length of one Element Length of one element (mm) 50 100 150 200 300 Bandwidth of H plane (°) 30 22 20 19 19
Gain (dB) 7.5 10.8 11.8 12.0 12.1
Preceding paragraphs have already pointed out the E-plane and H-plane in our design, so the relationship was redefined between the E and H planes for the leaky-wave antenna was shown as Fig. 4.6 by the coordinate of Fig. 4.8.
Fig. 4.8 The relationship between the E and H planes for the leaky-wave antenna
To satisfy system requirements, the number of antenna elements was set to 8 and antenna length was set to 15 cm based on data (Tables 4.1 and 4.2). To validate the array design, full-wave electromagnetic simulations were performed using the commercial software ZelandTM IE3D. The optimal design of one element of the leaky-wave antenna followed the data of Table 4.2. Simulation results indicate that the efficiency and antenna gain of the antenna array at 10.5GHz were 80% and 18.4dB, respectively.
4.3 The Measurement Result of Leaky-wave Antenna Arrays
The antenna pattern measurement of the leaky-wave antenna array was performed in the far-field antenna laboratory (is belonged to prof. Yu-De Lin) of department of Communication Engineering, National Chio-Tung University.
The test frequencies of the antenna arrays pattern measurement are 9.7GHz,
10.4GHz, 10.5GHz, 10.6GHz, and 11.0GHz. The antenna pattern measurement includes E-plane Co-polarization measurement, E-plane Cross-polarization measurement, and H-plane Co-polarization measurement.
The leaky-wave antenna array was realized by the printed circuit boards (PCB) process. Figure 4.9 displayed the photograph of the leaky-wave antenna array.
Fig. 4.9 The photo of the leaky-wave antenna array
The co-polarization represented that the observed signature when the transmitted and received polarizations are the same. The setup of the E-plane co-polarization measurement was shown in Fig. 4.10. The right side of Fig. 4.10 was a measured antenna array and the left side of Fig. 4.10 was a standard horn.
The standard horn was a transmitted antenna and the leaky-wave antenna array was received antenna. The polarizations of two antennas are the same. The
Fig. 4.10 Setup of pattern measurement for E-plane co-polarization
Fig. 4.11 The pattern of E-plane co-polarization of the leaky-wave antenna array
The cross-polarization represented that the observed signature when the transmitted and received polarizations are orthogonal. The setup of the E-plane cross-polarization measurement was shown in Fig. 4.12. The antenna pattern measurement of the E-plane cross-polarization was shown in Fig.4.13. The E-plane pattern was also an azimuth direction pattern.
Fig. 4.12 Setup of pattern measurement for E-plane cross-polarization
Fig. 4.13 The pattern of E-plane cross-polarization of the leaky-wave antenna array
The setup of the H-plane co-polarization measurement was shown in Fig.
4.14. The H-plane pattern was also an elevation direction pattern. The antenna pattern measurement of the H-plane co-polarization was shown in Fig. 4.15.
Fig. 4.14 Setup of pattern measurement for H-plane co-polarization
Fig. 4.15 The pattern of H-plane co-polarization of the leaky-wave antenna array
Since the pattern energy of H-plane cross-polarization of the leaky-wave antenna array was very low,which resulted in the measurement was omitted.
After the antenna pattern measurement, other important measurements are return loss and coupling efficiency measurement for the 8 stubs leaky-wave antenna array.
There are two leaky-wave antenna arrays in the measurement. The antenna arrays were labeled as A and B. The isolation measurement was adjusted the spacing of the two individual leaky-wave antenna arrays from the most left metal stripe of the antenna array B to the most right metal stripe of the antenna A (Fig. 4.16).
Fig. 4.16 The isolation measurement was adjusted the spacing of the two individual leaky-wave antenna arrays.
Fig.4.17 and Fig.4.18 represented the return loss measurement (from 1GHz to 20 GHz) for Antenna A and B. Since our system used the two antennas to isolate the coupling influence, we tuned the space between two antennas and measured the insertion loss (S21) for two antennas. Figs. 4.19-4.23 represented the coupling measurement for antenna arrays A & B Spacing from 0cm to 8cm.
Fig.4.17 Return loss measurement for the antenna array A
Fig.4.18 Return loss measurement for the antenna array B
Fig.4.19 Coupling measurement for antenna arrays A & B Spacing 0 cm
Fig.4.20 Coupling measurement for antenna arrays A & B Spacing 2 cm
Fig.4.21 Coupling measurement for antenna arrays A & B Spacing 4 cm
Fig.4.23 Coupling measurement for antenna arrays A & B Spacing 8 cm
It summarized the above measurements that the conclusions were derived the following description: the space of two antennas which increase over 4 cm, the insertion loss (S21) will be dropped to –48dB. And then the insertion loss (S21) of the space of two antennas is also represented the coupling effect of two antennas. Hence, the space of the leaky-wave antenna array is wider than one wavelength, and then the coupling effect can be neglected.
More significantly, the electromagnetic coupling of the two antenna arrays must be measured before combining the entire sensor system. Figure 4.24 plots the measured isolation of the two antenna arrays separated by only 5.0mm, revealing a coupling of less than 42dB. Table 4.3 shows the measured coupling versus the spacing between two antenna arrays, revealing that the coupling is insensitive to the spacing. Significantly, an attainable isolation value for a good circulator in the X-band is around 35dB, which is approximately 10dB below that obtained by the proposed two-antenna array approach. Figures 4.25 and 4.26 show the measured cut-plane on the main beam at 56° from the E-plane (yz plane) and the H-plane (xz plane) radiation patterns of the leaky-wave antenna array at 10.5GHz. The measurements in Fig. 4.24 demonstrate that the half power is about 15 dB and is bounded between -6.5° and +6.5°. Hence, the 3dB
beam-width of the antenna array was 13° in the E-plane, and that of the main-beam with a gain of 18.5dB was 56° from the broad side of the array in the H-plane. The first lobes of the E-plane radiation pattern were equal to 3.2 dB, because of the original design of the leaky-wave antenna. Since the path lengths of the differential feeding structure of the leaky-wave antenna were not equal, the side-lobes were not symmetrical. Nevertheless, the main direction of the beam is still 0° in the E-plane. In the paper, it applied the elevation pattern (H-plane) to achieve the range measurement. The azimuth resolution (E-plane) haven’t discussed in the dissertation. The radiation angle of the elevation pattern has no relation with range resolution. However, the radiation angle effects directly the echo power distribution and the signal-to-noise rate (SNR) of range measurement.
The FMCW front-end, which includes the CMOS transceiver and antenna arrays, was designed and experimentally characterized. The next section presents a practical example of obtaining the vehicle occupancy in TMS to indicate the capability of the proposed FMCW sensor system.
-80 -70 -60 -50 -40 -30 -20 -10 0
S21(dB) Antenna Array Antenna Array
S
Port1 Port2
TABLE 4.3 Coupling of Two Antenna Arrays in Different Spacing
Fig. 4.25 Measured E-plane radiation pattern of the cut-plane on the main beam at 56° of the 8-element antenna array.
-180 -120 -60 0 60 120 180
Fig. 4.26 Measured H-plane radiation pattern of the eight-element antenna array.
The csc2θ-type antenna pattern in system applications, originated with the surveillance radar with a fan-shaped beam. The azimuth (E-plane) beam angle is small, and the elevation (H-plane) beam angle is large [20]. The main lobe pattern of the H-Plane of the leaky-wave antenna is always at an oblique angle.
The oblique angle is then increased as the length of the antenna array. When the oblique angle was adjusted to a suitable value, the conditional cosecant-squared antenna pattern was realized. The characteristic curve of the cosecant-squared antenna pattern ranges between the two dashed-lines in Fig. 4 of another investigation [42]. The theory of the csc2θ pattern of a leaky-wave antenna has also been derived and proved elsewhere [42]. In the investigation, one element of the leaky-wave antenna is used to design a csc2θ pattern associated with the proposed method that the length of the antenna metal must be varied gradually to prove the design concept with reference to the numerical solution. When the lengths of the metallic antenna are larger than 80.0mm, the csc2θ patterns are obtained (Fig. 4.27).
-90 -60 -30 0 30 60 90
Elevation Angle θ (deg) -15
Fig. 4.27 Patterns of elevation angles vary with the stripe length of antenna from 25.0 to 150.0mm.
the antenna array is shorter, then the radiated energy cannot be immediately
the antenna array is shorter, then the radiated energy cannot be immediately