The simulated radiation patterns of the proposed MLWA is shown in Figure 27. The original MLWA pattern of simulation is shown in Figure 28.
The simulated results illustrate that the back lobes have been suppressed successfully, and the back lobes of the proposed LKW have been suppressed by about 10 dB at the scanning band from 4.25GHz to 4.8GHz compared to the original MLWA. The measurement radiation pattern of the proposed MLWA is shown in Figure 29. The gain difference between the main beam and back lobe is 10 dB. The back lobes of the proposed MLWA is suppressed indeed in experimental results.
It is noticeable to speak of the scanning angle of the main beams about the proposed MLWA and original MLWA. In simulated radiation pattern, the main beams of the proposed MLWA are slightly slanted to broadside direction compared to the original one. It is mainly due to the monopole of the proposed MLWA. Therefore the scanning region of the proposed MLWA is wider than the original MLWA. In simulated results, the scanning angle of main beam of the proposed MLWA is about 44°, and the original MLWA is about 20°. There is about 24° difference between the proposed MLWA and original MLWA. The measurement scanning region is shown in Figure 29.
Figure 28: The simulated radiation patterns of the proposed MLWA
Figure 29: The simulated radiation patterns of the conventional MLWA
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Figure 30: The measurement radiation patterns of the proposed MLWA
3.4 Conclusion
Here, a novel Method of Beam-Steering Range Greatly for Leaky-Wave Antenna is introduced. We can design the open terminal of original MLKW to improve the performance. The monopole is used in this experiment.
This method can effectively improve the performance from comparing the difference between Figure 27 and Figure 28. It not only decreases the back lobes, but increases the scanning region. After we design the proposed MLKW, we have to match the input matching. We design the slot under the feeding line of the proposed MLKW appropriately. Figure 30 shows the comparison about matching and without matching. Finally, we have completed the proposed antenna.
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Figure 31: The comparison S-parameter of matching and without matching
Chapter4
Design Of MIMO Antenna For Application In 2.4GHz
Future wireless communication systems should be capable of accom-modating higher data rates than the current systems owing to the advent of various multimedia services. The use of multielement antennas, such as multiple-input multiple-output (MIMO) antenna systems, is one of the effective ways for improving reliability and increasing the channel capacity.
However, it is very difficult to integrate multiple antennas closely in a small and compact mobile handset while maintaining good isolation between an-tenna elements since the anan-tennas couple strongly to each other and to the ground plane by sharing the surface currents on them. For a M×N MIMO communication system, the data throughput can be pushed up to K times, K=min(M,N) that of a single-input single-output (SISO) system, as long as the communication channels linked between the transmitter and the receiver are uncorrelated [29–31]. The correlation between the chan-nels depends not only on the propagation environment, e.g., multipath effect due to the reflection and diffraction of outdoor buildings or indoor partitions, and also on the coupling between the N or M antennas. High antenna coupling (or low isolation) would introduce signal leakage from one antenna to another, thus increasing the signal correlation between the channels. It will also decrease the antenna radiation efficiency due to the loss of the power dissipated in the coupled antenna port. The signal cor-relation between two receiver antennas can be reduced by increasing the antenna spacing. However, this spacing is usually limited, especially for a mobile terminal which has very restricted volume for the antennas. The other way to diminish the correlation is using multiple antennas with dif-ferent radiation patterns. It is better to have the patterns complementary to each other in space, so as to receive multipath signals from various directions.
For a long time, many papers have been focused on diminishing the coupling of antennas. In[32], the relation of the isolation and the
arrange-37
ment of two nearby antennas with different operating bands in a cellular handset were studied. Itoh and the co-workers used the defected ground structure (DGS) to increase the port isolation of polarized and dual-frequency patch antennas[11]. The above tackled the isolation problems of antennas operating at different frequencies. For decoupling two nearby antennas with the same frequency, many efforts have been done by us-ing the electromagnetic band gap (EBG) structures. Mushroom-like EBG structures are the ones that are usually inserted between patch antennas to prevent the propagation of surface waves for higher isolation and better radiation patterns[33–35]. These EBG structures provide conspicuous de-coupling effect, but suffer from complicated structures and large structure area. Possible loss may also be induced in the resonant EBG structures.
To reduce the coupling between two planar inverted F antennas (PIFAs), Diallo[12, 36–38] used a suspended metal strip linking the two antennas to cancel the reactive coupling between antennas. This neutralization tech-nique has been also extended to patch antennas by [39] and [40]. In[41], a decoupling circuit network was realized for two-element array by using external transmission lines. Although good isolation was achieved, only weak coupled antennas were tackled. The all-transmission-lines configura-tion also made the circuit bulky. A similar approach of connecting circuits between elements has also been used to improve the impedance matching of a phasedarray antenna over wide scan angles [42] since mutual coupling may vary the input impedance in different scanning angle. The mutual coupling of two closely packed antennas was reduced by etching slots on the ground plane [10]. The fish-bone like slots formed equivalent induc-tors and capaciinduc-tors on the ground plane, which prevented the flowing of the coupling ground current between the antennas. A large ground plane size was needed for sufficient isolation. In this paper, the MIMO antenna will be composed by quadrature hybrid and monopole. We will design the quadrature hybrid in 2.4GHz. It will have the important property of high isolation. On the other hand, it has low relation between signal port. This
system.
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4.1Basic Theory of Quadrature Hybrid
Quadrature hybrid are 3dB directional couplers with a 90° phase dif-ference in the outputs of the trough and coupled arms. This type of hybrid is often made in microstrip or stripline form as shown in Figure 31. It is also known as a line hybrid. The basic operation of the branch-line coupler is as follows. With all ports matched, power entering port 1 is evenly divided between ports 2 and 3, with a 90° phase shift between these outputs. No power is coupled to port 4 (the isolated port). We will use this isolated property to apply on the MIMO antenna. The branch-line hybrid has a high degree of symmetry, as any port can be used as the input port. The output ports will always be on the opposite side of the junction from the input port, and the isolated port will be the remaining port on the same side as the input port. The [S] matrix of a reciprocal four-port network matched at all ports has the following form:
S =
If the network is lossless, we can get equation from the unitarity, or energy conservation. Let us consider the multiplication of row 1 and row 2, and the multiplication of row 4 and row 3:
S13∗ S12+ S14∗ S23 = 0 (3a) S14∗ S13+ S24∗ S23 = 0 (3b) Now multiply (7a) by S24∗ and (7b) by S13∗ , and subtract to obtain
S14∗ (|S13|2 − |S24|2) = 0 (4) Similarly, the multiplication of row 1 and row 3, and the multiplication of row 4 and row 2, gives
S12∗ S23+ S14∗ S34 = 0 (5a) S14∗ S12+ S34∗ S23 = 0 (5b) Now multiply (9a) by S12 and (9b) by S34, and subtract to obtain
S23(|S12|2 − |S34|2) = 0 (6) One way for (8) and (10) to be satisfied is if S14 = S23 = 0, which results in a directional coupler. Then the self-products of the rows of the unitary [S] matrix of (6) yield the following equations:
|S12|2 +|S13|2 = 1, (7a)
Further simplification can be made by choosing the phase references on three of the four ports. Thus, we choose S12 = S34 = α, S13 = βe(jθ), and S24 = βe(jϕ), where α and β are real, and θ and ϕ are phase constants to be determined (one of which we are still free to choose). The dot product of rows 2 and 3 gives
S12∗ S13+ S24∗ S34 = 0, (8) which yields a relation between the remaining phase constants as
θ + ϕ = π± 2nπ. (9)
If we ignore integer multiples of 2π, there are two particular choices that commonly occur in practice:
(1)The Symmetrical Coupler: θ = ϕ = π/2. The phases of the terms having amplitude β are chosen equal. Then the scattering matrix has the following form:
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S =
(2) The Antisymmetrical Coupler: θ = 0, ϕ = π. The phases of the terms having amplitude β are chosen to be 180° apart. Then the scattering matrix has the following form:
Note that the two couplers differ only in the choice of reference planes.
Also, the amplitudes α and β are not independent, as (11a) requires that:
α2 + β2 = 1 (12)
We can find the S-parameter matrx from (13) and (14), and four ports are matching, the power supplied to port 1 is coupled to port 3 (the coupled port) with the coupling factor |S13|2 = β2, while the remainder of the input power is delivered to port 2 (the through port) with the coefficient
|S12|2 = α2 = 1−β2. In an ideal directional coupler, no power is delivered to ports 4 (the isolated port). Hybrid couplers are special cases of directional couplers, where the coupling factor is 3 dB, which implies that α = β = 1/√
2. There are two types of hybrids:
(1)The quadrature hybrid has a 90°phase shift between ports 2 and 3 (θ = ϕ = π/2) when fed at port 1, and is an example of a symmetrical coupler. Its [S] matrix has the following form
S =
(2)The magic-T hybrid or rat-race hybrid has a 180°phase difference be-tween ports 2 and 3 when fed at port 4, and is an example of an antisymmetrical coupler. Its [S] matrix has the following form:
S = 1
√2
0 1 j 0 1 0 0 j j 0 0 1 0 j 1 0
(14)
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Figure 32: The structure of quadrature hybrid coupler