Four dual-mode dual-band PSIRR bandpass filters are fabricated on a substrate with r = 10.2 and thickness = 1.27 mm. The design procedure for N = 5, 6, and 8 is shown in Figure 2-19.
Figure 2-20 plots the simulated and measured results of the N = 8 dual-mode dual-band filter with S = 112.5o. It is designed to have f1 = 2.5 GHz and f2 = 4.72 GHz and fractional bandwidths 1 = 6.04% and 2 = 5.51%. The circuit parameters
are the same as those used in Figure 2-14. The corresponding line widths for the sections Z1, Z2 and Z3, are 0.36, 3.80 and 0.93 mm, respectively. The outer radius of the ring is 5.78 mm and the area occupied by the circuit is only 58.3% of a
conventional dual-mode ring filter operating at f1. The gap size between the line-to-ring coupler and the ring resonator is 0.15 mm, and 1 = 51o and 2 = 39o. The line-to-ring coupler has to support the two passbands simultaneously.
The patches Zp are used to trim the bandwidths. As shown in Figure 2-14, the distance between the two peaks at f2 is larger than that at f1, so it needs more coupling.
In this case, however, stronger coupling with larger 1 and 2 will cause the response at f1 to be over-coupled. Thus, the patches are added at virtual short-circuited positions for the even mode resonance at f2, leading to a shorter distance between the two peaks so that the required coupling level needs no more increment. At the same time these positions are the open circuit points for the resonances at f1. The patches bring both resonances at f1 to shift down together, but the change of bandwidth is negligible. The size of Zp is 0.50.37 mm2.
In Figure 2-20, the measured insertion losses are 1.9 dB and 1.5 dB and return losses are better than 18 dB and 15 dB at f1 and f2, respectively. The four transmission zeros are at 2.42 GHz, 3.5 GHz, 4.55 GHz and 5.85 GHz, offering good transition responses. The first spurious in the upper rejection band occurs at 6.3 GHz, which is close to the theoretical value (point A) in Figure 2-12. Figure 2-21 shows the photograph of the experimental circuit.
A similar design procedure can be applied to the PSIRRs with N = 5 and 6. A
resonant frequency f3a is much larger than f3b, a wider upper stopband can be achieved if the resonance at f3b can be suppressed. This can be done by allocating the zero f5z very close to it, leading to the separation between the input and output S = 62.5o, as shown by the dashed line in Figure 2-15.
For the resonant spectrum of N = 5 as shown in Figure 2-11, the resonant frequencies f3a and f3b are higher than f2a and f2b, and a desirable upper stopband can be realized easily. Here, we have a degree of freedom to select S for the transmission zeros, so that a wide stopband can also be achieved at the same time.
Figure 2-22 presents the simulated and measured results of the PSIRR bandpass filter with N = 6. Both the line width and gap size of line-to-ring structure are 0.2 mm.
The radius of the ring is 5.67 mm, and its normalized area is 57.2%. The circuit has f1
= 2.49 GHz and f2 = 4.77 GHz with fractional bandwidths 1 = 8.11% and 2 = 4.54%.
The measured |S21| at f1 and f2 are 2.39 dB and 2.2 dB, respectively, and |S11| at both frequencies are better than 15 dB. The two pairs of zeros located at both sides of passbands are 1.93 GHz, 2.57 GHz, 4.35 GHz and 5.49 GHz. Since f3b is totally suppressed by f5z, the circuit demonstrates an improved upper stopband performance as compared with the previous filter with N = 8. Again, two Zp patches are incorporated into the circuit for tuning the passbands. The measured data show reasonably good agreement with the simulation. Figure 2-23 shows the photograph of the measured circuit.
Figure 2-24 shows the simulated and measured results of the experiment filter
with N = 5. The circuit has f1 = 2.48 GHz and f2 = 4.55 GHz with fractional bandwidths 1 = 4.03% and 2 = 3.77%. Figure 2-25 is the photo of PSIRR bandpass
filter with N = 5. The radius of the resonator is 6.04 mm, and it occupies about 65% of the area of a conventional ring resonator at the first frequency. The circuit also demonstrates an improved upper stopband performance. In the case, we have the freedom in choosing the position of transmission zeros as compared with the PSIRR filter with N = 6. The four transmission zeros located at both sides of passbands are 2.37 GHz, 3.27 GHz, 4.42 GHz and 6.71 GHz. The measured result has good agreement with the simulation.
Figure 2-26 plots the simulated and measured results of the filter with N = 4. The value of /(1/4) is selected as 0.4. Note that f1a and f1b support the first passband and f2a and f3b establish the second one. The latter two resonances are non-degenerate
modes, like the design in [14]. The circuit has f1 = 2.47 GHz and f2 = (f2a + f3b)/2 = 5.83 GHz with 1 = 5.39% and 2 = 7.07%, respectively. Note that the ratio f2/f1 =
2.36 > 2. The measured |S11| and |S21| are –10.2 dB and –1.6 dB at f1 and –31.1 dB and –2.18 dB at f2. The |S21| glitch around 4 GHz confirms the fact that f2b is suppressed by f3z. The peak at 6.75 GHz is f3a, which is 2.8 f1 and has about 7% error compared with the curve shown in Figure 2-17. The deviation could be due to the parasitic effects of the impedance junctions which are not taken into the account in the
Figure 2-27 shows the photograph of the test circuit.
Coupling Gap
Feed Lines ri
ro
Figure 2-1 The uniform impedance microstrip ring resonator with feed line
Frequency (GHz)
1 2 3 4 5 6 7 8
21
| S | (dB )
-50
-40
-30
-20
-10
0
Frequency (GHz)
21
| S | (dB)
-20 0
0.5 0.75 1 1.25 1.5 1.75 2 2.25 2.5 -40
-60
-80
Patch
90o
Figure 2-3 Simulated response of the UIR with ro = 12.3 mm, ri = 11.7 mm and patch area = 2×3 mm2 when the space between input and output ports is 90°.
P
S Q
T
P
Z ,P
Z R
Figure 2-4 Uniform impedance resonator using even- and odd-analysis.
S(degree)
(c)
(a) (b)
(d)
Figure 2-7 Four UIR circuits with line-to-ring coupling of different coupled line lengths. (a) length = 0o. (b) length = 10o. (c) length = 30o. (d) length = 50o. All electrical lengths are at the first resonant frequency.
f
mzo/f
Figure 2-8 Simulation results of four UIR circuits with different coupled-line lengths.
Figure 2-9 The periodic stepped-impedance ring resonator with line-to-ring coupling
Figure 2-10 The periodic stepped-impedance ring resonator with N = 6 in the even- and odd-analysis.
f
nb/f
of
na, /f
oResonant frequencies
/f
oR
S= 90
of
2z/f
oFigure 2-17 The N = 4 periodic stepped-impedance ring resonator with line-to-ring
f
5z/f
o compensate the difference between the bandwidh twopassband
Given f2 / f1by selection R Choose PSIRR N number
Add / Tune line-to-ring coupled line
to synthesis two passbands Start
Split the degenerate modes by adding perturbation Z3
Use pole-zero design graph to select sfor transmission zeros
Satisfactory return loss Yes End No
Figure 2-19 Design procedure for N = 5, 6, and 8.
21
|S | , | S | (dB )
11Simulation Measurement
8 7
6 5
4 3
2 1
0 -5 -10 -15 -20 -25 -30 -35 -40 -45 -50
Frequency (GHz)
Figure 2-20 Simulation and measured of PSIRR with N = 8.
Frequency (GHz) -50
-45 -40 -35 -30 -25 -20 -15 -10 -5 0
1 2 3 4 5 6 7 8
Measurement Simulation
> 3.26f1
11|S |, |S | (dB)21
Figure 2-22 Simulated and measured results of the bandpass filter with N = 6. Circuit parameters: Z1 = 91.3 , Z2 = 30.4 , Z3 = 27.3 , Zp = 0.26 mm2, 1 = 27o, and 2 = 63o.
Figure 2-23 Photograph of PSIRR bandpass filter with N = 6 for the result in Figure 2-22.
|S |, |S | (dB)1121
Frequency (GHz) -50
0
Measurement Simulated
1 2 3 4 5 6 7 8 9
> 3.56f -10
-20
-30
-40 -45 -35 -25 -15 -5
1
Figure 2-24 Performances of dual-mode dual-band bandpass filter with N = 5. Circuit parameters: Z1 = 90.5 , Z2 = 30.2 , Z3 = 77.1 , Zp =0.38 0.27 mm2, 1 = 40o, and 2 = 0o.
8 7
6 5
4 3
2 0 -5 -10 -15 -20 -25 -30 -35 -40 -45 -50
Frequency (GHz) Measurement Simulation
11
|S |, | S | (dB)
21Figure 2-26 Performances of dual-mode dual-band bandpass filter with N = 4. Circuit parameters: Z1 = 91.3 , Z2 = 30.4 , Z3 = 25 , = 3.6 mm, 1 = 40o, and 2 = 40o.
Figure 2-27 Photograph of PSIRR with N = 4.
Chapter 3
New Miniaturized Dual-Mode Dual-Band Ring
Resonator Bandpass Filter With Microwave C-Sections
In this Chapter, a brief introduction of the microwave C-section is firstly given.
The microwave C-section has nonlinear phase shift property in frequency and is suitable for development of dual-band devices [17]. In [17], the microwave C-section together with two transmission line sections is used to design to possess phase changes of 90 o and 270 o at the first and second frequencies, respectively. Here, each C-section is used to substitute a transmission line section of designated electrical
length. By proper design of input/output coupling configuration, two transmission zeros can be created on both sides of each passband. The proposed dual-mode dual-band bandpass filter has an area reduction of better than 70% as compared with a conventional dual-mode ring filter. Measurement results of two fabricated circuits validate the analysis and theoretical prediction.