Summary

In this chapter, we specify the near-field region discussed in this thesis. A simple

formulation has been presented for computing the coupling coefficients between two

antennas that are placed in the near field of each other and are arbitrarily oriented.

Although the formula is complicated to some extent, it could be regarded as a near-field

counterpart of the Friis transmission formula. Based on the proposed formula and

program, all the information we need to compute the near-field coupling coefficient are

the 3D radiation patterns of each antenna and their relative orientation.

Chapter 3

Comparisons between the Formulation and HFSS Simulation

3.1 Introduction

In order to verify the proposed formulation, three classic scenarios in the UHF band are considered in this chapter. The design frequencies of all the antennas are at 915 MHz. The associated coupling coefficients between the transmitting and receiving antennas are computed and compared to those simulated using Ansoft HFSS. Please note that, in the HFSS simulation, the quantity |S21|² is used for comparison, which is obtained by assuming port 2, namely the receiving antenna in our cases, being perfectly

matched to its load impedance. For consistency, this condition can be included in our formulation merely by setting ΓRL = 0. Also, it must be mentioned that according to the

condition in (2.7) the coupling coefficients can be computed only when the antenna

separation is larger than the mean value of the largest dimensions for the transmitting and receiving antennas. Therefore, d ≥ 85 mm is chosen in the following examples.

3.2 Side-by-Side, Parallel Half-Wave Dipoles

Consider two identical, y-directed half-wavelength dipole antennas, one of which is placed at the origin and the other on the z-axis with a separation d. The former is

chosen as the transmitting antenna, while the latter is the receiving antenna. Since it is not difficult to derive the normalized vector far-field pattern of an ideal y-directed half-wavelength dipole based on its well-known z-directed counterpart, they are given as

Substituting (3.1) and (3.2) into (2.12) yields the desired spherical wave coefficients Bn. The computed coefficients Bn diminishes significantly for higher order terms when n > 7 leading to fast convergence of (2.7). In the HFSS simulation, the configuration of the dipole is shown in Fig 3.1, and the corresponding patterns at 915 MHz are shown in Fig 3.2.

Fig. 3.1 Geometry of the HFSS simulated dipole.

(a) (b)

Fig. 3.2 Simulated radiation patterns of the dipole at 915MHz.

(a) x-z plane and (b) y-z plane.

calculated coupling coefficient (by close-form pattern expression)

simulated S21 (by HFSS)

Fig. 3.3 Coupling coefficient versus antenna separation for polarization-matched dipoles.

The near-field coupling coefficient as a function of antenna spacing d computed by the proposed method and those obtained via HFSS are depicted in Fig. 3.3. Excellent agreement can be observed verifying the proposed formulation.

3.3 Side-by-Side, Polarization-Mismatched Half-Wave Dipoles

In the preceding subsection, the two dipoles are polarization matched corresponding to the best case in a two-dipole system. However, in most cases, they are arbitrarily oriented. Besides, to demonstrate the capability of our method, a scenario

having polarization-mismatched dipoles is also considered. In the current case, the receiving dipole lying on the y-z plane is rotated by 20° around its phase center. This

can be accounted for in our formulation simply by transforming the coordinate system of the vector far-field pattern of the receiving dipole accordingly. Using (2.13),

θA andφ_A defined in Fig. 2.3 are 20° and 90°, respectively. Accordingly the corresponding rectangular components of the receiving antenna are

cos 20 sin 20

xR R

yR R

zR R

f f

f f

f f

φ

θ

θ

⎧ = −

⎪ = ° ⋅

⎨ ⎪ = − ° ⋅

⎩

(3.3)

Likewise, Bn thus obtained diminishes significantly for n > 9 leading to fast convergence of (2.7). The coupling coefficients thus obtained are plotted in Fig. 3.4.

One can see that the coupling coefficients are smaller here than in the preceding case because of the polarization mismatch between the two dipoles.

50 100 150 200 250 300 350 400

calculated coupling coefficient (by close-form pattern expression)

simulated S21 (by HFSS)

20°

Fig. 3.4 Coupling coefficient versus antenna separation for polarization-mismatched dipoles.

3.4 Polarization-Matched Square Loop and Half-Wave Dipole

Here, a square loop antenna having its perimeter equal to a wavelength is used to replace the transmitting dipole in Section 3.2. The loop antenna is centered at the origin with the loop lying on the x-y plane and oriented in such a way that the resultant polarization is aligned with the y-directed receiving dipole. In the HFSS simulation, the structure of the square loop antenna and the patterns are shown in Fig 3.5 and Fig 3.6, respectively.

x y

Fig. 3.5 Geometry of the HFSS simulated square loop.

(a) (b)

Fig. 3.6 Simulated radiation patterns of the square loop antenna at 915MHz.

(a) x-z plane and (b) y-z plane.

We observe that the simulated pattern in the y-z plane is doughnut-shaped, while the x-z plane pattern is omni-directional with a slightly shaking at ±90°. This circumstance indicates that its normalized co-polarized component of the far-field pattern can be represented as an array composed of two parallel dipoles a quarter wavelength apart. Consequently, the radiation patterns of the square loop can be approximated using the principle of pattern multiplication with the element factor

2 2

while the array factor can be derived as

sin sin

1

^j2

AF e

π θ φ

= +

(3.6) The coupling coefficient for this setup can thus be computed as a function of the antenna spacing. The results are compared with those simulated and shown in Fig. 3.7.

In this example, the error is larger than previous two cases due to the far-field pattern of square loop is an approximation, which is not a exact solution used in previous cases.

50 100 150 200 250 300 350 400

calculated coupling coefficient (by close-form pattern expression)

simulated S21 (by HFSS)

Coupling Coefficient (dB)

Antenna Separation d (mm)

Fig. 3.7 Coupling coefficient versus antenna separation for polarization-matched square loop and dipole.

3.5 Summary

In the previous three scenarios, the agreement between the computed results and those simulated by HFSS indicates that the proposed method can be utilized to determine the near-field coupling coefficient as the relative orientation, the antenna spacing, and the far-field patterns of the transmitting and receiving antennas are known.

Chapter 4

Application in Near-Field UHF RFID System

4.1 Introduction

A RFID system is a spontaneous wireless data collection technology with a long history [18]. Depending upon their operating principle, RFID systems are classified into three categories: passive, semi-passive, and active. A passive RFID system is the least complex and cheapest, hence widely used for many applications. Without a power supply of a passive tag its own, the required energy to turn on the tag chip depends upon the electromagnetic field coupling from the reader. Accordingly, two different coupling techniques are further categorized: near-field coupling and far-field coupling.

Low frequency (LF, 125-134 kHz) and high frequency (HF, 13.56 MHz) RFID systems are short-range systems based on near-field coupling. On the other hand, Ultra-high frequency (UHF, 860-960 MHz) and microwave (2.4 GHz and 5.8 GHz) RFID systems are typically long-range systems based on far-field coupling. LF and HF RFID systems have been deployed in the market for many commercial applications.

However, the larger size of the antennas used in the LF/HF band systems confines their further development. Thus, it is straight forward to reduce antenna size by designing the system in a higher frequency band, such as the UHF band. In addition, the near-field

UHF RFID systems have other superiorities, including higher data rate, and lower manufacturing cost, making them suitable for item-level tagging.

The near-field UHF RFID system is composed of a reader and a tag just as in the ordinary RFID systems. The simplified system architecture is depicted in Fig. 4.1. The power generated by the reader circuitry Preader is transferred to the reader antenna, and then acquired by the tag antenna through near-field coupling. The power absorbed by the tag chip Pchip can be expressed as [19]

chip reader reader chip

P = P × τ × × C τ

(4.1) where τreader and τchip are the impedance mismatch coefficients of the reader and tag between the front-end circuitry and the associated antenna, respectively. They can be expressed as

Please note that since both the impedances of tag and chip are complex, we use a modified power wave reflection coefficient proposed by Kurokawa [20]. Equation (4.2) also indicates that the maximum power transfer occurs at conjugate impedance match between both components.

ZS

Z^T RFID

reader

RFID tag chip

Z^R Z^C

Coupling ZS

Z^T RFID

reader

RFID tag chip

Z^R Z^C

Coupling

Fig. 4.1 Simplified architecture of near-field RFID systems.

C is the coupling coefficient between the reader and tag antennas. With the aid of the proposed formulation (2.7), (2,12), the coupling coefficient C can be readily calculated.

To verify the results by experiment, a near-field UHF RFID system is implemented in this chapter. In Section 4.2, we present a broadband square loop array with a back- reflector and it is used as the reader antenna. In Section 4.3, two different tag antenna designs [21], [22] are used individually in the system. We compared a series of experiments with the proposed formulation in Section 4.4. The measurement results are in good agreement with those computed by the proposed method and those simulated by HFSS as well. Furthermore, some factors are found to be crucial for improving the power coupling level in Section 4.5. For practical applications, such as Point of Sale (POS), the proposed formulation is capable of determining the near-field read range and the read reliability, which is introduced in Section 4.6. Finally, we summarized the measured results and findings in Section 4.7.

4.2 Reader Antenna

Typically, a loop antenna is favorable for a near-field reader antenna. However, the square loop with perimeter of a wavelength demonstrated in Section 3.4 is not an appropriate design due to poor concentration of power. Therefore, we develop a loop array with back-reflector to aggregate the radiated power.

The geometry of the proposed reader antenna for the near-field UHF RFID system is shown in Fig. 4.2, and the photographs are shown in Fig 4.3. Two printed square loop antennas, of which the perimeters are equal to a wavelength, are back-to-back connected by a coplanar strip (CPS) of length Lcps. The two arms of the CPS are connected respectively at their midpoints to the inner and outer conductors of the feeding coaxial cable. The coaxial cable is fed from the direction normal to the antenna plane. Although a balun could be added to slightly improve the radiation performance, the proposed design directly fed via a coaxial cable can still provide satisfactorily higher gain and well-shaped radiation pattern. As one may expect, the design radiates bi-directionally;

however, most RFID reader antennas require unidirectional radiation pattern. To produce unidirectional radiation pattern and further increase the antenna gain, an electrically large, planar conducting sheet is utilized as a back reflector for the loop array as shown in Fig. 4.2 The spacing H between them is set to be approximately a quarter wavelength in free space.

2 mm 76 mm 76

mm

L

_cps

=68 mm 4 mm

400 mm

400 mm

H = 82 mm (Spacer) Loop array

2 mm 76

mm 76

mm

L

_cps

=68 mm 4 mm

400 mm

400 mm

H = 82 mm (Spacer) Loop array

Fig. 4.2 Geometry of two-element square loop array with a back reflector.

(a) (b)

Fig. 4.3 Photographs of two-element square loop array with a back reflector.

A prototype antenna designed around 915 MHz was fabricated on an FR-4 substrate with dielectric constant εr = 4.4 and thickness h = 0.6 mm. A copper sheet of dimensions 400×400 mm² is used as the back reflector and placed at a distance H = 82

mm from the loop array. Throughout the design process, simulations are carried out on HFSS. The simulated and measured input return losses of this antenna are shown and compared in Fig. 4.4. We can find that the design is well matched within a wide frequency range, and the measured 10-dB return loss bandwidth is 19.1% (848-1022 MHz). The peak gains measured at 920 MHz and 930 MHz are 10.2 and 10.1 dBi, respectively. Since the radiation pattern remains nearly the same throughout the return loss bandwidth, for simplicity, Fig. 4.5 depicts the x-z and y-z plane patterns measured at 920 MHz only.

0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3

-50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0

simulated measured

|S11| (dB)

Frequency (GHz)

Fig. 4.4 Simulated and measured input return losses of the reader antenna.

0

90

180 270

-10 dB

simulated E-phi simulated E-theta measured E-phi measured E-theta

10 dB

(a)

0

90

180 270

-10 dB

simulated E-phi simulated E-theta measured E-theta measured E-phi

10 dB

(b)

Fig. 4.5 Simulated and measured radiation patterns of the proposed reader antenna at 920MHz. (a) x-z plane and (b) y-z plane.

4.3 Tag Antennas

Passive tags utilize the coupled energy from a reader to power up the chip circuit.

To have a superior power transfer between the tag antenna and the chip, the input impedance must be conjugate matched to the chip impedance, which is highly capacitive generally. The capacitive reactance of chip impedance makes the matching task become difficult between tag antenna and chip. Thus the design guidance of tag antennas is miniaturized as well as a well-coupled power. Two different tag antenna designs, referring to [21] and [22] are implemented. Each of them is used in our near-field experiment setup.

4.3.1 Folded Dipole with a Closed Loop

The first one is a folded dipole with a closed loop [21] whose main advantage is its tunable input impedance to achieve conjugate match for various commercial tag chips.

A prototype antenna design at 915 MHz is depicted in Fig. 4.6, and the photograph of the antenna, of which the total antenna area is 64.4 × 27.6 mm², is shown in Fig. 4.7.

Please note that, to facilitate measuring the coupling coefficient through the vector network analyzer (VNA), the antenna is designed for 50 Ω instead of being conjugate

matched to the highly capacitive tag chips and is fed by a section of CPS connected to a balun connected to the coaxial cable. This additive balun degrades the

25.76 mm

27.6 mm

64.4 mm

9.2 mm 28.52

mm

Balun

Fig. 4.6 Geometry of the folded dipole antenna.

Fig. 4.7 Photograph of the folded dipole antenna.

pattern of the tag antenna inevitably, which will be stated and discussed at Section 4.3.3.

The proposed folded dipole was fabricated on the FR-4 with thickness of 0.6 mm.

The simulated and measured return losses of the folded dipole with the Balun are shown and compared in Fig. 4.8, and the radiation patterns of x-z plane and y-z plane are

shown in Fig. 4.9. Due to the fabrication errors and the uncertainty of the dielectric constant of FR4, the resonant frequency of the prototype antenna slightly shifts from the design frequency of 915 MHz to 920 MHz. The peak gain measured at 920 MHz is 2.2 dBi. We can utilize this tag for near-field applications by operating at a lower output power of reader, so the tag responds only to stronger fields in the vicinity of the reader antenna.

Fig. 4.8 Simulated and measured return losses of the folded dipole with the Balun

0.70 0.75 0.80 0.85 0.90 0.95 1.00 1.05 1.10

-14 -12 -10 -8 -6 -4 -2 0

Return Loss (dB)

Frequency (GHz)

Simulation Measurement

Fig. 4.9 Simulated and measured radiation patterns of the folded dipole with the Balun

4.3.2 Meander Circular Loop

The other design used is a meandered circular loop [22]. A print planar type circular loop antenna is able to reduce size by adding stubs attached to the structure. The prototype antenna designed at 915 MHz is depicted in Fig. 4.10. It is also designed for 50 Ω, and the same feeding structure is used for measurement’s sake. The fabrication of

the proposed meander loop was on the FR-4 with thickness of 0.6 mm. Fig. 4.11 shows the return losses obtained from simulation and measurement, and Fig. 4.12 shows the radiation patterns of the x-z plane and y-z plane. Both the return losses and radiation patterns of CPS-fed meander loop are evaluated with the Balun, and the peak gain measured at 930 MHz is 2.7 dBi.

Fig. 4.10 (a) Photograph of 930-MHz meander loop. (b) Geometry of meander loop fed by CPS and balun.

0.70 0.75 0.80 0.85 0.90 0.95 1.00 1.05 1.10

-50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0

Frequency (GHz)

Simulation Measurement

Return Loss (dB)

Fig. 4.11 Simulated and measured return losses of the meander loop with the Balun 16.5 mm

27 mm

(a) (b)

Balun 1 mm

(a) (b)

Fig. 4.12 Simulated and measured radiation patterns of the meander loop with the Balun at 930MHz (a) x-z plane and (b) y-z plane.

4.3.3 Microstrip-to-CPS Transition

A coplanar stripline (CPS) is a balanced transmission line which can be used for

balanced fed-in antennas such as dipoles and loops. On the other hand, a microstrip line

is an unbalanced transmission line, and is one of the most widely used transmission

lines in microwave circuits. In many cases, when a transition (balun) is used between

the microstrip line and the CPS, the overall antenna performance is limited by the balun

structure. At this section, we digress and discuss the balun used in the above-mentioned

tag antennas.

The proposed balun operates from 890 MHz to 945 MHz with an insertion loss

ratio (S ) less than 1 dB and return loss (S ) better than 10 dB for back-to-back

coupled CPS microstrip line microstrip feed

truncated ground plane (backside)

L1

L2

W1

W2

L3

y x z

Fig. 4.13 Proposed structure of the microstrip-to-CPS transition.

transition. In spite of a narrow band, however, its bandwidth is wide enough for our

application. Figure 4.13 shows the back-to-back uniplanar microstrip-to-CPS transition.

It consist of a 50 Ω microstrip line (W1 = 1.3 mm) which branched into two paths. The characteristic impedance of each microstrip branch is chosen as 100 Ω (W2 = 0.265 mm) for easy fabrication. The differential 180° phase difference between two microstrip line

branches can be accomplished by introducing a delay line where L₂−L₁=λ_g 4 [23]

with λg the guided wavelength in the microstrip. This result indicates that the dominant

mode of coupled-microstrip line is odd mode, which can be subsequently utilized as a

fed-in structure to balanced antenna. The gap of the CPS is 0.4 mm, strip width is 3 mm,

and the CPS characteristic impedance is 50 Ω verified by HFSS.

The limitation of the proposed balun is that it degrades the radiation patterns. In

Fig. 4.9 (b), the pattern in y-z plane is supposed to be omni-directional instead of

reflecting fields toward the positive z-direction. A similar phenomenon can be found in

Fig 4.12 (b). These inaccuracies are attributed to the truncated ground plane which

reflects the power from tag antennas, hence excessive power radiates in the y direction.

Two attempts have been made for eliminating such excessive power. The first one is to

extend the length of CPS. In our design, L3 is chosen as λ₀ 4 82 ≈ mm. The other

manner employs a 45° tapered ground plane, as shown in Fig 4.13. The current flowing

on both sides of tapered ground planes are canceled out due to their opposite phases.

Further optimization of the balun and antenna design is possible depending on the

requirements of particular application.

4.4 Measurement Results

The aforementioned reader and tag antennas are utilized to measure the coupling

coefficient in the near-field UHF RFID system. The experiment setup is shown in Fig.

4.14. The measurements are performed in an anechoic chamber, and the associated

coupling coefficients are obtained by measuring the |S21|² at the antenna terminals by the

HP8753D VNA depicted in Fig. 4.15. We introduce the experiments in two subsections

here, that is, the coupling coefficients versus longitudinal displacements and transverse

displacements, respectively.

Anechoic Chamber

Network Analyzer

HP 8753D RX TX (Fixed)

d

(20 to 500 mm, sample a point for every 5 mm )

Anechoic Chamber

Network Analyzer

HP 8753D RX TX (Fixed)

d

(20 to 500 mm, sample a point for every 5 mm )

Anechoic Chamber

RX Network Analyzer

HP 8753D RX TX (Fixed)

Variable d

Anechoic Chamber

Network Analyzer

HP 8753D RX TX (Fixed)

d

(20 to 500 mm, sample a point for every 5 mm )

Anechoic Chamber

Network Analyzer

HP 8753D RX TX (Fixed)

d

(20 to 500 mm, sample a point for every 5 mm )

Anechoic Chamber

RX Network Analyzer

HP 8753D RX TX (Fixed)

Variable d

Fig. 4.14 Measurement setup for the near-field RFID system.

Fig. 4.15 Photograph of HP8753D VNA.

4.4.1 Coupling Coefficient versus Longitudinal Displacement

First of all, the coupling coefficients are measured for various antenna separations

d, and two sets of data are acquired respectively using the two tag antennas. During the

measurement, the reader antenna is fixed, while the tag antenna is moved from d = 85

mm to 400 mm with an incremental step of 5 mm. Besides, the antennas are kept

polarization matched with the main beam maximum aimed at each other. A photograph

of the experiment setup is shown in Fig. 4.16. Also, note that the 3D far-field patterns

Eθ, Eφ needed in the formula require a finer sampling step of 1° due to the numerical integration, and therefore they are obtained by transforming the near-field measurement

data.

Fig. 4.16 Photograph of the measurement setup in anechoic chamber.

Tag antenna

Reader antenna

The coupling coefficients for the case using the folded dipole as tag antenna are

measured at 920 MHz, which is the resonant frequency of the fabricated folded dipole.

The measured, calculated, and full-wave simulated coupling coefficients are plotted in

The measured, calculated, and full-wave simulated coupling coefficients are plotted in

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