Multi-layered Complementary Conducting Strip Transmission Line

Recently, a new artificially engineered synthetic transmission line – the so-called complementary conducting strip transmission line (CCS TL) – was reported to be an effective means of miniaturizing microwave circuits [52]. The CCS TL has the following characteristics. It firstly provides wide design choices for making characteristic impedance of the transmission line, without changing the process parameters and material constants. Second, the meandered CCS TL exhibits less bending and adjacent coupling effects, as indicated by the slower change in characteristic impedance against the width variation in the TL than the conventional meandered microstrip used in the same fashion (See Fig. 5 in [52]). Therefore, a compact microwave circuit can be established using the meandered CCS TL, finally achieving miniaturization.

The CCS TL is made from a unit cell, which has dimensions that are much smaller than the operating wavelength. As shown in Fig. 3.1, a unit cell contains a mesh ground plane and a central patch with at least two series arms for cells in series (Fig. 3.1 (a)) and bent (Fig. 3.1 (b)) connection to the adjacent cells. The etched portion of the meshed ground plane complements to the central patch of the signal layer, forming a CCS TL.

Additionally, Fig. 3.2 shows a new multi-layer complementary conducting strips (CCS) transmission line (TL) configuration made of the meandered CCS TL realized by two metal-layers (Fig. 3.2 (a)), whose guiding characteristics have been well documented [52], and will not be repeated here.

(a)

(b)

Fig. 3.1 Unit cells of complementary conducting strip transmission line (CCS TL): (a) For series connection. (b) For bent connection.

W P W_h S

P

W S W_h

(a)

(b)

Fig. 3.2 Synthetic complementary conducting strip transmission line (CCS TL): (a) Meandered CCS TL. (b) Sandwiched meandered CCS TL.

M1 SUB1 M2

M2 SUB2 M3

M4 SUB3

On the other hand, the sandwiched CCS TL, which is realized by two meshed ground planes on the top and bottom surfaces (Fig. 3.2 (b)), is first time reported. All the meshed ground planes are connected by plated through-vias. The procedure for designing sandwiched CCS TL is similar to the meandered CCS TL reported in [52].

By applying various structural parameters, including period of CCS unit cell (P), the width of the central path (W), the width of the connecting arm (S), and the etch area of the mesh ground plane (W_h X W_h), the sandwiched CCS TL also provides wide design choices for making characteristic impedance of the transmission line, without changing the process parameters and material constants. Fig. 3.3 shows experimental results for comparing the guiding characteristics between the sandwiched CCS TL and conventional stripline in the identical laminated substrates. Clearly, the sandwiched CCS TL can provide wider impedance range than conventional meander CCS TL based on the same width of the signal line. Further more, the variation of propagation constant of the meandered CCS TL by changing the width of the signal is relatively smaller than that of the conventional meandered stripline. Notably, the slow-wave factor of the sandwiched CCS TL in the meander form is 2.188, exceeding the physical limit of the conventional stripline about 8%, revealing the potential of CCS TL for miniaturizing the planar circuits.

(a)

(b)

Fig. 3.3 The guiding characteristics of the sandwiched CCS TL and the conventional stripline: (a) real part of characteristic impedances, (b) normalized phase constants.

0.5 1 1.5 2 2.5 3 3.5 4 4.5 5

Notably, a four-layer substrate configuration was adopted throughout this chapter.

In such a configuration, Figs. 3.2 (a) and 3.2 (b) share a common meshed ground plane M2. Based on this integration scheme, the CCS TLs in different layers can be independently controlled for various circuit designs. However, attention must be paid to the isolation of the stacked CCS TLs in different layers. The perfect solid ground plane provides the highest shielding capability of any mesh ground plane. An investigation on the shielding capability of two isolated circuits using meandered CCS TLs in different layers follows.

Two filters with independent functions are designed using CCS TLs and integrated in the same four-layer substrate configuration. The first is the lowpass filter (LPF), which occupies M1 and M2 layers (Fig. 3.2 (a)). The second is the bandpass filter (BPF), which utilizes the M2, M3 and M4 layers (Fig. 3.2 (b)). These symmetrical filters are designed following the similar procedure to be reported in the next section. In Fig. 3.2, every substrate has an equal thickness of 0.06 mm (SUB1-through-SUB3). The area of overlapping of the two filters is approximately 95

% of the total area. Figures 3.4 (a) and (b) show the intrinsic frequency responses of two stand-alone filters based on the measured and simulated results.

(a)

(b)

(c)

Fig. 3.4 Multi-function module incorporating four-layer meandered CCS TLs. (a) Frequency responses of 2.4 GHz LPF in M1 and M2 metal-layers. (b) Frequency responses of 2.4 GHz BPF in M2, M3 and M4 metal-layers. (c) Measured

1 1.5 2 2.5 3 3.5 4 4.5 5

In the experiments, one filter is measured using the two-port vector network analyzer (VNA) and the other is terminated by two chip 50Ω resistors. The full-wave

simulations using Zeland^TM IE3D follows the same procedure. The cutoff frequency of the LPF is 2.75 GHz, and the out-band rejection is below 30 dB from 4.25 GHz to 4.7 GHz. The insertion-loss is approximately 0.92 dB, a little higher than the simulated value of 0.45 dB. The return-loss is below –10 dB from 2.38 GHz to 2.51 GHz. On the other hand, in Fig. 3.4 (b) the center frequency of the BPF is 2.51 GHz, and the return-loss is below –11.5 dB from 2.11 GHz to 2.91 GHz. The measured insertion-loss is about 1.48 dB, which is 0.39 dB higher than the simulated value.

Good agreement between the measurements and simulations for the two filters, show that the structural parameters and material constants are very close to the design values, as will be reported in the next section. Additionally, the transmission between port 3 and either port 1 or 2 is measured to evaluate the cross coupling between LPF and BPF. Figure 3.4 (c) shows the measured transmission coefficient across two filters.

Based on the measured results presented in Figs. 3.4 (a) and (b), the BPF passes the energy above 2.11 GHz with low reflection and the LPF rejects signals above 2.75 GHz. The electromagnetic energy can be distributed in a four-layer configuration between 2.11 GHz and 2.75 GHz. Figure 3.4 (c) plots the measurements for the adjacent-port coupling (|S₃₂|) and cross-port coupling (|S₃₁|). The filter is symmetrical,

so only port 3 is applied when port 4 is terminated. Although Fig. 3.4 (c) reveals the relatively high electromagnetic energy transmission between the two filters in different layers from 2.11 GHz to 2.75 GHz, the adjacent-coupling is maintained below -23 dB and the cross coupling is below -29 dB. Therefore, Fig. 3.4 (c) verifies that passive circuits in different layers of the stacked meandered CCS TLs can be well isolated from each other. In the case study, an isolation of more than 23 dB is achieved.