Chapter 1 Introduction
1.2 Reduced-Size Impedance-Transforming Broadband 180 o Hybrid
The 180o hybrid ring coupler, which is also known as the rat-race ring, is an essential component in microwave circuits, such as balanced mixers, multipliers, push-pull amplifiers and antenna feed network, etc. In chapter four, a reduced-sized impedance-transforming broadband 180o hybrid ring coupler is proposed. With a unit element at each port and an ideal phase inverter, the broadband 180o hybrid ring with impedance transformation between the input and output ports is proposed. After formulating the scattering matrix of the 180o hybrid ring, the values of circuit parameters are derived by the optimization method to fit the 180o hybrid ring to a Chebyshev response. We replace each line section by three sections of stepped impedance transmission lines using CPS/interdigital CPS structure to realize an ideal 180 o phase inverter and to reduce the length of each line section. The circuit analysis, design procedure, simulation and measurement results will be discussed.
Chapter 2
360 o Reflection-Type Phase Shifter
2.1 Introduction
Reflection-type phase shifters have many advantages such as good matching, low insertion loss, simple control, and low DC power consumption, so it is popular in phase shifter design. Using active components such as varactor diodes as reflection loads was first introduced in [1]. Varactors are commonly used as reflection loads so that the phase shift is limited to the capacitance tuning range of the varactor. The maximum phase shift can be extended by resonating the varactor with an inductance or changing the port impedance of the reflection load. But even when these methods are utilized, it is difficult to achieve a phase shift of 360o by a single varactor.
In order to get 360o phase shift range, several approaches have been proposed. The most popular approach is to cascade two series resonant varactors with a quarter- wavelength transmission line [2] [3], which is shown in Figure 2.1(a). For the dual varactor termination, the phase shift range of the reflection load is doubled. In [4]
(Figure 2.1(b)), it is shown that using tunable short-terminated combline filters as reflection load also provide double insertion phase. Another approach is the
combination of two series connected varactors separated by a quarter-wave transmission line and an impedance-transforming quadrature coupler [5] (Figure 2.1(c)). The required varactor capacitance variation is smaller if the impedance ratio of the impedance-transforming quadrature coupler is enlarged. Instead of the resonant load, a ladder-type reflection type phase shifter with six varactors and seven quarter-wave transmission shown in Figure 2.1(d) is proposed [6]. The total phase shift is the sum of the tunable phase of each varactor, so it can achieve large phase shift without inductors.
In addition to all of the above, using two different series tuned varactors in a parallel connection to achieve 360o phase shift was provided by B. T. Henoch and P.
Tamm [7]. The author obtained a 360o phase shifter which has minimum variation of total phase shift versus frequency. In [8], the original varactor impedance was transformed by the transmission line, and linear voltage-phase relationship is obtained.
In this chapter, we propose a full 360o phase shifter with the reflection load using two resonating loads in a parallel configuration, so there are four varactors in total. The circuit schematic is shown in Figure 2.2. The proposed phase shifter is designed with excellent match and linear voltage-phase relationship at Ku band, where parasitic effects of all components such as resistors and varactor diodes have to be considered.
Detail design procedure and measurement results will be discussed.
(a) (b)
(c)
(d)
Figure 2.1: Different approaches for reflection loads of the 360o phase shifter. (a) Dual varactor termination for doubling phase shift range [3]. (b) Tunable short-terminated combline filter [4]. (c) In-series cascaded connection of varactors with the impedance-transforming quadrature coupler [5]. (d) Ladder-type reflection loads [6].
Figure 2.2: Circuit schematic of the proposed 360o reflection-type phase shifter.
2.2 Design Procedure
2.2.1 Reflection-Type Phase Shifter
As shown in Figure 2.3, a generic reflection-type phase shifter is composed of a 3-dB 90o hybrid coupler and two identical reflection loads. Due to the 3-dB hybrid 90o coupler, input signal is equally divided by the coupler in quadrature phase. The phase of reflected signal is determined by the reflection coefficient of the reflection load.
Because the reflection loads are identical, two reflected signals are combined in-phase at output port. On the other hand, two reflected signals are cancelled out each other at input port. Thus, the phase shift provided by the phase shifter is equal to the phase shift of the reflection load.
Figure 2.3: Generic reflection-type phase shifter
2.2.2 Series Resonant Varactor
The proposed phase shifter is designed at the center frequency of 12.45 GHz.
MA-COM MA46H120, which is a gallium arsenide flip chip varactor diode, is used as the termination loads of the phase shifter. The circuit model is shown in Figure 2.4, where LS, RS, and CP are the series inductance, series resistance, and package capacitance of the varactor diode. The tuning range of varactor capacitance CJ is from 0.2pF to 1pF. However, suffering form parasitic effects of LS, RS, and CP, the characteristic of the varactor is not simply a capacitance at high frequency. Thus, the first step is to measure the characteristics of the varactor.
Figure 2.4: The circuit model of MA-COM MA46H120 varactor
Figure 2.5 shows two different ground structures the varactor diode mounted. The gray lines represent the microstrip line. Via holes connect the top metal to the bottom, which is a metallic ground plane. The varactor is mounted between the microstrip line and the ground area, and its reverse bias voltage is applied on the left side.
(a) (b)
(c) (d) Figure 2.5: Two ground structures of termination loads.
(a) Layout of the load with RF virtual ground.
(b) Equivalent circuit of the load with RF virtual ground.
(c) Layout of series resonant loads
(d) Equivalent circuit of series resonant varactor.
Figure 2.5(a) shows that the top view of the ground structure, composed of a shunt connection of a fan stub and a quarter-wave short stub, is a RF virtual ground and DC return. A fan stub can achieve a “perfect short” over a moderate bandwidth. On the other hand, the short stub using via holes acts like a “perfect open” for RF signals, so there is no via hole effect. Thus, we can derive the characteristics of the varactor by measuring the reflection coefficient. Figure 2.5(b) shows its equivalent circuit.
Measured equivalent reactance varies from -23Ω to -61Ω with tunable reactance ratio of 2.7 and total phase shift of 52o at the bias voltage range of 0V to 10V. Parasitic resistance varies from 3Ω(V=0V) to 5Ω(V=10V).
In order to increase the phase shift range of the varactor, a modified ground structure is utilized, which is shown in Figure 2.5(c). The diameter of via hole is 10mil. Using ground structure in Figure 2.5(c) an inductive series reactance is added to the equivalent circuit of varactor. This via inductance can be designed to form a series resonant circuit with the varactor at the center frequency
0
0
1 L CS
ω = (2.10)
where LS and C0 represent the via inductance and the average capacitance of the varactor, respectively. With the resonance of the varactor, the impedance variation is dramatically improved and large phase shift is achieved. The circuit model is shown in Figure 2.5(d).
The characteristic of reflection loads with RF virtual ground and with via holes for ground are measured against bias voltage from 0V to 10V at 12.45GHz. Measured phase shift of reflection coefficient of varactor diodes is shown in Table 2.1. As predicted, the varactor with a RF virtual ground operates like a capacitance and has a phase shift range of 52o. However, the phase shift range of the resonant varactor is improved to 119o. The later is 67o more than the former. The reflection coefficient of the resonant varactor is plotted for the voltage range of 0V to 10V in 1V steps on the Smith Chart in Figure 2.6, where the equivalent resistance varies with its bias voltage, as listed in Table 2.2.
Table 2.1: Measured phase of reflection coefficient of varactors at 12.45GHz
∠ Γ (degree) Bias
Voltage(V) Varactor Resonant Varactor
0 -130 109
Table 2.2 Measured resistance of the resonant varactor against bias voltage.
Bias voltage(V) 0 2 4 6 8 10
Resistance(Ω) 1.4 1.7 2.1 2.6 3.2 3.8
Figure 2.6: The reflection coefficient of the resonant varactor on the Smith Chart
2.2.3 Parallel Resonant Load
According to [2], if two branches of the varactors, each varactor being tuned with a series inductor to resonate at the low and high bias voltage (Vmin and Vmax) respectively, are connected in parallel to form a parallel resonant circuit at a control voltage between Vmin and Vmax, the phase shift could cover 360o when the varactors are biased form Vmin and Vmax. This can be realized if the original resonant loads at different bias voltage are transformed by adding the transmission line with different length [9] (Figure 2.7). However, controlling the inductance of lump elements is so difficult at high frequency that transmission line is adopted in the proposed circuit.
(a) (b)
Figure 2.7: Two series resonant varactor forming a parallel resonant circuit using (a) lump elements (b) transmission lines.
By tuning the length of the transmission line, the phase of each branch can be controlled arbitrarily. The reflection coefficient of each branch is shown in Figure 2.8(a), where V0 represents the point of equal control voltage when Γ1 and Γ2 are complex conjugate of each other. Instead of the maximum and minimum bias voltage of 0V and 10V, the series resonance is designed at bias voltage of 2V and 9V, which will be explained later. These two branches are connected in parallel to form a 360o reflective network when the varactor is biased from 2V to 9V, as shown in Figure 2.8(b). It is apparent that the reflection coefficient of the reflection load doesn’t follow a constant ρ circle on the Smith Chart due to the series resistance RS of the varactor.
The resistance is 1.7Ω and 3.5Ω at the series resonance. However, at the parallel resonance the return loss is much higher because the equivalent parallel resistance is
2 2
0 0
0 0 0 0
0 0
( ) //( )
2 2 2
P
R X X0
R R jX R jX
R R
= + − = + ≈ (2.1)
where R0, X0 represent the equivalent resistance and reactance of one branch at bias voltage of V0. For R0 = 2.8Ω and X0 = 19Ω, the parallel resistance RP is approximately 65Ω and the return loss is 20dB.
(a) (b)
(c)
Figure 2.8: Two series resonant varactor forming a parallel resonant circuit. (a) Two series resonant varactor. (b) Parallel resonant circuit. (c) Modified parallel circuit.
2.2.4 Changing Port Impedance of Reflection Loads
In order to increase RP and reduce the return loss at the parallel resonance, the port impedance of the reflection load is changed. For RS and RP giving the same reflection at series and parallel resonance, the port impedance is chosen to be
0 S P
Z = R R (2.2) It shows that RS is 3Ω at the series resonance point or bias voltage of 9V. Thus, from (2.2) the optimized port impedance should be 14Ω to have a minimum return-loss variation. The reason for choosing the series resonance at bias voltage of 2V and 9V is to avoid extreme RS when bias voltage is 0, 1, and 10V. The reflection coefficient of the modified parallel circuit is shown in Figure 2.8(c).
For transforming port impedance from 50Ω to 14Ω, a quarter-wave impedance transformer is used with its characteristic impedance of 26Ω. This is too low for microstrip line to be realized. Therefore, two cascaded quarter-wave impedance transformers are used in the proposed phase shifter. The principle of impedance transforming is illustrated in Figure 2.9. The equation of the input impedance Zin
which is related to the characteristic impedance Z1, is derived as
2 2
Figure 2.9: Two cascaded quarter-wave impedance transformers
If the input impedance Zin is designed as 14Ω and port impedance Z0 of 3-dB hybrid 90o coupler is 50Ω, from (2.13) the characteristic impedance Z1 is obtained to be 90Ω. Thus, we can use two cascaded quarter-wave impedance transformers with characteristic impedance of 90Ω and 50Ω to transform port impedance from 50Ω to 14Ω over a moderate bandwidth.
2.3 Simulation Results
The proposed 360o reflection-type phase shifter is fabricated on a substrate with a dielectric constant of 3.58 and thickness of 20 mils. Full-wave EM simulation (Sonnet) is used to model the effects of via holes and tee-junctions of the branch line coupler and transmission lines. The circuit layout of the proposed circuit is shown in Figure 2.10 and the design parameters and physical dimensions are listed in Table 2.3, where W and L denote the width and length of the microstrip line, and D denotes the diameter of the via hole.
Figure 2.10: The circuit layout of the proposed 360o reflection-type phase shifter.
Table 2.3 Physical dimensions of the proposed 360o reflection-type phase shifter.
W1 W2 W3 W4 W5 W6 W7 W8 D
66 46 14 90 116 45 49 45 10
L1 L2 L3 L4 L5 L6 L7 L8 L9
134 138 88 152 152 118 46 217 28 (Unit: mil)
The Simulated results of relative phase shift and insertion loss against control voltage at 12.45GHz are shown in Figure 2.11. With the shunt connection of two resonant loads, the phase shift of the phase shifter is more than 360o when bias voltage is raised from 0V to 10V. It shows that the insertion loss with bias voltage
below 3V is much smaller than other biasing conditions. In order to minimize the fluctuation of the insertion loss, the phase shifter is designed to operate at the bias voltage of 2V-9V, and within the bias voltage a total phase shift of 360o can be achieved. The scattering parameter S21 is plotted on the Smith Chart in Figure 2.12.
Figure 2.11: Simulated relative phase shift and insertion loss of proposed 360o phase shifter at 12.45GHz.
Figure 2.12: Simulated S21 of the proposed 360o phase shifter at 12.45GHz.
2.4 Fabrication and Measurements
The 360o reflection-type phase shifter was fabricated on RO4003 substrate with a 20-mil thickness and a dielectric constant of 3.58 using a copper etching process. The diameter of via hole is 10 mils, and four flip-chip varactor diodes are mounted using a low temperature indium solder. The photograph of the proposed phase shifters is shown in Figure 2.13.
Figure 2.13: Photograph of the proposed 360o reflection-type phase shifter
2.4.1 Performance against Control Voltage
Figure 2.14 shows measured performance with respect to varactor control voltage of the proposed 360o phase shifters at 12.45 GHz. The measured maximum phase shift is 450 o for the control voltage range of 0V-10V. It shows that for the control voltage range of 3V to 9V, a phase shift of 360o is achieved, and the average insertion loss is 5.1 dB and its variation is ±1.3 dB. The scattering parameter S21, which is plotted in Figure 2.13(c), follows nearly constant ρ circle on the Smith Chart except for the
extreme control voltages. Figure 2.15 shows the comparison of relative phase shift between the proposed 360o phase shifter, series resonant varactor, and a single varactor.
360o
(a)
2.6dB
(b)
(c)
Figure 2.14: Measured results of the proposed 360o phase shifter against control voltage at 12.45GHz. (a) Relative phase shift. (b) Insertion loss. (c) S21 on Smith Chart.
Figure 2.15: Comparison of relative phase shift between the proposed 360o phase
2.4.2 Performance against Frequency
The measured frequency response is measured in Figure 2.16 from 12.2 to 12.7 GHz.
The measurement is plotted at control voltage of 3V-9V, where a full 360o phase shifter is achieved. It shows that the maximum phase shift is larger than 360o, insertion loss is 4.8 1.6 dB, and return loss is better than 14 dB over 500MHz of the voltage stage of 3V-9V.
±
(a)
(b)
(c)
Figure 2.16: Frequency response of the 360o phase shifter. (a) Relative phase shift. (b) Insertion loss. (c) Return loss.
Chapter 3
Ku band Phased array
3.1 Introduction
Phased array plays an important role in microwave applications such as radar systems and smart antenna systems of wireless mobile communication. In a phased-array antenna system, phase shifters are used to control the radiation direction. These phase shifters have to meet high requirements for continuously adjustable phase shift.
Design of these phase shifters requires consideration of many factors such as large phase shift, low insertion loss, low phase error, bandwidth, low insertion-loss variation, simple control, low DC power consumption, and low cost for commercial wireless communication.
Reflection-type phase shifters (RTPS) have the advantages of simple control, low reflection, and low insertion loss. Using active components such as varactor diodes as reflection loads was first introduced in [1]. Thus, phase shift is restricted by tunable capacitance of varactor diodes. In order to maximize phase shift range of reflection loads, many approaches were reported. But when working on an effort to increase the phase shift range of phase shifters, insertion-loss variation increases. This
insertion-loss variation can be eliminated by adding a compensating resistance parallel to the reflection load [2]. Furthermore, in adaptive antenna receivers, the insertion loss and the insertion-loss variation can be compensated by the variable-gain low-noise amplifiers (VGLNA) located in front of the phase shifters.
In this chapter, a Ku band (12.2-12.7GHz) phased array using reflection type analog phase shifter is proposed. First, a phase shifter with large phase shift is discussed. Instead of using two varactors for each reflection load of the proposed 360o phase shifter in the previous chapter, the phase shifter uses only one varactor as the reflection load. We extends the maximum phase shift of the varactor, which was originally limited by the capacitance tuning range, from 52o to 286o by resonating the varactor with via inductance and using impedance transformers between the branch line coupler and the varactor. The circuit analysis, design procedure, and tradeoffs between phase shift range and insertion-loss variation are elaborately discussed.
Finally, a 4-element phased array with a group of dipole antennas and the proposed phase shifters is fabricated to verify the proposed phase shifter. Low noise amplifiers (LNA) are used to compensate the insertion loss of phase shifters.
3.2 Theory of Reflection-Type Phase Shifter
The basic building block of the reflection-type phase shifter composed of a 3-dB 90o
hybrid coupler and two identical reflection loads has been discussed in chapter two. In this chapter, however, the phase shifter uses only one varactor for each reflection load.
The analysis, design procedure, and fabrication of the phase shifter will be discussed.
3.2.1 Analysis
In the simplest design, a single varactor used as reflection loads is shown in Figure 3.1(a). The phase of reflection coefficient is determined by the maximum capacitance variation of the varactor diode. The phase shift can be increased by adding an inductance LS in series to the varactor, forming the series resonant circuit (Figure 3.1(b)). The maximum phase shift range is achieved at the resonance frequency
0
0
1 L CS
ω = (3.1)
where C0 represents the average capacitance value of the varactor.
To simplify this model, the parasitic resistance of the varactor and series inductance is combined as RS. Also, the inductance LS includes the parasitic inductance of the varactor. In the ideal case (RS = 0), the phase shift of reflection coefficient follows the unity circle of the Smith Chart. But for real varactor diodes with finite Q, the effective series resistance RS must be included in the circuit model. In this case, the reflection coefficient follows a constant resistance circle on the Smith Chart, and this series
resistance determines the insertion loss and insertion-loss variation with the varactor bias voltage.
(a) (b) (c)
Figure 3.1: Circuit model of reflection loads, (a) Varactor. (b) Series resonant varactor.
(c) Series resonant loads with compensating resistance.
A method for compensating the insertion-loss variation is to shunt a resistance RP to the series resonant load as shown in Figure 3.1(c). This compensating resistance is used to transform the constant resistance circle locus to a constant ρ circle on the Smith Chart. Then, the insertion loss maintains nearly constant and insensitive with the change of varactor bias voltage.
3.2.2 Design Formula
The reflection coefficient Γ of the reflection load in Figure 3.1(c) is given by
0 0
where XL is equal toωLS −1ωCvar. Thus, phase shift and return loss of the reflection
Since reflected signals are combined from two identical reflection loads at the output of the 3-dB 90o hybrid coupler, the scattering parameter S21 of the phase shifter is obtained as jα2 Γ, whereαis the loss of the 3-dB 90o hybrid coupler. Phase shift
From (3.5) and (3.6), the varactor reactance determines not only the relative phase shift but also the insertion loss. The maximum phase shift Δφ is For a given varactor capacitance range, the maximum phase shift can be increased by lowering the impedance level Z0 at the input of the reflection load, while the input
From (3.5) and (3.6), the varactor reactance determines not only the relative phase shift but also the insertion loss. The maximum phase shift Δφ is For a given varactor capacitance range, the maximum phase shift can be increased by lowering the impedance level Z0 at the input of the reflection load, while the input