Chapter 3 An Effective Way to Reduce the Thermal Noise
3.3 Reducing Thermal Noise of NMOS
In CMOS technology, the substrate parasitic impedance can induce the substrate thermal noise of RFIC circuits due to the leaky current through the drain/source to the substrate. We propose the new and effective method that is adding
Chapter 3 An Effective Way to Reduce Thermal Noise of NMOS Transistors an external and large resistance at the substrate node of NMOS to reduce the thermal noise of the substrate node injecting the drain node in the NMOS. In this thesis, we do no add this resistance at the PMOS. The reason is the source node of PMOS that connects to VDD. Since the source node connect to VDD, the thermal noise of substrate at PMOS increase. The source of NMOS connects to ground, so this phenomenon does not exist. To explore the method, a small signal equivalent circuit model of the substrate with an external added resistor is developed and is shown in Figure 3.5. In order to simplify the circuit and the component be negligibly small, we neglect the Rs
and Rds.
Figure 3.5 Equivalent circuit model of the substrate with an added resistor Rbx, which is located between the substrate node and the source node of the
RF NMOS transistor.
The impedance of the substrate,
Z
sub, is derived and given as：0 0
Chapter 3 An Effective Way to Reduce Thermal Noise of NMOS Transistors From equation (3-4), increase of the added external resistance, Rbx lead to reduction of the equivalent substrate resistance Rsub and theC is not affected by Rsub bx. So we know that Rsub is proportion to Rbx. Figure 3.6 shows the simplified equivalent circuit model of the substrate with an added resistor Rbx.
Figure 3.6 Simplified equivalent circuit model of the substrate with an added resistor Rbx.
Chapter 3 An Effective Way to Reduce Thermal Noise of NMOS Transistors
Figure 3.7 (a) shows the simplified equivalent voltage noise circuit model with the added external resistor and Figure 3.7 (b) shows the simplified equivalent current noise circuit model with the added external resistor.
Figure 3.7 the simplified equivalent noise circuit model with the added external resistor Rbx：(a) the voltage noise model and (b) the current noise model.
We choose the current noise model since the noise factor of the LNA and the phase noise of LC-VCO are based on current noise model to calculate. Here, the proposed method is applied to ultra-wideband (UWB) LNA and worldwide interoperability for Microwave Access (WiMAX) LC-VCO to validate its effectiveness. Figure 3.8 (a) illustrates the proposed UWB LNA architecture with an
Chapter 3 An Effective Way to Reduce Thermal Noise of NMOS Transistors external resistance Rbx added to the transistor M1. To explore the noise figure of the LNA, the noise factor is derived first and is equal to the ratio between the input noise power and the output noise power of the circuit.
Figure 3.8 Circuit schematics (a) Proposed UWB LNA (b) Proposed WiMAX LC-VCO In both circuits, the external resistor is added between the body and the source.
According to the proposed UWB LNA and using the substrate model shown in Figure 3.7 (b), after some derivations the noise factor of the LNA is given by
2 gate noise, and C is the correction coefficient for the gate noise and drain noise, and ω0 is the center frequency, ωT is the cutoff frequency. C2 is the correction coefficient
Chapter 3 An Effective Way to Reduce Thermal Noise of NMOS Transistors
for the gate noise and substrate noise. From equation (3-19), we find that the noise factor is direct proportion toZsubandZsubis inverse proportion to
R
bx. Therefore, thenoise factor,F , is inverse proportion to
R
bx. Figure 3.8 (b) shows the proposed WiMAX LC-VCO using an external resistance Rbx of transistor M1. According to the Hajimiri-Lee phase noise model and after some derivations, phase noise of the LC-VCO is given by resonator’s thermal noise. It is noted that with the external resistance insub2/Δ is a fpart of inn2/Δf equal to 4kTR g that is decreased with the external resistance sub mb2 according to equation (3-17). We propose the new method is effective since the most noise contribution is provided by NMOS in this LC-VCO topology.
In simulation, the TSMC 0.18-μm 1P6M CMOS process, the low power UWB LNA design is proposed. A low supply voltage of 1.5V is chosen, and the total power consumption is 9.0mW. This proposed method not only reduces the noise figure but also improve the input matching performance in the LNA. The simulation results are shown as Figure 3.9 (a).In the Figure 3.9 (b), it shows that the noise figure (NF) is smaller when Rbx=30kΩ to compare with the case without Rbx. It is found that the
Chapter 3 An Effective Way to Reduce Thermal Noise of NMOS Transistors noise figure is at least less than 2.7dB in 6.0~10.6GHz and its minimum value is 2.28dB at 6.5GHz. Furthermore, the simulation result shows that there is about 0.1 dB to 0.3 dB of noise figure (NF) reduction with common source UWB LNA.
(a)
(b)
Figure 3.9 Simulation results (a) S-parameters versus signal frequency of LNA; (b) Noise figure versus signal frequency with and without Rbx.
Chapter 3 An Effective Way to Reduce Thermal Noise of NMOS Transistors A low power and low phase noise WiMAX LC-VCO is proposed. A low supply voltage of 1.2V is chosen, and the core circuit power consumption is 0.996mW.
In the Figure 3.10, it shows that the phase noise of LC-VCO is smaller when Rbx=30 kΩ to compare with the case without Rbx. It is found that when LC-VCO operates at 3.5 GHz, there is about 7.0 dB and 4.0 dB of phase noise reduction at 100 kHz and 1 MHz offset frequency, respectively. In addition, the proposed LC-VCO operates at 3.5 GHz with phase noise of -121 dBc/Hz at 1 MHz offset frequency.
The phase noise versus Rbx as shown in Figure 3.11, we can find that when the value of Rbx is about larger than 30 kΩ, the phase noise almost limited. Therefore, it is a reason that why we choose the value of Rbx to be 30 kΩ. In addition, this proposed method is effective through mathematical derivations and numerical simulations for UWB LNA and WiMAX LC-VCO. The performance of the propose LNA and LC-VCO are summarized in Table 3.1 and Table 3.2, respectively, with comparison to other recently published papers.
Figure 3.10 Simulated results of phase noise versus offset frequency with and without Rbx.
Chapter 3 An Effective Way to Reduce Thermal Noise of NMOS Transistors
Figure 3.11Simulated results of phase noise versus Rbx.
Table 3.1
Summary of LNA performance and comparison with published LNAs.
Ref. Tech. BW
Chapter 3 An Effective Way to Reduce Thermal Noise of NMOS Transistors
Table 3.2
Summary of LC-VCO performance and comparison with published LC-VCOs.
Ref. Tech. Freq.
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
Chapter 4 Design of a Dual-Band LC-VCO for 2.5/3.5 GHz WiMAX
4.1 Introduction
A critical building block of almost any wireless or wireline transceiver is the local oscillator (LO). When use with a mixer, the LO allows frequency translation and channel selection of radio frequency (RF) signals. The LO is typically implemented as a phase-locked loop (PLL) as shown as Figure 4.1, wherein a voltage-controlled oscillator (VCO) is phase-locked to a high-stability crystal oscillator [30].
Figure 4.1 Block diagram of a PLL-based frequency synthesizer.
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
In the design of the frequency synthesizer, the most critical building block is the VCO, which dominates the PLL performance, such as phase noise and tuning range. The VCO is usually embedded in a PLL as a tunable frequency synthesizer to provide clean, stable, and more precise carrier signals for frequency up/down-conversion [31]. A high-frequency (HF) CMOS VCO has strict requirements in the transceivers of wireless communication systems. Low power consumption and low phase noise are the challenges in VCO designs. Typically, a VCO is usually comprised of a gain element and a resonator. The resonator determines the oscillation frequency, and when it is composed of energy-storing inductors and capacitors, it is often referred to as an LC tank. A voltage-controlled varactor diode allows the oscillation frequency of the VCO to be varied.
Recently, the demand for high-quality performances but low-cost solutions is raising in the transceivers of modern wireless communication systems [32].
Low–power operation can extend the lifetime of the battery and save money for consumers. The low power consumption can be achieved by reducing the supply voltage and/or the current in the VCO core circuit. Although the low voltage operation can relies on scaling down metal-oxide-semiconductor (MOS) threshold voltage VT, the low voltage limits the signal amplitude, which in turn limits the signal-to-noise ratio (SNR) and degrades the VCO performance. Kwok and Luong [33] proposed a transformer-feedback oscillator, which swing the output signals dynamically above the supply voltage and below the ground potential to increase the carrier power and to lower the phase noise. This approach does not reduce the VT but in fact it increases the effective dynamic drain-to-source voltage at a fixed DC voltage. In addition, the another challenge in designing VCOs is minimizing phase noise while maintaining smallest power consumption [34].
The low power is an important concern, but the phase noise performance must
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
be low enough since the most critical performance specification for an oscillator is phase noise. In a receiver, the phase noise of the LO limits the ability to detect a weak signal in the presence of a strong signal in an adjacent channel. In a transmitter, phase noise results in energy being transmitted outside of the desired band. To achieve low phase noise, we have discussed and proposed the new and efficient method to reduce the phase noise of a VCO without increasing power consumption in Chapter 3.
In this chapter, we will focus on how to design a low-power-consumption low-cost and low-phase-noise dual-band LC-VCO for WiMAX. In order to conform with Taiwan giving fresh impetus to WiMAX, we design a dual-band LC-VCO covering 2.5 GHz and 3.5 GH. The section 4.2 briefly describes the current-reused LC-VCO topology that can only operate with only half the amount of DC current compared to those of the conventional LC-VCO topology. The current-reused dual-band LC-VCO combined with an external and large resistor at the substrate node of NMOS is proposed in the section 4.3. In section 4.4, the simulated and measured results are compared.
4.2 Proposed Voltage Controlled Oscillator Architecture
Figure 4.2 shows two typical LC tank oscillators. Figure 4.2(a) uses all-NMOS cross-coupled pair to provide negative-GM and Figure 4.2(b) employs all-PMOS cross-coupled pair. In both structures, MOS coupled pair is an active element to compensate for the losses of the inductor and the capacitor.
The phase noise of PMOS cross-coupled pair oscillator is lower than NMOS structure since the intrinsic noise of PMOS is lower than NMOS. Nevertheless, the output power of NMOS cross-coupled pair oscillator is larger than PMOS structure.
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
To sum up, we can use the NMOS and PMOS cross-coupled pairs that is called complementary cross-coupled pair) to provide negative GM. There are several reasons why the complementary structure is superior to the all-NMOS structure [35].
Figure 4.2 Two typical LC tank oscillator structures.
1. The complementary structure offers better rise- and fall-time symmetry. It makes less up-conversion of 1/ f noise and other lower frequency noise sources.
2. The complementary structure offers higher transconductance for a given current, which results in a better start-up behavior.
3. The complementary structure also exhibits better noise performance for all bias points illustrated in Figure 4.3
As long as the oscillator operates in the current-limited regime, the tank voltage swing is the same for both oscillators. However if we desire to operate in the voltage-limited region, the all-NMOS structure can offer a larger voltage swing.
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
Figure 4.3 Phase noise for the complementary and All-NMOS.
Figure 4.4 illustrates the schematic of the complementary cross-coupled LC-VCO without the tail current source, which is adopted in this work. From the phase noise point of view, this topology reveals better noise performance than the one in Figure 4.3. This is due to the fact that the 1/f 3 noise of the topology without the tail current can only originate from the flicker noise of the MOS transistor switches.
Figure 4.4 Complementary cross-coupled LC-VCO without the tail current source.
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
These switches are expected to feature lower flicker noise than the tail current source that dominates the 1/f 3 noise, for two main reasons. First, the switches operate in triode region for large portions of the oscillation period; hence, they exhibit lower current flicker noise than the tail transistor that continuously operates in saturation.
Second, switched MOS transistors are known to have lower flicker noise than transistors biased in the stationary condition [36]. Nevertheless, the main drawback of this topology is a higher sensitivity of the frequency to the voltage supply (frequency pushing). This effect can be alleviated by using a supply voltage regulator.
Here, we propose the current-reused LC-VCO that uses both NMOS and PMOS transistor in cross-coupled pair as a negative conductance generator to achieve low power consumption easily. As shown in Figure 4.5, the series stacking of NMOS and PMOS allows the supply current to be reduced by half compared to that of the conventional LC-VCO while providing the same negative conductance. This topology is not only low-power-consumption but also low-cost since it only used one inductor and two MOS transistors, but the conventional LC-VCO used two inductors and four MOS transistors.
Figure 4.5 The current-reused LC-VCO.
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
The conventional and current-reused LC-VCOs operate at 3.4GHz to 3.7GHz as shown in Figure 4.6 and the tuning sensitivity (KVCO) of both topologies are shown in Figure 4.7.
Figure 4.6 Simulated tuning range of the conventional and proposed LC-VCO at 3.5 GHz.
Figure 4.7 Simulated KVCO of the conventional and proposed LC-VCO at 3.5 GHz.
Figure 4.8 shows simulated phase noise for both the conventional and current-reused LC-VCOs which operate at 3.4GHz to 3.7GHz. The simulated values for the conventional LC-VCO and current-reused LC-VCO -119 dBc/Hz and -117 dBc/HZ, respectively, at 1MHz offset frequency. In addition, the power consumption of the
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
conventional topology and the current-reused topology are 1.973mW and 0.996mW, respectively. Figure 4.9 shows the output power of conventional and current-reused LC-VCO. It is found that the minimum values of output power are -1.13 dBm and -2.11 dBm, respectively, in the conventional topology and the current-reused topology.
Therefore, we know that the phase noise and the output power performances of conventional LC-VCO are better than current-reused LC-VCO, but the power consumption of current-reused topology is lower than the conventional topology.
Figure 4.8 Simulated phase noise of the conventional and proposed LC-VCO at 3.5 GHz.
Figure 4.9 Simulated output power of the conventional and proposed LC-VCO at 3.5 GHz.
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
Although the proposed topology can operate with only half amount of DC current compared to that of the conventional topology, the phase noise of current-reused topology is higher than conventional topology. In the modern wireless communication systems, the low power is an important concern, but the phase noise performance must be low enough since the most critical performance specification for an oscillator is phase noise. Hence, the current-reused LC-VCO combines with the Chapter 3 proposed that is adding an external and large resistor, which is located between the substrate node and the source nod of NMOS transistor. As shown in Figure 4.10, the current-reused LC-VCO combined with the external and large resistor Rbx is proposed.
Figure 4.10 Current-reused LC-VCO combined with the external resistor Rbx.
The conventional, current-reused, and proposed LC-VCOs operate at 3.4GHz to 3.7GHz as shown in Figure 4.11. Figure 4.12 shows the simulated tuning sensitivity (KVCO) for conventional, current-reused, and proposed LC-VCOs which operate at 3.5GHz.
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
Figure 4.11 Simulated tuning range of the conventional, current-reused, and proposed LC-VCO at 3.5 GHz.
Figure 4.12 Simulated KVCO of the conventional, current-reused, and proposed LC-VCO at 3.5 GHz.
Figure 4.13 shows simulated phase noise for the conventional current-reused and the proposed LC-VCOs which operate at 3.4GHz to 3.7GHz. The simulated values for the conventional, current-reused, and proposed LC-VCOs are -119 dBc/Hz, -117 dBc/HZ, and -121 dBc/Hz at 1MHz offset frequency. In addition, the power consumption of the conventional topology is 1.973mW, but the current-reused and the proposed topologies are both only 0.996mW.
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
Figure 4.13 Simulated phase noise of the conventional, current-reused, and proposed LC-VCO at 3.5 GHz.
Figure 4.14 Simulated output power of the conventional, current-reused, and proposed LC-VCO at 3.5 GHz.
Figure 4.14 shows the output power of current-reused and proposed LC-VCO. It is found that the minimum values of output power are -1.13 dBm, -2.11 dBm and -2.16 dBm in the conventional, current-reused, and proposed topologies. Therefore, we know that the phase noise and the output power performances of conventional LC-VCO are better than current-reused and proposed LC-VCOs, but the power
Chapter 4 Design of a Dual Band VCO for 2.5 GHz and 3.5 GHz WiMAX
consumption of current-reused and proposed topologies is lower than the conventional topology.
From the simulated, the current-reused LC-VCO which combines with the Chapter 3 proposed that is adding an external and large resistor, which is located between the substrate node and the source nod of NMOS transistor can achieve low phase noise performance without increasing power consumption. Although the output power of proposed LC-VCO is the least less than the others, its still can be allow applying wireless communication systems. In order to achieve the low-power and low-phase-noise performance, we choose the current-reused LC-VCO combined with the external resistor to implement the dual-band LC-VCO. This proposed topology is not only to achieve low-power and low-phase-noise easily but also realizing to low-cost.
4.3 A Dual-Band LC-VCO for 2.5 GHz/3.5 GHz WiMAX
As wireless applications proliferate, demands for low-cost wireless communication which can support multiple bands and multiple standards with minimal hardware implementations are rapidly increasing [37]. In response to this, multiband terminals using multiple RF transceivers have been reported. This, however, increase die area or chip count in a radio, which, in turn, increases cost and complexity of radios. Another of the major issues in a dual-band transceiver is the
As wireless applications proliferate, demands for low-cost wireless communication which can support multiple bands and multiple standards with minimal hardware implementations are rapidly increasing [37]. In response to this, multiband terminals using multiple RF transceivers have been reported. This, however, increase die area or chip count in a radio, which, in turn, increases cost and complexity of radios. Another of the major issues in a dual-band transceiver is the