Chapter 3 The Broadband LTCC Doubly Balanced Mixer
3.4 Simulated results of the double-balanced mixer
The simulated results of the double-balanced mixer were obtained using the circuit simulator (Agilent ADS). Because the circuit is symmetrical, we can exchange the RF and LO ports. Figure 3.4-1 shows that conversion loss is less than 8dB in the RF frequency range of 2.4 to 6.4GHz. Figure 3.4-2 shows that conversion loss is less than 8dB in the RF frequency range of 1.75 to 6.5 GHz.
0 Figure 3.4-1 Conversion loss vs. RF frequency for RF balun center tap
IF=300MHz,PLO=14dBm IF=300MHz,PLO=14dBm
Figure 3.4-2 Conversion loss vs. RF frequency for LO balun center tap
Figure 3.4-3 and Figure 3.4-4 show the RF-IF, RF-LO and LO-IF isolation of the LTCC mixer. The LO-RF isolation and LO-IF isolation for the center tap of the RF secondary served as the IF output are good as discussed in Chapter 3.2. For the center tap of the RF secondary served as the IF output, the RF-IF isolations over the frequency range of 1.5 to 2.8GHz are poor. The same result (LO-IF isolation) can be obtained for the center tap of the LO secondary served as the IF output. The main factor is the design of the capacitor. For the center tap of the RF secondary served as the IF output, the capacitors aren’t perfect grounding for RF signals at low frequencies. Hence, the RF-IF isolation isn’t good at lower frequencies. Increasing capacitor values will improve the RF-IF isolation, but degrade the conversion loss of the IF signal.
0 10 20 30 40 50 60 70 80
1 2 3 4 5 6 7
RF frquency(GHz)
isolation(dB)
8 RF-IF RF-LO LO-IF
Figure 3.4-3 LO-IF, LO-RF, RF-IF isolation vs. RF frequency for RF balun center tap
0
Figure 3.4-4 LO-IF, LO-RF, RF-IF isolation vs. RF frequency for LO balun center tap
0
Figure 3.4-5 Conversion loss vs. IF frequencies for RF balun center tap
Figure 3.4-5 shows the IF bandwidth of the LTCC double-balanced mixer. The capacitors restrict the IF bandwidth. The larger IF results in larger conversion loss.
Chapter 4
Combline Filter with Capacitive Cross-coupling
4.1 Theory of the typical combline filter
A typical tapped combline filter is shown in Figure 4.1-1[16-20]. The lines are each short circuited to ground at the same end while opposite ends are terminated in lumped capacitors. As the capacitors are increased the shunt lines behaves as inductive elements and resonate with the capacitors at a frequency below the quarter- wave frequency. At the resonant frequency of the filter, the lines are significantly less than a quarter wavelength length. Thus, the larger the loading capacitances, the shorter the resonator lines, which results in a more compact filter structure with a wider stopband between the first passband and the second passband. If the capacitors are not present, the resonator line will be λ0/4 long at resonance, and the structure will have no passband. This is because the magnetic and electric couplings totally cancel each other out in this case.
In this type of filter, the second pass band occurs when the resonator line elements are somewhat over a half-wavelength long. So, if the resonator lines are
8
0/
λ long at the primary passband, the second passband will be centered at somewhat over four times the midband frequency of the first passband. If the resonator line elements are made to be less thanλ0/8 long at the primary passband, the second passband will be even further removed.
‧‧‧‧‧
Figure 4.1-1 Typical combline bandpass filter
1 2
Figure 4.1-2 Transformation of equivalent circuit of coupled line
θ
Figure 4.1-3 Schematic of J-inverter
input
Figure 4.1-4 Equivalent circuit of the combline filter
‧‧‧ CN
Figure 4.1-5 J inverter equivalent circuit for the combline filter
Figure 4.1-6 Lump element equivalent circuits of the combline filter
…
Figure 4.1-2 can be equivalent to a J-inverter with J = Y0cotθ while
as shown in Figure 4.1-3. Then, the combline filter in Figure 4.1-1 can be convert to the equivalent circuit as shown in Figure 4.1-4. The J inverter equivalent circuit for the combline filter is shown in Figure 4.1-5. Figure 4.1-6 is the lump element equivalent circuits of the combline filter.
90o
θ ≠
4.2 Phase relationships
Let the phase component of the Y-parameter S21 be denoted Φ21. Consider the series capacitor of Figure 4.2-1(a) as two port devices. The signal entering port 1 will undergo a phase shift upon exiting port 2. This is Φ21, and it tends toward . For the series inductor as shown in Figure 4.2-1(b), the phase shift is . For the shunt inductor/capacitor pairs in Figure 4.2-1(c), the phase shift at off-resonance frequencies is dependent on whether the signal is above of below resonances. For signals below the resonance frequency, the phase shift tends toward . However, for signals above resonance frequency, the phase shift tends toward .
90o
+ 90o
−
90o
+ 90o
−
The three-resonator structure of Figure 4.3-1 and Figure 4.3-2, which represents a cascaded triplet (CT) section using a capacitive cross-coupling between resonators 1 and 3. Path 1-2-3 is the primary path, and path 1-3 is the secondary path that follows the capacitive cross-coupling. In Figure 4.2-2(a), the phase shifts for two paths are given in Table 4.1. Above resonance, the two paths are in phase, but below resonance, the two paths are out phase. This destructive interference causes a transmission zero on the low-side skirt as shown in Figure 4.2-2(b). Stronger coupling between 1 and 3 causes the zero to move up the skirt toward passband. Decreasing the coupling moves it farther down the skirt.
Port 2 Port 1
Y=jωC
Φ21=+90∘
(a)
Port 2
Port 1 Φ21=-90∘
Y=-j(1/ωL)
(b)
Φ21=ang(Y21)
Resonant frequency -90∘
90∘ Port 2
Port 1
(c)
Figure 4.2-1 Phase shifts for series capacitor, series inductor and shunt inductor/capacitor pairs (a) Series capacitor, (b) Series inductor, (c) Shunt inductor/capacitor pairs
90∘
90∘ 90∘
+/- 90ο
3 2
1
dB
(a) (b)
Figure 4.2-2 CT section (a) Multi-path coupling diagram for CT section with capacitive cross-coupling (b) Possible frequency response
Below Resonance Above Resonance
90+90+90 = 270° 90-90+90=90°
90° 90°
Path 1-2-3
Result Path 1-3
In phase Out phase
Table 4.1 Total phase shifts for two paths in a CT section with capacitive cross-coupling
4.3 Design of the LTCC three-poles combline filter with cross-coupled capacitor
To meet the specifications in Table 1.1, the filter should generate the zero at 2.1GHz. As discussed in Chapter 4.2, we can design a CT-type filter using a capacitive cross-coupling between resonators 1 and 3 to generate the zero in the low-side skirt. The proposed structure in this design is shown in Figure 4.2-1, which shows the combline filter with cross-coupled capacitor modified from the typical combline filter. The coupling between adjacent resonators can be tuning easily with the direct-coupled capacitors. Edge coupling has restricted the bandwidth of the typical combline filter. The modified combline filter can achieve larger bandwidth
than typical edge-coupled combline filter through the direct-coupled capacitors.
Figure 4.3-2 shows the equivalent circuit of the modified combline filter.
SL3
Figure 4.3-1 Modified combline filter
C10 C20 C30
Figure 4.3-2 Equivalent circuit of the modified combline filter
Step 1: Choose the optimum loading capacitors for this design. The larger the loading capacitances, the shorter the resonator lines, which results in a more compact filter structure with a wider stopband between the first passband and the second passband. However, the circuit dimension restricts the values of the
MIM capacitors.
Step 2: Design the inductors of the resonators using transmission lines and these resonators resonate at 2.45GHz.
Step 3: Choose C12, C23, and C13 to meet the specifications in Table 1.1.
Step 4: Fine-tune all component values.
The corresponding component values in Figure 4.3-1 are C10 = C30 = 4.18 pF, C20 =3.93 pF, C12 = C23 = 0.9 pF, C13=0.319 pF and the length and width are 73 mil×6 mil for SL1, SL2 and SL3. Figure 4.3-3 shows that the simulation results using the circuit simulator (AWR Microwave Office). Figure 4.3-3 indicates that the low-side skirt zero is at 2.1GHz and return loss in the passband (2.4-2.5GHz) is less than –30dB.The filter also suppress the second harmonic, third harmonic and all the lower stopband signals. This design meets the specifications of the Table 1.1.
dB
Figure 4.3-3 Simulated response of the combline filter with Microwave Office
Dgree
Figure 4.3-4 Applying Y-parameter to analyze the transmission zeros
SL1 SL2 SL3
I/O port I/O port
C13
C23 C12
C30 C10 C20
Figure 4.3-5 Combline filter and cross-coupled capacitor
These transmission zeros occur at the frequencies where of and of the reminder part of the filter has the same magnitude, but opposite phase as shown in Figure 4.3-4 and Figure 4.3-5. This means that of the three-pole combline filter with capacitive cross-coupling at these frequencies will be zero. The location of zero
Y21 C13 Y21
Y21
agrees with discussion in Chapter 4.2. After the circuit simulation, we convert these values into the LTCC structure and simulate with the fully 3-D EM simulator (HFSS).
The ceramic substrate of the LTCC has dielectric constant of 33, stripline ground plane spacing of 16.2 mil and the conductor (thickness:0.4mil) is silver. The 3-D structure of the combline filter is shown in Figure 4.3-6. The size of the LTCC filter is 100 mil 75 mil× ×32.4mil.
Layer 1
Via effect
Layer 6 Layer 2
Layer 4 C
Layer 5 23
C30
Layer 3
C13
C20
C12
C10
C13
Layer 7
Ground plane
Figure 4.3-6 3-D structure of the LTCC combline filter
Extra zero
dB
Figure 4.3-7 EM simulated responses of the LTCC combline filter
From the EM simulated responses in Figure 4.3-7, the low-side skirt zero is at 2.1GHz and return loss in the passband (2.4-2.5GHz) is less than –30dB.
The simulation results meet the specifications in Table 1.1. Figure 4.3-7 indicates that the filter generates one extra transmission zero in the lower-side skirt. As shown in Figure 4.3-6, the extra transmission zero may be caused by the via-holes of the stripline. The modified combline filter is connected with the inductance L as shown in Figure 4.3-8.
C10 C20 C30
C12 C23 C13
SL3
SL2
SL1
I/O port I/O port
Figure 4.3-8 Modified combline filter with the inductance L L
dB
Figure 4.3-9 Simulated responses of the combline filter with Microwave Office
Degree
Figure 4.3-10 Applying Y-parameter to analyze the transmission zeros The filter with inductance L generates the extra zero in the lower-side skirt as shown in Figure 4.3-9. Besides, the inductance L also makes the transmission zero in the high-side skirt to lower frequency. The Y-parameter response is shown in Figure 4.3-10. The concept of using series L in the resonator is similar to that of [4,20]. The
value of the inductance L is 0.0065nH for the response in Figure 4.3-9. The inductance L is extremely small. Figure 4.3-12 shows the photograph of the LTCC combline filter. Figure 4.3-11 shows the comparison of the measured results and EM simulated results with GSG probe. The insertion loss in the passband (2.4-2.5GHz) is about 2.5dB. The return loss is less than –10dB over the frequency of 2.35 to 2.55GHz.The suppression of the measured results for second and third harmonic are less than –30dB.
dB
Figure 4.3-11 Comparison of measured results and EM simulated results (GSG probe)
Figure 4.3-12 Photograph of the LTCC combline filter
Chapter 5
Substrate and Shielding Box Effects
5.1 Substrate effects
In chapter 4, the responses of the LTCC bandpass filter are simulated and measured with GSG probe. But considering the practical environment, the LTCC bandpass filter must be mounted on the substrate. The overall performance should include the parasitic of the substrate. The substrate for the filter is Rogers RO4003, where the dielectric constant is 3.38 and the thickness is 20mil. Figure 5.1-2 shows the LTCC filter and substrate environment. As discussed in Chapter 4, the inductance L will cause the shift of the transmission zeros and second passband. Figure 5.1-1 shows the overall schematic of the LTCC filter.
C10 C20 C30
C12 C23 C13
I/O port I/O port
LTCC ground plane
Inductance L System ground
plane
Figure 5.1-1 Overall schematic of the LTCC filter
via holes I/O port
I/O port
I/O port I/O port
LTCC band pass filter
Figure 5.1-2 Measured environment with substrates
dB
Figure 5.1-3 EM simulated results for substrate environments
dB
Substrate environments
GSG probe
Figure 5.1-4 EM simulated responses with GSG probe and substrate environments
Inductance L
dB
small
large
Inductance L large small
Figure 5.1-5 EM simulation result with various via-holes
Figure 5.1-4 indicates that increasing inductance L will make the zero in the high-side skirt to lower frequency and the zero in the low-side skirt to high frequency.
From the simulated results in Figure 5.1-5, the number of plated via-holes can control the parasitic via-hole inductance. If the number of plated via-holes increases, the
parasitic via-hole inductance L decreases. The first transmission zero in low-side skirt shift to a lower frequency and the transmission zero in high-side skirt shift to a high frequency. Although the series inductance L will affect the response of the LTCC filter, we can fine-tune the values of the inductance of the transmission line, load capacitors, and coupled capacitors in the LTCC filter to meet the specifications for holding the same series inductance L. In Figure 5.1-6, the solid lines are the measured response of LTCC three-pole combline filter with the substrate. The insertion loss in the passband (2.4-2.5GHz) is about 2dB. The return loss is less than –10dB over the frequency of 2.25 to 2.55GHz.The suppression of the measured results for second and third harmonic are less than –20dB.
dB
Figure 5.1-6 Comparison of measured results and EM simulated results (mounted on the substrate)
Figure 5.1-7 Photograph of the LTCC combline filter mounted on the substrate
5.2 Shielding box effect
Figure 5.2-1 indicates that the responses of LTCC bandpass filter are affected by variations of box sizes. Table 5.1 shows the frequencies of the transmission zeros. The transmission zero at 2.1GHz generated by cross-coupled capacitor isn’t affected by different box sizes. When increasing the box size, the zero at the low-side skirt move toward high frequency and the zero at the high-side skirt move toward low frequency.
For GSG probe measurement, the responses are affected seriously by box size (parasitic effect).
dB
small large
Box size Box size
small large
Figure 5.2-1 EM simulated results with various box sizes
Box size 100×75 mil 100×80mil 100×85mil 100×90mil
Left zero 0.83GHz 1.00GHz 1.08GHz 1.19GHz Right zero 8.84GHz 7.7GHz 6.97GHz 6.32GHz Table 5.1 Various box sizes vs. the locations of transmission zeros
Chapter 6 Conclusion
In Chapter 2, the measured results of the modified Marchand balun agree well with the simulated results. Some of the possible reasons cause the difference between the simulation and measured results are: the air gap between two layers, the inductance of the via used to connect bottom transmission lines and ground, non-identical physical lengths of the two extended output ports and misalignment between two substrates. This kind of balun showed a good performance in spite of imperfect fabrication facilities leading to a misalignment between the two substrates.
Thus, this design allows considerable flexibility in the design procedure and yields reasonable tolerances for fabrication. Adding the transmission line between two couplers improves the performance of the Marchand balun significantly.
In addition, the compensated transmission line can be added between the LTCC spiral broadside coupled stripline to increase the bandwidth of the Marchand balun in Chapter 3. Then, the broadband LTCC double-balanced mixer can be implemented with two Marchand balun. The simulated results indicate that the LTCC double-balanced mixer can achieve broadband bandwidth.
In Chapter 4 and 5, the measured results of the three-pole combline filter with cross-coupling measured with GSG probe and the substrate environments are shown in Figure 6-1. The solid lines are the measured results with substrate environments and the dot lines are the measured responses with GSG probe. These measured results agree with the simulated results in Figure 5.1-4 except the location of zeros. The ceramic substrate of the LTCC has dielectric constant of 33. Under such high substrate dielectric constant, the performances of the filter are sensitive to the shrinkages of the LTCC. In a free sintering process, the LTCC is shrinking in X- and
Y- and Z-direction (thickness). Therefore, the shrinkage of the LTCC may cause the error of the dimension. In Figure 5.1-6, the suppression of the combline with the substrate environment is –24dB for second harmonic. Decreasing the series L (more via-holes) can move the high-side zero toward higher frequency. Then, the suppression can achieve –30dB.
Figure 6-1 Measured results with GSG probe and substrate environments
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