Chapter 2 Basic Antenna Theory
3.4 Simulation and measurement results
Fig. 3.17 shows the simulation reflection coefficient at different distances, Fig. 3.18 shows simulation antenna efficiency at 900 MHz with different distances are 5%, 20% and 58%, Fig. 3.19 shows the simulation antenna efficiency at 1800 MHz with different distances are 17%,45% and 78%, Fig.
3.20 shows the simulation antenna gain at 900 MHz with different distances are -8.5dBi,-1.5dBi and 4.5dBi, Fig. 3.21 shows the simulation antenna gain at 1800 MHz with different distances are -0.3dBi,4.5dBi and 5dBi, Fig. 3.22 shows 900 MHz simulation E-plane pattern with different distances, when the antenna closer to the human body model, the electromagnetic waves are absorbed more serious. Fig. 3.23 shows 900 MHz simulation H-plane pattern with different distances, when the distance of the antenna and human model is 5 mm, 15 mm and 50 mm, antenna radiation power affected the situation is more serious, Fig. 3.24 shows 1800 MHz simulation E-plane pattern at different distances, Fig. 3.25 shows 1800 MHz simulation H-plane pattern at different distances, table 3-1 shows simulation Antenna efficiency and gain with different distances.
The measurement structure as shown in Fig. 3.26, In this thesis, antenna measured by SATIMO measurement system, Fig. 3.27 shows the measurement reflection coefficient at different distances, Fig. 3.28 shows measurement antenna efficiency at 900 MHz with different distances are 4%, 22% and 34%, Fig. 3.29 shows the measurement antenna efficiency at 1800 MHz with different distances are 22%, 50% and 53%, it can find the antenna closed to the head model, antenna radiation energy is absorbed, Fig. 3.30 shows the measurement antenna gain at 900 MHz with different distances are -9.5dBi, -2.5dBi and 0.2dBi, Fig. 3.31 shows the measurement antenna gain at 1800 MHz with different distances are 2dBi, 4.3dBi and 5dBi, it can find the antenna gain easy to impact of the low frequency, when the distance are 50mm, the head model similar a reflection plane, antenna gain will raise, Fig.
3.32 shows 900 MHz measurement E-plane pattern with different distances, Fig. 3.33 shows 900 MHz measurement H-plane pattern with different distances, it can find the antenna closed to the head model, antenna radiation energy is absorbed, Fig. 3.34 shows 1800 MHz measurement E-plane pattern at different distances, Fig. 3.35 shows 1800 MHz measurement H-plane pattern at different distances.
3.5 Summary
The transmission distance and quality in the wireless transmission is very
importance, but the wireless communication standard for wireless transmission distance is very stringent, so it can’t heighten the wireless transmitter power, if want to extend the transmission range, height antenna reception sensitivity and antenna gain are methods, but the human body will be reflection and absorption in the antenna radiation and cause attenuation of the radiation performance, so the related communication system design must consider antenna placed of near human body, the communication system need to improve the quality of communication, and to limit the specific absorption rate (SAR) on the human body, through the numerical simulation results, hoping to reduce the phone's SAR value with the achieve better communication quality to make a balance.
Fig.3.1 PIFA antenna structure.
Frequency(GHz)
0.5 1.0 1.5 2.0 2.5 3.0
|S 11|(dB)
-25 -20 -15 -10 -5 0
Simulation Measurement
Fig.3.2 Reflection coefficient of dual-band PIFA antenna.
Fig.3.3 Dual-band PIFA antenna Smith Chart.
Fig.3.4 PIFA antenna efficiency at 900MHz.
900MHz 1800MHz
Frequency(GHz)
0.80 0.85 0.90 0.95 1.00
Ef fi cenc y (linear )
0.0 0.2 0.4 0.6 0.8 1.0
Simulation Measurement
Fig.3.5 PIFA antenna efficiency at 1800MHz.
Fig.3.6 PIFA antenna gain at 900MHz.
Frequency(GHz)
1.70 1.75 1.80 1.85 1.90
Ef fi cenc y (linear )
0.80 0.85 0.90 0.95 1.00
Gain(dB i)
Fig.3.7 PIFA antenna gain at 1800MHz.
Fig.3.8 PIFA antenna in E-Plane pattern at 900MHz.
-35 -30 -25 -20 -15 -10 -5 0
1.70 1.75 1.80 1.85 1.90
Gain(dBi)
Fig.3.9 PIFA antenna in H-Plane pattern at 900MHz.
Fig.3.11 PIFA antenna in H-Plane pattern at 1800MHz.
Fig. 3.12 Simulation electric field at 900MHz.
-35 -30 -25 -20 -15 -10 -5 0
Fig. 3.13 Simulation surface current at 900MHz.
Fig. 3.14 Simulation electric field at 1800MHz.
Fig. 3.15 Simulation surface current at 1800MHz.
Fig.3.16 Simulation structure.
Fig.3.17 Simulation reflection coefficient at different distances.
Fig.3.18 Simulation antenna efficiency at 900MHz with different distances.
Frequency(GHz)
0.80 0.85 0.90 0.95 1.00
Ef fi c ienc y
Fig.3.19 Simulation antenna efficiency at 1800MHz with different distances.
Fig.3.20 Simulation antenna gain at 900 MHz with different distances.
Frequency(GHz)
1.70 1.75 1.80 1.85 1.90
Ef fi c ienc y
0.80 0.85 0.90 0.95 1.00
Gain( dBi)
Fig.3.21 Simulation antenna gain at 1800MHz with different distances.
1.70 1.75 1.80 1.85 1.90
Gain(dBi)
-35 -30 -25 -20 -15 -10 -5 0
Fig.3.23 Simulation 900 MHz H-plane pattern at different distances.
Fig.3.24 Simulation 1800 MHz E-plane pattern at different distances.
-35 -30 -25 -20 -15 -10 -5 0
Fig.3.25 Simulation 1800 MHz H-plane pattern at different distances.
Table 3.1 Simulation antenna efficiency and gain with different distances.
900MHz
Fig.3.26 Measurement structure.
Frequency(GHz)
0.5 1.0 1.5 2.0 2.5 3.0
|S 11|( dB)
-20 -15 -10 -5 0
PIFA 5mm 15mm 50mm
Fig.3.27 Measurement reflection coefficient with different distances.
Antenna
Fig.3.28 Measurement antenna efficiency at 900 MHz with different distances.
Fig.3.29 Measurement antenna efficiency at 1800 MHz with different
Frequency(GHz)
0.80 0.85 0.90 0.95 1.00
Ef fi c ienc y (linear )
1.70 1.75 1.80 1.85 1.90
Ef fi c ienc y (linear )
Fig.3.30 Measurement antenna gain at 900 MHz with different distances.
Fig.3.31 Measurement antenna gain at 1800 MHz with different distances.
Frequency(GHz)
0.80 0.85 0.90 0.95 1.00
Gain( dBi)
1.70 1.75 1.80 1.85 1.90
Gain( dBi)
Fig.3.32 Measurement 900 MHz E-plane pattern with different distances.
Fig.3.33 Measurement 900 MHz H-plane pattern with different distances.
-40 -35 -30 -25 -20 -15 -10 -5
Fig.3.34 Measurement 1800 MHz E-plane pattern with different distances.
Fig.3.35 Measurement 1800 MHz H-plane pattern with different distances.
Chapter 4
Simulation and Measurement of RE for PCB
4.1 Introduction
The electromagnetic interference can be divided into three elements, it includes the noise source, the receiving, and the transmission path, the three elements are electromagnetic interference, electromagnetic susceptibility, and transmission mode. In the printed circuit board, the electromagnetic interference noise model can be divided into conduction and radiation mode, the two kind’s interferences except on different transmission medium and frequency. In the past, the electromagnetic interference definition is mutual influence of the systems to system, but the circuit system is increasingly complex, a circuit system is often not only a circuit subsystem exists, the system gradually unable to ignore interaction between the subsystems, so the electromagnetic interference problem does not only exist in systems to system, it must consider to the subsystem problem, the problem can be get the best resolution.
4.2 Design and analysis of PIFA antenna for the electric field distribution
Continuation of the previous chapter, when the antenna radiation, the
ambient electronic components will interfere and affect the product performance, EMC with the current, electric and magnetic fields and electromagnetic radiation coupling phenomena are related, coupling occurs in two ways. Are electric field coupling and magnetic coupling, the antenna from the radiation source coupled to the circuit noise, it could be the magnetic field coupling or both, how to control them? It must understand the coupling mechanism, and how to impact product. An example of communication system is shown in Fig. 4.1, the electronic circuit structure is very complex, it include many coaxial cable and electronic components, it will to cause interference if it near the antenna, EMI radiation is not broadband, the system will excite the resonance, the external cable maybe excitation the resonance.
Fig. 4.2 shows in simulation the antenna radiation, the current distribution of the coaxial cable; it can be find the current distribution of the different polarization. Fig. 4.3 is the simulation the sensitive electronic components from antenna radiation to receive the interfere and sense current, it can find the current strength in the simulation area, to improve the electromagnetic interference and enhance product performance, this chapter use PIFA antenna of the previous chapter to simulate the antenna radiation with electric field in the distribution of different distances, simulated distance at 0.3 cm, 0.5 cm and 1 cm, the effects of different distances of the surrounding electric field and current distribution.
4.3 Simulation and test results
This part, use the GEMS to simulate the current density on the PCB, the GEMS is based on the FDTD, the simulation distance is 0.3 cm, the simulation area is shown in Fig. 4.4, it use a PIFA antenna to replace PCB, to simulated electric field at 900 MHz and 1800 MHz is shown in Fig. 4.5 and Fig. 4.6. Fig.4.7 is shown the hardware implementation of the antenna, using the probe to scan the surface and recording results, the aperture field for meander line PIFA with simulated and measured at 900 MHz and 1800 MHz are shown in Fig. 4.8 and Fig. 4.9. The results of simulation and measurement are similar.
4.4 Summary
The wireless communications are more popular in nowadays. Most of the size of cellular in communications is compact with smaller size. The RF components on the PCB of cellular will have mutual electromagnetic interference. If the components of receiver are interfered by near field RF components, the total isotropic sensitivity (TIS) of cellular will be degraded.
The antenna is compact and near the PCB of cellular phone. The efficiency of antenna will also be degraded. Not only the total radiation power (TRP) but also the TIS will be reduced. The performance, such as throughput, bit error rate (BER) of cellular communication systems will be degraded due to the low TRP and TIS. The near field EMI of PCB is quite important for compact size communication systems, but how to measure the electromagnetic interference and locate the EMI sources on PCB is not an easy job, in next
surface current density on the PCB.
Fig.4.1. Structure communication system.
Fig.4.2. Simulated electric current at 900 MHz.
Fig.4.3. Simulated electric current.
Fig.4.4. Simulated area.
Fig.4.5. Simulated surface current density at 900 MHz.
Fig.4.6. Simulated surface current density at 1800 MHz.
Fig.4.7. Aperture field scan of antenna.
Fig.4.8. Simulation and measurement of aperture field at 900MHz.
Fig.4.9 Simulation and measurement of aperture field at 1800MHz.
Chapter 5
Build High Resolution Circuit Image
5.1 Introduction
In this chapter, how to solve the EMI of PCB to improve the performance of communication systems? By special arrangement of RF components on PCB will enhance the TIS and TRP of communications. How to measure the electromagnetic interference and locate the EMI sources on PCB is not an easy job. This chapter proposes the aperture field technique to get the high resolution surface current density on the PCB. In order to verify the technique, simulation the PIFA antennas with surface current density and near aperture field on PCB are analyzed by GEMS. Except for the simulation, PIFA antennas are implemented and aperture field are collected by planar near field range. By using aperture field transformation to the surface of PCB, the surface current density on the PCB is obtained.
5.2 Diffraction field by NF aperture
Fig. 5.1 shows in the aperture antenna radiation diagram, while the S is the radiation source region, the position vector of source point
,
on the aperture is:
x ˆ y ˆ
(5.1)
The aperture field intensity on the source point is：
(5.2)
,
A
is the magnitude of field intensity, ,
is the phase of the field intensity, Fˆ is the polarization unit vector of source point.The diffraction field intensity at fixed
R, ,
is：
(5.3)
When the boundary conditions not any restrictions, if the location of the observation point placed in the far field range, because distance faraway, the radiation source to the observation point R and the origin point from the observation point r can be regarded as nearly parallel to line, 𝑛̂ ∙ 𝑟̂ = 𝑛̂ ∙ 𝑅̂ = cos θ , 𝑎𝑛𝑑
1 𝑟
≪ k =2π λ
, then(5.4)
Whether radiation source or the distribution of the radiation field, even integration of the radiation field, use the Cartesian coordinates to expression is most appropriate, in the integral particular solution, there are three most common and convenient way of expression the plane, there are y-z plane, x-z plane, x-y plane, as shown in the Fig.5.2, if the three planes were used to analyze the distribution of the radiation field, the integral will be different.
Nevertheless, even the different integral of the different plane, the calculated result will be the same, the actual distribution of the radiation field. At different plane respectively：
n s d d
(5.5)
(5.6)
(5.7)
Radiation from the source to the different observation points of the path integral ：
(5.8)
Because it inability to know the actual surface current distribution, so using the equivalence principle, by the actual near field electric field transform to equivalent surface current.
Fig. 5.3 shows a simple example of the equivalence principle. In Fig. 5.3 (a) within the region of S is the actual source of the radiation field distribution, it include the flux current J and M, the M is imagine of the virtual field source, exterior is the free space, let S region exterior the radiation field and the medium retain, while the internal is presume no any radiation field, the internal become a free space, as shown in Fig. 5.3 (b). As the S region exterior still presence a radiation field, so the boundary of the S region, it will
The transformation formula structure as shown in Fig. 5.4, using measurement plane induct the magnetic field energy, equivalent to the surface current distribution on the PCB, as shown in equation 5.10.
J⃗=ẑ × (H
x
x̂+Hy
ŷ+Hz
ẑ) (5.10)From equation 5.10, it can equivalent to the surface current.
𝐽
𝑥 =−𝐻 𝑦
(5.11a)𝐽
𝑦 =𝐻 𝑥 (5.11b)
5.3 Simulation and test results
In this part, it will use simulation tools to simulated the near field electric field and surface current distribution in the different distances are 0.3 cm, 0.5 cm 1 cm, as shown in Fig.5.5, the simulated surface current for PIFA at 900 MHz and 1800 MHz is shown in Fig.5.6 and Fig.5.7. It can find the distribution of the surface current at different frequency, the Fig.5.8 and Fig.5.9 are show the simulated magnetic field density with the distance at 0.3 cm in 900 MHz and 1800 MHz, the Fig.5.10 and Fig.5.11 are show the simulated magnetic field density with the distance at 0.5 cm in 900 MHz and 1800 MHz, Fig.5.12 and Fig.5.13 are show the simulated magnetic field density with the distance at 1cm in 900 MHz and 1800 MHz, sorting out the simulation data of the magnetic field density, it can use the results, by the
current density on the PCB.
The surface current density is transformed from the aperture field by the formula. Fig.5.14 as shown in transformation formula structure with different scope, the transformation scope are 3.5 mm*3.5 mm and 6.5 mm*6.5 mm, when the measurement plane and antenna surface to farther, the integration range must to raise, the Fig.5.15 and Fig.5.16 are show the transformed surface current with the distance at 0.3 cm in 900 MHz and 1800 MHz and the integration range is 3.5 mm*3.5 mm, the Fig.5.17 and Fig.5.18 are show the transformed surface current with the distance at 0.3 cm in 900 MHz and 1800 MHz and the integration range is 6.5 mm*6.5 mm, it can find integration range will affect intensity of the surface current, the Fig.5.19 and Fig.5.20 are show the transformed surface current with the distance at 0.5 cm in 900 MHz and 1800 MHz, Fig.5.21 and Fig.5.22 are show the transformed surface current with the distance at 1cm in 900 MHz and 1800 MHz, The results of transformed surface current from transformation and simulation are quite similar.
5.4 Summary
Most of radiation emission (RE) from PCB will cause the system fail to pass the EMI regulation and degradation the sensitivity in communication system. Since the size of cellular communication system is smaller and smaller, the sensitivity of system is degraded by near field EMI. This chapter proposes a technique to predict the high resolution surface current density on PCB by planar aperture field above the PCB and current density on the PCB.
The aperture field of PCB antenna with dual bands are measured by planar near field scanner. The measured data is transformed to the surface of PCB to get the surface current density on the PCB. The results of PCB current density from both measurement and simulation are quite similar.
Fig.5.1 the diffraction field by near field aperture.
Fig.5.2 (a) Plane position and rectangular aperture antenna radiation analysis: x-z plane.
dz
dx
Fig.5.2 (b) Plane position and rectangular aperture antenna radiation analysis: y-z plane.
Fig.5.2 (c) Plane position and rectangular aperture antenna radiation
analysis: x -y plane.
dy
dx dy
dz
Fig.5.3 Equivalence principle.
Fig.5.4 Transformation formula structure.
Fig.5.5 PIFA antenna structure.
Fig.5.6 Surface current at 900 MHz.
Fig. 5.7 Surface current at 1800 MHz.
Fig. 5.8 Magnetic field at distance 0.3 cm in 900 MHz.
Fig. 5.9 Magnetic field at distance 0.3cm in 1800MHz.
Fig. 5.10 Magnetic field at distance 0.5 cm in 900 MHz.
Fig. 5.11 Magnetic field at distance 0.5 cm in 1800 MHz.
Fig. 5.12 Magnetic field at distance 1 cm in 900 MHz.
Fig. 5.13 Magnetic field at distance 1 cm in 1800 MHz.
Fig.5.14 Transformation formula structure with different scope.
Fig. 5.15 Transformed surface current at distance 3mm in 900 MHz.
Fig. 5.16 Transformed surface current at distance 3mm in 1800 MHz.
Fig. 5.17 Transformed surface current at distance 3mm in 900 MHz.
Fig. 5.18 Transformed surface current at distance 3mm in 1800 MHz.
Fig. 5.19 Transformed surface current at distance 5mm in 900 MHz.
Fig. 5.20 Transformed surface current at distance 5mm in 1800 MHz.
Fig. 5.21 Transformed surface current at distance 1cm in 900 MHz.
Fig. 5.22 Transformed surface current at distance 1cm in 1800 MHz.
Chapter 6 Conclusion
The EMI / EMC problem is prompted by the current on the conductor change to generated electromagnetic radiation; similarly, the external electromagnetic field energy can also be changes the circuit current. Most of high-speed and fast rise time signal will produce the EMI / EMC problems, these problems will be connected to the equipment the cable, the typical solution is shield and filter on the input and output signals on power line. This thesis proposes the aperture field technique to get the high resolution surface current density on the PCB, by using aperture field transformation to the surface of PCB, the surface current density on the PCB is obtained. Almost all of EMI interference is produce current in the product of the somewhere, if these currents can be properly controlled, it only contains work required for the harmonic, reduce the high frequency harmonic unnecessary interference, the EMI / EMC problems will be solved. If during the design phase, can consideration the EMC problem, it will be save time and cost.
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