新型低功率與高雜訊抑制電生理訊號量測系統之前端電路設計與分析

(1)

新型低功率與高雜訊抑制電生理訊號量測系統之前端電路設計與分析

學生:林韋霆指導教授:吳重雨教授

電子工程學系電子研究所碩士班

中文摘要

在無線網路的蓬勃發展下，許多結合無線網路的應用也由此應運而生。居家醫療看戶及遠距醫療即為其中的一項應用，而建構『電生理訊號量測與監控系統之無線化與晶片化』即為實現居家看護與遠距醫療的第一步。當一個生醫量測系統能夠結合無線網路時，我們便能夠在任何地方和任何時間下，輕易的監控一個人的身體狀況。

本論文將設計一個應用於『電生理訊號量測與監控系統』晶片之前端電路，

在此前端電路中，包括了兩個部份: 一個用以抑制人體雜訊之儀表放大器以及一個低頻的帶通濾波器。在本論文中，提出一個擁有高共模抑制比的儀表放大器，將有效抑制人體中大量的低頻雜訊，而後端的低頻帶通濾波器的設計將使用『跳蛙架構』 (Leapfrog Structure)來減低元件飄移對於電路表現的影響。

此電生理訊號量測之前端電路，使用台灣積體電路 0.35 微米製程模擬，此電

路第一級的儀表放大電路操作於 3.3 伏特，功率消耗為 0.1 毫瓦，頻寬為 2 至 16

千赫茲，有0 至 60dB 之電壓增益，共模拒絕比率可達到 200 分貝至 250 分貝以抑

制人體中之大量低頻雜訊。在此儀表放大器的後端，是一個可濾出不同所需人體

(2)

訊號之低頻濾波器，而此操作於3.3 伏特下之低頻濾波器，功率消耗為 0.13 毫瓦，

濾波器頻寬為50 赫茲至 2 千赫茲，並有可調整之介於 0~60 分貝的電壓增益。整

體的前端電路消耗0.23 毫瓦，有 200 至 250 分貝之共模拒絕比率，頻寬為 2 至 16

千赫茲。

(3)

THE DESIGN AND ANALYSIS OF NEW LOW POWER HIGH CMRR CMOS FRONT-END CIRCUITS FOR

ELECTROPHYSIOLOGICAL SIGNAL MESUEMENT SYSTEM

Student: Wei-Ting Lin Advisor: Prof. Chung-Yu Wu

Department of Electronics Engineering & Institute of Electronics National Chiao Tung University

ABSTRACT

With the growing development of wireless networks, many applications combined with wireless networks come into existence. Home nursing and remote medical care is one of the applications. And constructing the electrophysiological signals measurement and control system is the first step to realize the home nursing and remote medical care.

While the measurement system is combined with wireless network, we can easily monitor the physical condition of people.

In this thesis, we will design the front-end circuit of the electrophysiological signals measurement and control system. The front-end circuit is composed of two parts, the instrumentation amplifier(INA) to compress the noise of human bodies and the low- frequency bandpass filter. In this thesis, we will announce a instrumentation amplifier with high common-mode rejection ratio (CMRR) which can effectively compress the noises in human bodies. The low-frequency bandpass filter behind the instrumentation amplifier uses leapfrog structure to decrease the influence on circuit performance due to devices variation.

(4)

The front-end circuits of the electrophysiological signals measurement simulates with TSMC 0.35um technology. The first stage INA operates at 3.3V, consumes the power of 0.1mW. Its bandwidth is between 2~16 kHz, and its CMRR archives the magnitude 200~250 dB to suppress the enormous low-frequency noises in human bodies. Behind the INA is the low-frequency bandpass filter which filters different human-body signals. The low-frequency bandpass filter operates at 3.3V and consumes 0.13mW. Its bandwidth is from 50Hz to 2kHz and has the tunable gain between 0~60dB.

(5)

ACKNOWLEDGEMENTS

首先我要感謝我的指導老師吳重雨教授多年來耐心的指導與鼓勵，使我能夠順利完成碩士學位。在吳教授循序漸進的諄諄教誨下，除了讓我獲得許多積體電路設計的專業知識外，更學習到挑戰困難及解決問題的態度與方法，而在與國外的學術交流中，也讓我因此拓展了國際的視野。這些的訓練與磨練也讓我在交大電子所的這幾年來，成長許多。

此外，在這段求學過程中，感謝 307 實驗室豐富的軟硬體資源以及幫忙建立這些資源的人員們，由於你們辛苦的維護與建立這些軟硬體資源，讓我能夠在這樣的環境中，順利的利用這些資源來完成我的研究以及碩士論文。

其次我要感謝施育全、鄭秋宏、廖以義、黃冠勳、林俐如、王文傑、江政達、

蘇烜毅、虞繼堯、黃鈞正等學長姐們，對我在研究上面的指導以及幫助。另外還要感謝陳勝豪、陳旻珓、陳煒明、黃祖德、蔡夙勇、楊文嘉、顏承正、蕭淵文、

盧台祐、陳建文、吳書豪、曾瑋信、李宗霖、丁彥、謝致遠、黃如琳、蕭勝文、

莊凱嵐、郭秉捷、林棋樺等實驗室同學們，一起研究課業以及在生活上的相互幫助，陪伴我度過這些年來的研究生涯，希望大家都能在學業或工作上順順利利。

另外，我要感謝我的朋友，丕哥、毓康、思瑋、俊憬、雲清、銘德、一斌、

牛、智堯、志偉、小山、清大口琴社的夥伴以及附中 875 的好同學們，謝謝你們，

有你們陪伴的出遊、打球、以及生活上的分享及幫助，讓我在研究之餘，也能夠有一個非常快樂開心的休閒生活。

最後，我要感謝我親愛的父親林炳輝，母親廖伶玉，弟弟林哲雋以及一直耐心陪伴著我的資穎，謝謝你們對我無怨無悔的支持，我才能一路這樣的走過來，

你們一向是我努力的原動力，我愛你們，尤其是媽媽的細心栽培與無悔的付出，

才會有今天的我。把這份論文獻給我最愛的你們。

林韋霆誌於風城交大九五年七月

(6)

Table Captions

Table 2.1 The size of the DDGm amplifier in Fig. 2.10...27

Table 2.2 The specification of the low-frequency bandpass filter ...28

Table 2.3 Gm values of the Gm amplifier in Fig. 2.17 ...35

Table 2.4 The size of the DDGm amplifier in Fig. 2.10...37

Table 3.1 Simulation results of the INA...50

Table 3.2 Simulation results of the low-frequency BP filter...52

Table 3.3 Simulation results of the front-end circuit of the electrophysiological signal measurement system...54

(9)

Figure Captions

CHAPTER 1

Fig. 1.1 Schematic block diagram of the biotelemetry system...3

Fig. 1.2 Conventional Three-op instrumentation amplifier...5

Fig. 1.3 The symbol and the matrix form of second generation current conveyor ...5

Fig 1.4 Conventional current-mode instrumentation amplifier ...6

Fig. 1.5 Current-mode instrumentation amplifier using current cancellation by an operational amplifier ...7

Fig. 1.6 Current-mode instrumentation amplifier using current inversion...8

Fig. 1.7 Current-mode instrumentation amplifier...9

Fig. 1.8 Current-mode instrumentation amplifier using bias current sensing of opamps .. ...10

Fig. 1.9 Choice of filter type as a function of the operating frequency range ...12

Fig. 1.10 The Nauta’s Gm amplifier ...13

Fig. 1.11 The rail-to-rail Gm amplifier ...14

Fig. 1.12 The conventional Gm amplifier ...15

Fig. 1.13 Common-drain amplifier (voltage follower)...16

Fig. 1.14 Common-drain amplifier with output loads...16

CHAPTER 2

Fig. 2.1 The symbol for Differential Difference Operational Amplifier...19

Fig. 2.2 The symbol for Differential Difference Operational Transconductance Amplifier ...20

Fig 2.3 Expression for the new approach of instrumentation amplifier ...21

Fig 2.4 Subtraction of input signals using two DDGm ...22

(10)

Fig. 2.5 The new structure of high CMRR instrumentation amplifier ...23

Fig. 2.6 The function of the first stage and waveform of V_X₁ and V_X₂ ...23

Fig. 2.7 Flipped Voltage Follower (FVF) ...25

Fig. 2.8 The DDGm amplifier using FVF method ...26

Fig. 2.9 (a) The prototype LC network of the 5th-order Elliptic bandpass filter (b) The LC network after element value normalization. (c) The LC network after element position change for leapfrog structure ...29

Fig. 2.10 (a) The LC network of the 5th-order BP Elliptic filter (b) The block diagram of leapfrog structure for the 5th-order BP Elliptic filter (c) The block diagram of leapfrog structure for the 5th-order BP Elliptic filter ...30

Fig. 2.11 The simulation result of 5^th –order Elliptic LC ladder ...31

Fig. 2.12 Circuits implementations for functions of the functions in Fig 2.10 ...32

Fig. 2.13 The two port network of Gyrator ...33

Fig. 2.14 (a) The realization of a grounded inductor (b) The realization of a floating inductor...33

Fig. 2.15 (a) The realization of a grounded inductor composed of Gm and capacitor (b) The realization of a floating inductor composed of Gm and capacitor....34

Fig. 2.16 The low-frequency bandpass Gm-C filter with leapfrog structure ...35

Fig. 2.17 The circuit of the Gm amplifier ...36

Fig. 2.18 The circuit of the Gm amplifier connected with MOS resistor...37

Fig. 2.19 The method to eliminate the background current of the Gm amplifier...39

Fig. 2.20 The finite Ro of Gm amplifier in the Gyrator ...40

Fig. 2.21 The Gm amplifier with a capacitor load ...41

Fig. 2.22 The transfer function of the filter with ideal and nonideal Gm...42

(11)

Fig. 2.23 The output stage of the Gm amplifier ...43

CHAPTER 3

Fig. 3.1 The gain comparison of the conventional voltage follower and FVF versus ...44

RG Fig. 3.2 The frequency response of the Gm amplifier with different Gm values...45

Fig. 3.3 The frequency response of the Gm amplifier with nA/V Gm values ...45

Fig. 3.4 The calculated and HSPICE simulated Iout versus R_G...46

Fig. 3.5 The differential-mode signal of node V_X₁ and V_X₂ ...47

Fig. 3.6 The common-mode signal of node V_X₁ and V_X₂...47

Fig. 3.7 The frequency response of the proposed INA structure...48

Fig. 3.8 The frequency response of the proposed INA structure with four corners ...48

Fig. 3.9 The HSPICE Monte-Carlo simulation of the frequency response of the proposed INA structure ...49

Fig. 3.10 Simulation results with the comparison of the ideal and the realistic filter ...51

Fig. 3.11 Simulation results of the low-frequency BP filter with four corners ...52

Fig. 3.12 The differential and common mode simulation results front-end circuit of the electrophysiological measurement system ...53

(12)

CHAPTER 1 INTRODUCTION

1.1 BACKGROUND

With the increase of ageing population, chronic diseases, and need for further integration of handicapped through rehabilitation, monitoring, homecare, the development of healthcare and health delivery is constantly progressing.

As the combination of medical care and integrated circuits, the biotelemetry system for monitoring and measuring electrophysiological signals are paid more and more attention nowadays. When a biomedical system is integrated with radio transceivers, we can easily monitor the patient’s physical condition everywhere. So that home nursing and remote medical care will be realized [1]-[2].

In the biomedical measurement system, the measurement and wireless transformation of electrophysiological signals can be realized for a single chip by SoC technology. This technique makes people who need nursing can wear the biomedical measurement system. Therefore we can monitor the electrophysiological signals and transfer the signals to the Remote host anywhere at any time.

During the last few years the research and development on smart wearable for remote health monitoring have been accelerated through consequent public and private financial support. Remote health monitoring could lead to a significant reduction of total cost in healthcare by avoiding unnecessary hospitalizations and ensuring that those who need urgent care get it sooner. To meet the requirements of smart wearable worn

(13)

by the subject, it is necessary to develop the miniaturized, low power, and wireless biomedical sensors.

Portable biomedical sensors of detecting EMG (Electrocardiography), EEG (Electroencephalography), and ECG (Electromyography) signals have been developed and commercialized. However, for implementing the remote health monitoring under the normal quality of life, these biomedical sensors must be wireless link between the subject and the majority of the signal processing system, and be miniaturized via System-on-Chip (SoC) technology.

A biotelemetry system consisting of portable and stationary parts has been developed to monitor and analyze the biomedical signals ECG and EMG [3]. This system was not miniaturized by using SoC technology yet. A particularly strong emerging trend is the move toward SoCs that can provide low-data-rate, short-range wireless communications. Moreover, highly integrated, mixed-signal SoC solutions are ideal for medical OEMs looking to increase the functionality of implantable and portable medical applications.

Because the biomedical measurement system is realized for chip, we can easily combine it with our mobile system, like PDA, notebook, PC, and mobile phones.

Through proper software we can make these consuming IT products become valuable homecare and remote medical apparatus.

Consequently, many researches on the biomedical systems are proposed. Fig.1.1 shows the block diagram of the whole biotelemetry system. In the biotelemetry system, RF signals from the portable subsystem will be transmitted to the stationary system which monitors people’s physical conditions.

(14)

Fig. 1.1 Schematic block diagram of the biotelemetry system

(15)

1.2 REVIEW ON ELECTROPHYSIOLOGICAL SIGNAL MEASUREMENT

SYSTEM

1.2.1 Review on Instrumentation Amplifier Structures (1) Voltage-mode Instrumentation amplifiers

The most often used instrumentation amplifier is constructed of three operational amplifiers and seven resistors showed in Fig. 2.1. The first stage with two opamp op1、

op2 and 、、 is responsible for amplifies input signals 、 .And the voltage gain is

R1 R₂ R_G V_in + V_in −

⎟⎟⎠

⎜⎜ ⎞

⎝

⎛ +

=

G

d R

A 2R¹

1 (1.1)

So we can adjust which is tunable to obtain the gain we want. The function of the second stage with op3 and is to counteracts common-mode signals and add differential signals. So we can obtain good common-mode rejection ratio (CMRR) and take away the noises we don’t want.

RG

6 3 ~ R R

But there exists a disadvantage in this kind of structure. In Fig2.1, we can find the output signal V_out is

⎟⎟⎠

⎜⎜ ⎞

⎝

⎛ +

+ ⋅

×

⎟⎟+

⎠

⎜⎜ ⎞

⎝

⎛−

×

= ₊ ₋

3 4 3 6 5

6 3

4

R R R R R V R R V R

V_out (1.2)

As a result, we need a very good resistors matching of R₃ ~ R₆ to make

5 6 3 4

R R RR = if we want V_out almost equals to zero when V₊ =V₋ ≈V_CM ⋅A_d-- otherwise the CMRR will degrade seriously.

Another problem in this kind of structure is that when we have gain error or the gain is not high enough in op1 and op2, we will have output voltage error. Ref [4]

provides a good method to lower the output voltage error of this kind of structure.

(16)

R

G

R

1

R

6

R

5

R

4

R

3

R

2

V

out

V − + V

in

+ V

in

− V

op3 op1

op2

Fig. 1.2 Conventional Three-op instrumentation amplifier

(2) Current-mode Instrumentation amplifiers

Because of the disadvantage of the conventional voltage-mode instrumentation amplifier, many researches on current-mode instrumentation amplifier comes into existence. In most of the current-mode instrumentation amplifiers, the second generation current conveyor is an important block and the idea of current subtraction is also widely used [5]. We’ll introduce these concepts in the following sections.

⎥ ⎥

⎥

⎦

⎤

⎢ ⎢

⎢

⎣

⎡

⋅

⎥ ⎥

⎥

⎦

⎤

⎢ ⎢

⎢

⎣

⎡

±

=

⎥ ⎥

⎥

⎦

⎤

⎢ ⎢

⎢

⎣

⎡

z x

y

z x y

v i v

i v i

0 1 0

0 0 1

0 0 0

CCII+

Y

X

Z

i

x

i

z

v

x

v

z

v

y

Fig. 1.3 The symbol and the matrix form of second generation current conveyor

(17)

(a) Conventional Current-mode Instrumentation amplifiers

Current conveyor is widely used in many current-mode circuits [6]. The conventional current-mode instrumentation amplifier is also constructed of the second generation current conveyor (CCII). The conventional current-mode instrumentation amplifier is showed in Fig. 1.4. It has the voltage gain , and it doesn’t need a good resistor matching to reach high CMRR. The amplifier’s bandwidth is large with high voltage gain as current conveyors are operating in open loop without the gain-bandwidth product limitation.

1 2/ R R

Unlike the voltage-mode instrumentation amplifier mentioned before, the CMRR of the current –mode instrumentation amplifier is independent of the voltage gain. And due to these characters, the current –mode instrumentation amplifier is good for low-gain and wide-bandwidth applications.

CCII+

Y

X

Z

CCII+

Y X

Z

+

_

A1

A2 A3

R1 in +

V

in − V

R2

i out

Vout

Fig. 1.4 Conventional current-mode instrumentation amplifier

(18)

(b) Current-mode Instrumentation amplifier using CCII with current cancellation

Although the conventional current-mode instrumentation amplifier doesn’t need good resistor matching, its CMRR is not high enough [6]. So we can use current subtraction to improve it. It can effectively improve CMRR of the current-mode instrumentation amplifier. Fig. 1.5 is the current-mode instrumentation amplifier using current cancellation by operational amplifier. Once there is common-mode current flowing to output, it will use to cancel the common-mode signal. And we can adjust the value of to improve the CMRR. But the disadvantage of this structure is that the frequency range of CMRR will be restricted by the operational amplifier. So the frequency range would be limited, and it won’t be such wideband as the structure mentioned before.

i1

i2

R3

CCII+

Y

X

Z

CCII+

Y X

Z

+

_

A1

A2

A3 R1

in + V

in

− V

R2

Vout

i1

Fig. 1.5 Current-mode instrumentation amplifier using current cancellation by an operational amplifier

(19)

(c) Current-mode Instrumentation amplifier using CCII for current inversion

As mentioned in section (2), the structure of current cancellation by an operational amplifier can’t achieve wide bandwidth. This problem can be solved by current inversion method [6]. Current inversion can be realized by inverting the second conveyor output current with an additional positive second-generation current conveyor CCII and summing it then up with the first conveyor output current as presented in Fig.

1.6. The RC network adding to the first current conveyor can make the summing range more accurate. Besides, this network can compensate for the high frequency phase shift of the added conveyor but the resistance can also compensate for the systematic current transfer error, due to the finite input and output impedance of the conveyor.

Comparing to the conventional current-mode instrumentation amplifier, this current inversion structure is also insensitive to the gain error of the CCII.

R3

CCII+

Y

X

Z

CCII+

Y X

Z

A1

R1 A2

in + V

in − V

CCII+

Y X

Z

+

_

A3 R2

i out

Vout

R3

A4 C1

Fig. 1.6 Current-mode instrumentation amplifier using current inversion

Another structure published in [7] uses the similar concept to eliminate the current of common mode. The structure is showed in Fig. 1.7 and the node Z of A1 is

(20)

connected to the node X of A2. It forms a feedback and it minimizes the output current of A1. Therefore the common-mode gain would be compressed.

CCII+

Y

X

Z

CCII+

Y X

Z

A1

A2 R1

in + V

in − V

2i^o

i o

i^o 2i o

RL

Vout

Fig. 1.7 Current-mode instrumentation amplifier

(d) Current-mode Instrumentation amplifier using bias current sensing of opamps

In measuring small biomedical signals, differential amplifiers are often applied to overcome the interference of noises in human bodies. Therefore current-mode instrumentation amplifiers using differential operational amplifiers are also presented in many papers. In Ref [8], the method of bias current sensing is presented and the schematic of this structure is presented in Fig. 1.8. The input stage consists of two identical opamps connected as unity-gain buffer. The outputs of the two opamps are connected via like the conventional three-op INA. The input voltages transfer to current through and the current flows through the bias circuit of op1. After and flow through the two current mirrors and the subtraction of these two branches of current. The current will flow through and forms the output voltage . The function of op3 is to prevent the value of affecting the output impedance of the first stage of op1、op2.

R7

ix R₇ i_x

x BIAS i

I + i_x

ix R₈

Vout R₈

(21)

There are two main advantage of this structure compared to the three-op INA. The first is that no resistor matching is required to achieve high CMRR. The second advantage is that this structure doesn’t require ideal opamps; the design requires only that the opamps identical. But the goal is difficult to attain, but it is easier than having an ideal opamp.

+

_

+

_

R^L Vout in +

V

in − V

VCC

VEE

VCC

VEE

A1

A2

RG

ix

IBIAS x BIAS i I +

x BIAS i I +

IBIAS

x BIAS i I +

ix

Fig. 1.8 Current-mode instrumentation amplifier using bias current sensing of opamps

But there is a serious disadvantage of this kind of structure that we need a precise current mirror to make the current I_BIAS + and i_x subtracted to perfectly.

Once the two current mirrors don’t match well, we will not have a good CMRR. So the design of the current mirror is the main point of this structure to obtain good performance.

IBIAS i_x

Another kind of current-mode instrumentation amplifier structure is also published in [9]. It uses the similar method of current-mode instrumentation amplifier using current inversion we introduced before.

(22)

1.2.2 Review on Bandpass Filter Structures

(1) Comparison on Different Filter Types

From the derivations of mathematical sentences, we can get different transfer functions and responses. There are four essential filter types widely used in the design of filters: Butterworth、Chebyshev、Inverse Chebyshev and Elliptic filter. According to different requirements, we can choose the type we need. Here we’ll introduce these four filter types.

Filter Type

Order Elliptic > Inv-Che > Chebyshev > Butterworth Passband

performance Butterworth > Inv-Che > Chebyshev or Elliptic Group delay Elliptic > Butterworth > Inv-Che > Chebyshev

(2) Design considerations

We have introduced the passive and active filters and analysis the advantages and disadvantages of them. We can choose the type we need from the characteristics from them and the following are the consider factors we must consider when designing a filter:

1. The technology desired for the system implementation.

2. Availability of dc supplies for the active devices, and power consumption.

3. Cost

4. The range of frequency of operation.

5. The sensitivity to parameter changes and stability.

6. Weight and size of the implemented circuit.

7. Noise and dynamic range of the realized filter.

Also, filtering requirements at very high frequencies where ultra fast sampling and digital circuitry may not be realistic and economical may require analog techniques.

We’ll show the choice of filter types as a function of the operating frequency range. Fig.

(23)

1.9 shows the adequacy frequency range of different type filters.

Discrete analog active filters Switched-capacitor active filters

Integrated analog active filters Passive LC filters

Distributed (waveguide filters

10¹ 10² 10³ 10⁴ 10⁵ 10⁶ 10⁷ 10⁸ 10⁹ 10¹⁰ 10¹¹ 1

Frequency,Hz

Fig. 1.9 Choice of filter type as a function of the operating frequency range

1.2.2 Review on Gm amplifiers (1) Different Gm amplifiers

Gm amplifier is a widely used and important analog block in analog circuits like filters and other current-mode circuit. The Gm amplifier transfers voltage to current and is also called the operational transconductance amplifier. The following is the review on some Gm amplifiers.

(a) Nauta’s Gm amplifier

There is a Gm amplifier proposed in [10]-[11] by Nauta and shown in Fig.

1.10.From Fig. 1.10, we can find that the Gm amplifier has no internal nodes, so it can achieve high bandwidth and operate at high frequency filters or robust low frequency filters. The Gm value of the amplifier is shown as below:

( )

_⎟^⎟

⎠

⎞

⎜⎜

⎝

⎛

⎟⎟⎠

⎜⎜ ⎞

⎝ + ⎛

−

=

p p n n p n tp tn dd

m L

W L k W k V V V

G ' ' (1.3)

Although the Gm amplifier has wide bandwidth, it has some disadvantage that its Gm value is decide by MOS parameters and is easily changed by process variation. In Gm-C filters, we need precise Gm value to maintain the performance of the filter. Also the Gm value decided by MOS parameters is fixed and cannot be tuned after been

(24)

manufactured. From (1.3), we can find that if we need large Gm value, we need either large size MOS or to consume more power.

Vdd Vdd

Vdd Vdd Vdd Vdd

Vin+

Vin-

Vout-

Vout+

I-

I+

I3 I4 I5 I6

I1

I2

Fig. 1.10 The Nauta’s Gm amplifier

(b) Rail-to-rail Gm amplifier

Another Gm amplifier proposed in [12] is a rail-to-rail Gm amplifier shown in Fig.

1.11. It can operate with full input swing and used in high input signal applications. Its Gm value is shown as below:

( )( )

'

2 '

' ₁ ₂

p

tn tn

cm na n

m K

V V V V V K

G K − + −

= (1.4)

This rail-to-rail Gm amplifier can have multiple outputs by put parallel output stages as M6 and M6a. However, it has the same problem like Nauta’s Gm amplifier that its Gm value is fixed and will be affected by process variation. Besides, it also consumes large power or occupies large size MOS when large Gm value is going to be realized.

(25)

M2a

M6a

M3a

M1

M5 M5a

M4

M2

M6 M3

M1a

M4a

Vin+ Vin-

V1 V2

VDD

Io2 Io1

Vb

Fig. 1.11 The rail-to-rail Gm amplifier

(c) Conventional Gm amplifier

To solve the problem of Gm value that changed by process variation, the conventional Gm amplifier shown in Fig. 1.12 is widely used [13]. It uses two voltage follow and a resistor to transfer the input voltage into output current we want. From Fig.

1.12, we can find the Gm value is RG

1 . Once we replace by MOS resistor, we can

tune the Gm value by the gate voltage of the MOS resistor. Although we solve the problem of imprecise Gm value caused by process variation, there is another problem of this kind of Gm amplifier.

RG

Because we always put a resistor at the input stage which is a voltage follow, the resistor’s value will affect the gain of the source follow. Once we put a small at the source of the MOS, we will not have a unity gain of source follow.

RG

(26)

M1

M3 V^o1

M2

M4 V^o2

Vⁱⁿ¹ Vⁱⁿ²

R^G

Fig. 1.12 The conventional Gm amplifier

(2) Voltage follower

The most commonly used voltage follower is shown as Fig. 1.13. It is widely used in many operational amplifiers for voltage buffer and input stage for operational transconductance amplifier like Fig. 1.14. This voltage follower is also used in many analog circuits. However, it has a drawback that would influent the performance of the analog circuits. The voltage gain of this circuit is

L m

L m i

o

Z g

Z g V V

≈ +

1 (2.4)

When g_mZ_L is large enough

i o

V

V will approach one. But while the node is

connected with output loads (R or C at high frequency), the term will change and unit gain

Vo

ZL

≈1

i o

V

V will be affected by those loads.

(27)

Vi

Vo

I

^b

Fig. 1.13 Common-drain amplifier (voltage follower)

Vi

V^o

I

^b

RL C_L RI

Fig. 1.14 Common-drain amplifier with output loads

1.3 MOTIVATION

The electrophysiological signals of human body are very small and the frequency is low. The common voltage range is from 50 micro volts for the brain wave (EEG) to 5 mV for some heart voltage (ECG). This rarely goes down to 0.5 micro volts for evoked potential like retina voltage (ENG) or hearing hair cell to the brain steam audio nerve voltage. Besides, the frequency range is usually low, no more than 2 KHz for most cases and seldom extended to 40 KHz.

Besides, there are a lot of low frequency noises in the human body. The frequency of these noises overlaps the frequency range of human body signals. Therefore, the most important part of the front-end portable system is the instrumentation amplifier (INA).

(28)

We use the INA to compress the noises and amplify the electrophysiological signals we need. So we have to design a high common-mode rejection ratio (CMRR) instrumentation amplifier which is the first stage of the portable system. Another point is that once our signal is very small, input offsets voltage and input noise play an important role in our instrumentation amplifier. Once the input offset voltage or input-referred noises become higher than the signals, we will not get any output signals.

Behind the instrumentation amplifier is the low-frequency bandpass filter to filter the signals we want. Human body signals are very low frequency (about 50Hz to 2 kHz), and the bandpass filter needs very large capacitors or resistors. It will increase the difficulty for put the large devices on chip or increase the size of the chip and the costs.

To overcome this problem, we use some method in the thesis without using large devices.

Another problem is about the power of the circuits. Like most of the mobile systems, we need a low power design to make it more convenient to be portable. So the low power is another design consideration.

1.4 THESIS ORGANIZATION

This thesis contains four chapters. Chapter1 introduces the background of nursing system and remote medical care. Besides, the structure and function of the front-end circuit of the electrophysiological signals measurement system are also mentioned. At the end of Chapter1, some architectures of instrumentation amplifier which are already proposed to archive high CMRR will be described. Also the new architecture design of INA and low frequency filter of this thesis are presented. In Chapter2, we’ll describe the circuits to construct the INA and the low frequency bandpass filter.Chapter3 shows the

(29)

simulation results of the front-end circuit of the electrophysiological signals measurement system. Finally, chapter4 describes the conclusion and future work.

(30)

CHAPTER 2 ARCHITECTURE AND CIRCUITS DESIGN

2.1 ARCHITECTURE DESIGN OF THE INSTRUMENTATION AMPLIFIER

2.1.1 Differential Difference operational amplifier (DDA)

Differential difference operational amplifier (DDA) is the extension concept of the differential operational amplifier. Its expression is showed in Fig. 2.1 and the difference between opamp and DDA is that instead of two single-ended inputs as in the case of op-amps, it has two differential input ports

(

V₂ −V₁

)

and

(

V₄ −V₃

)

.The output voltage of the DDA can be written as:

( ) ( )

[

V₄ V₃ V₂ V₁

]

A

V_O = _O − − − (2-1) A is the open-loop gain of the DDA. When a negative feedback is introduced to O

V1 or V4, the basic equation that characterizes the operation of the DDA is obtained as :

3 4 1

2 V V V

V − = − (2-2) when A_O →∞

The DDA is a widely used block in many circuit applications because it has four inputs to have more function changing.

V

1

V

2

V

3

V

4 ⁺^_

V

^O

+ + + _ _ __ +

Fig. 2.1 The symbol for Differential Difference Operational Amplifier

(31)

2.1.2 Differential Difference operational transconductance amplifier (DDGm)

Like the differential difference operational amplifier which is extended from operational amplifier, differential difference operational transconductance amplifier (DDGm) is extended from operational transconductance amplifier. Fig. 2.2 is the expression of DDGm and its function is:

( ) ( )

[

V₄ V₃ V₂ V₁

]

G

I_O = _m − − − (2-3) G is the transconductance m

V

1

V

2

V

4

V

3

I

o

Fig. 2.2 The symbol for Differential Difference Operational Transconductance Amplifier

When the output has a resistor loading, we can obtain the same function (2-2).

The differential difference operational transconductance amplifier also has multiple functions and we’ll use the DDGm in our new structure of instrumentation amplifier.

2.1.3 New structure design of the instrumentation amplifier

We have introduced some structures of instrumentation amplifiers in chapter1;

most of the structures transfer the voltages of input signals into current in the first stage.

Then they amplify the current and subtract the current in the second stage. If we can do

(32)

the action of current transformation and the subtraction of the input signals simultaneously at the first stage then amplify the signals at the second stage, the common mode signals will be compressed twice. Fig. 2.3 shows the structure of this idea.

Subtraction of input signals

Gain Stage Y

1

Y

2

X

1

X

2

Z

Y

1

Y

₂

^X

₁

( V

_cm

) Y

1

Y

₂

X

₂

( 2 V

_d

)

Fig. 2.3 Expression for the new approach of instrumentation amplifier

As showed in Fig. 2.3, we subtract input signals in the first stage and the noises (common-mode signals) will almost equals to zero with differential signals become double. As a result, before entering the gain stage, the noises have been eliminated and the gain stage also has the ability to compress common-mode signals. Under this structure, common mode rejection ratio would be high.

The function of the idea for first stage can be realized by our new circuit – differential difference operational transconductance amplifier. Once we connect the differential difference operational transconductance amplifier like Fig. 2.4, we can get the output signal as:

( ) ⁽ ⁾

[ ]

(

m a

)

m

(

ref in

)

o

in m ref m o m a

o

o ref m o

V V

G R G V

V G V G V R G

V

V V V V G I

∆

−

= +

⇒

∆

− +

−

=

⇒

−

=

1

2 1

(33)

( ) (

in ref

)

ref a

m a m o

V V

V V R V

G R V G

−

∆

≈

−

− −

= ₁ ₂

1

(2.3)

And we can get a voltage subtraction function using this method to cancel the common-mode signals.

V

ref

V

ref

V

1

V

2

V

^o

DDGm

R

a

Fig. 2.4 Subtraction of input signals using two DDGm

After using the function showed in equation (2.3), we can design a new structure of high CMRR instrumentation amplifier. Fig. 2.5 is the structure of the new high CMRR INA using DDGm. The main function of the first stage is to cancel the common-mode and doubles the differential signal like Fig. 2.6 showed. Briefly speaking, we can say that the first stage constructed with two DDGm is to enhance the differential signal and to compress the common-mode signals (noises). Then the second stage with Gm amplifier and is to provide the gain for signals. Equation (2.4) derives the node voltage of Vx1 and Vx2.

RG

(34)

X1

V

o

+ _

2

V

X

Vⁱ¹ Vⁱ²

V

ref

V

ref

DDGm1

DDGm2

Gm

R

a

R

G

Opamp

1st stage 2nd stage

Fig. 2.5 The new structure of high CMRR instrumentation amplifier

Differential Input

Common Input

+ _

1

V

X

2

V

X

1

V

X

2

V

X

Gm

V

o

V

o

Fig. 2.6 The function of the first stage and waveform of V_X₁ and V_X₂

dm dm

i dm i dm

X V V V

V ₁_, = ₁_, − ₂_, =2

dm dm

i dm i dm

X V V V

V ₂_, = ₂_, − ₁_, =−2

, 0

1 , 2 ,

2_cm = _i _cm− _i _cm ≈

X V V

V

, 0

2 , 1 ,

1_cm = _i _cm− _i _cm ≈

X V V

V (2.4)

(35)

2.2 CIRCUIT DESIGN OF THE DIFFERENTIAL DIFFERENCE OPERATIONAL TRANSCONDUCTANCE AMPLIFIER

2.2.1 The Flipped Voltage Follower (FVF)

The most commonly used voltage follower is shown as Fig. 1.13. It is widely used in many operational amplifiers for voltage buffer and input stage for operational transconductance amplifier like Fig. 1.12. This voltage follower is also used in many analog circuits. However, it has a drawback that would influent the performance of the analog circuits.

To solve this problem, the paper [14] claimed a new voltage follower circuit called flipped voltage follower (FVF) shown in Fig. 2.7. In Fig. 2.7 the current through M1 is held constant and independent on the output current. It could be described as a voltage follower with shunt feedback. Neglecting body effect and the short-channel effect,

is held constant, and voltage gain is unity. Unlike the conventional voltage follower, FVF is able to source a large amount of current, but its sinking capability is limited by the biasing current . The large sourcing capability is due to the low impedance at the output node, which is approximately

1

VSGM

Ib

1 2 1

1

o m m

o g g r

r = , where and

are the transconductance and output resistance of transistor, respectively. This value is in the order of 20-100 .

gmi

roi

Ω

And we will use this circuit to construct our differential difference operational transconductance amplifier.

(36)

V^o Vⁱ

I

^b

M1 M2

Fig. 2.7 Flipped Voltage Follower

(37)

2.2.2 The differential difference operational transconductance amplifier

Using the flipped voltage follower (FVF), we can design a DDGm circuit shown in Fig. 3.8. The magnitude of Gm can be derived as follow:

, 2 2

4 3 2 2

1 1

i i X i

i X

V V V

V

V V +

+ =

=

( )

2

4 3 2 1 2 1

i i i i X X

V V V V V

V + − +

=

−

⇒

( ) ( )

m i i i i

o R

V V V I V

2

2 4 3

1− − −

=

⇒

m

m R

G 2

= 1

⇒ (2.4)

Vbias1

Vbias2 Vbias2

Vi4 Vi3

Vi2 Vi1

M1 M2

M3 M4 M5 M6

M7 M8

M9 M13

M12

M10 M11

M15

M14 M16

Vout

Vx1 Vx2

R^m

Io

Fig. 2.8 The DDGm amplifier using FVF method

In Fig. 2.8, are the cascode current mirror. The stacked MOS can decrease the channel length effect of and enhance the accuracy of the mirrored current

16

~ 13 M M

(38)

ratio. On the other hand, the output resistance of an ideal transconductance amplifier is infinite, so the stacked MOS M9 M~ 16 can increase of the DDGm circuit. r_o

MOS size m

M1、M2 (0.5/10) 10

M3、M4、M5、 M6 (0.5/10) 1

M7、M8 (0.7/10) 1

M9、M10 (0.4/9) 1

M11、M12 (0.4/4) 1

M13、M14 (1/4.2) 3

M15、M16 (1/4) 5

Table 2.1 The size of the DDGm amplifier in Fig. 2.10

2.3 STRUCTURE DESIGN OF THE LOW-FREQUENCY BANDPASS FILTER 2.3.1 Specification of the low-frequency bandpass filter

For the electrophysiological measurement system, we can set the spec of the low-frequency bandpass filter as Table 2.2.

From the Table 2.2, we can find the frequency range is from 50 Hz to 2 kHz, the stopband attenuation is about 35 to 40 dB, and the passband ripple is from 3 to 8dB.

From the spec we set, we can use the Inverse Chebyshev or the Elliptic filter to realize it.

By the comparison of the two LC networks of Inverse Chebyshev and the Elliptic filter we derived, we can find that the element values of Elliptic filter is much more realizable than the f Inverse Chebyshev filter. So finally, we choose the 5th-order Elliptic filter structure to design our desired low-frequency bandpass filter.

(39)

Filter Spec

Low corner frequency 50 Hz High corner frequency 2 kHz

Stop band ratio 1.98

Stop band attenuation 35~40 dB

Pass Band Ripple 3~10 dB

Gain >0dB Type of the Filter 5th order Elliptic

Table 2.2 The specification of the low-frequency bandpass filter

2.3.2 LC network and the leapfrog structure

From the spec of Table 2.1, we can derive the transfer function of the low-frequency bandpass filter showed in equation (2.5). And from equation (2.5),we can derive the LC network in Fig. 2.9. In Fig. 2.9, we can find that the element values of the capacitors won’t be able to be realized on the integrated circuits. So we have to scale the element values of Fig. 2.9 into the values that can be made on chip reasonable. Fig.

2.9(b) is the circuit after normalizing the element values from Fig. 2.9 (a); we make all the capacitor values from 0.1pF to 10pF using the normalization method of equation (2.6) and (2.7).

( ) ( )

( )

⁸ ⁴

( )

¹¹ ³

( )

¹⁴ ²

( )

¹⁷

( )

¹⁹

5 6

3 16 5 11

151 . 6 128

. 1 095

. 6 702

. 5 544

. 1 7241

055 . 1 184

. 5 677

+

⋅ +

⋅

S S

S (2.5)

L

S R

R C C

= ⋅

' (2.6)

L

S R

R L

L'= ⋅ ⋅ (2.7) As we mentioned before, we’ll use the leapfrog structure to decrease the effects of

新型低功率與高雜訊抑制電生理訊號量測系統之前端 電路設計與分析