Chapter 1 Introduction
1.3 Introduce to The Standards on Wireless Communication System
1.3.1 BlueTooth
In early 1998, a consortium of companies including Ericsson, IBM, Intel, Nokia, and Toshiba formed a special interest group, codenamed
"Bluetooth". The group's goal was to develop a low-cost, flexible wireless platform for short-distance communication (< ~10 meters). The Bluetooth 1.0 specifications were released on July 26, 1999, but the technology has only recently become cheap enough for widespread use. The cost of a Bluetooth radio chip has dropped from $20 and is now approximately $5.
Spectrum is divided up into 79 channels spaced 1 MHz apart. Data is transmitted at 1 Mbps. For security benefits and noise reduction, a Bluetooth transmitter employs frequency hopping, switching channels up to 1600 times a second.
Bluetooth is capable of point-to-point or point-to-multipoint communication. This flexibility allows Bluetooth to be used in a wide variety of applications. Because power consumption is always a concern for mobile devices, Bluetooth has three power classes that can be used depending on how far apart the communicating devices are from one another.
In 2002 Ericsson's Bluetooth technology had finally won a standard with the IEEE global standards body, a much needed shot in the arm for the fledgling wireless Personal Area Network (PAN) technology. The standard, 802.15.1, will lend validity to Bluetooth devices, and enable vendors to better support the hardware and software involved. Bluetooth devices based on this standard will suffer fewer compatibility issues than current implementations.
The IEEE licenses part of the current standard, authored by the Bluetooth SIG, as a basis for its 802.15 standard. As a result, 802.15 devices will be fully compatible with Bluetooth v1.1 devices.
Over the next few years, Bluetooth's use is expected to significantly grow. The specifications for Bluetooth 2.0 had been finalized for a couple years. Bluetooth 2.0 had been designed to complement existing Bluetooth devices and will offer data transmission rates up to 12 Mbps.
Chapter 2 The Principal Concepts of Designing Low Noise
Amplifier and Mixer
In this chapter, the principal concepts on designing Low Noise Amplifier and Mixer will be introduced. Since RF receiving front-end circuits usually get extremely weak signals from free space. The extremely weak signals are susceptible to noise, and always greatly affect overall performance. Therefore, before we begin to design RF receiving front-end circuits, the most important thing for us is to realize why noise is generated, and how to reduce the effect of noise. Besides, while designing each block such as LNA, mixer, we should have system view for acquiring the best overall performance. Hence, some design considerations and characteristics will be carefully taken into account in this chapter.
2.1 Introduction Noise Sources
In this subsection, only the intrinsic noises will be introduced. They are caused by small current and voltage fluctuations produced within devices themselves. The extraneous man-made signals that could be a problem in high-gain circuit will be excluded. The existence of noise is basically due to the fact that electrical charge is not continuous, and the
discrete amount is equivalent to electron charge.
The study of noise is important because it represents a lower limit to the size of electric signal that can be amplified by a circuit without significant deterioration in signal quality. Noise also results in an upper limit to the useful gain of amplifier, because if the gain is increased without limit, the output stage of the circuit eventually begins to enter saturated region.
2.1.1 Shot Noise
Shot noise is always taken place in diodes, bipolar transistors and MOSFETs, and has relations with the conduct current on them. The origin of shot noise can be seen by considering the carrier concentrations in a diode biased in forward region. An electrical file ξ exists in the depletion region and a voltage (ψ0 – V) exists between the p-type and n-type regions, whereψ0 is the build-in potential and V is the forward bias on the diode. The forward current of the diode I is composed of holes from the p region and electrons from n region, which have sufficient energy to overcome the potential barrier at the junction. Once the carriers have crossed the junction, they diffuse away as minority carriers.
The passage of each carrier across the junction, which can be modeled as a random event, is dependent on the carrier having sufficient energy and a velocity directed toward the junction. Thus external current I, which appears to be a steady current, contains a large number of random independent current pulses. If the current is examined on a sensitive oscilloscope, the trace appears as Fig. 2.1, where ID is the average current.
Fig. 2.1 Diode current I as a function of time
The fluctuation in I is termed shot noise and is generally specified in terms of its mean-square variation about average value. This is written as i2, where
(2.1)
It can be shown that if a current I is composed of a series of random independent pulses with average value ID, then the resulting noise current has a mean-square value
(2.2)
Where q is the electronic charge ( C) and is the bandwidth in hertz. This equation shows that the noise current has a
mean-square value that is directly proportional to the bandwidth x (in hertz) of the measurement. Thus a noise-current spectral density x (with units square amperes per hertz) can be defined that is
constant as a function of frequency.
2.1.2 Thermal Noise
The mechanism producing thermal noise is totally different from ID
t
Diode Current I
shot noise. In conventional resistors it is due to the random thermal motion of the electrons and is unaffected by the presence or absence of direct current, since typical electron drift velocities in a conductor are much less than electron thermal velocities. Since this source of noise is due to the thermal motion of electrons, we expect that it is related to absolute temperature T. In fact thermal noise is directly proportional to T (unlike shot noise, which is independent of T), as T approaches zero, thermal noise approaches zero.
In a resistor R, thermal noise can be shown to be represented by series voltage generator as shown in Fig. 2.2a, or by a shunt current generator as in Fig. 2.2b. These representations are equivalent and
(2.3)
(2.4)
Where k is Boltzmann’s constant. At room temperature V-C. Equation 2.3 and 2.4 show that the noise spectral density is again independent of frequency and, for thermal noise, this is true up to 1013 Hz. Thus thermal noise is another source of white noise. Note that the Norton equivalent of 2.4 can be derived from 2.3 as
(2.5)
A useful number to remember for thermal noise is that at room temperature (300°K), the thermal noise spectral density in a 1-KΩ resistor is V2/Hz. Another useful equivalence is that the thermal noise-current generator of a 1-K Ω resistor at room temperature is the same as that of 50μA of direct current exhibiting shot noise.
Thermal noise as described above is a fundamental physical phenomenon and is present in any linear passive resistor. This includes conventional resistors and the radiation resistance of antennas, loudspeakers, and microphones. In the case of loudspeakers and microphones, the source of noise is the thermal motion of the air
molecules.In the case of antennas, the source of noise is the black-body radiation of the object at which the antenna is directed. In all cases, (2.3) and (2.4) give the mean-square value of the noise.
Fig. 2.2 Alternative of Thermal Noise
2.1.3 Flicker Noise
Flicker noise in one of noise found in all active device, as well as in some discrete passive elements such as carbon resistors. The origins of flicker noise are varied, but it is caused mainly by traps associated with contamination and crystal defects. These traps capture and release carriers in a random fashion and the time constants associated with the process give rise to a noise signal with energy concentrated at low frequencies.
Flicker noise, which is always associated with a flow of direct current, displays a spectral density of the form
(2.6) where
= small bandwidth at frequency f I = direct current
K1 = constant for a particular device a = constant in the range 0.5 to 2 b = constant of about unity
If b = 1 in (2.6), the noise spectral density has a 1/f frequency dependence (hence the alternative name 1/f noise), as shown in Fig. 2.3.
It is apparent that flicker noise is most significant at low frequencies,
although in devices exhibiting high flicker noise levels, this noise source may dominate the device noise at frequency well into the megahertz range.
Fig. 2.3 Flicker Noise spectral density versus frequency
It was noted above that flicker noise only exists in association with a direct current. Thus in the case of carbon resistors, no flicker noise is present until a direct current is passed through the resistor (however, thermal noise always exists in the resistor and is unaffected by any direct current as long as the temperature remains constant). Consequently, carbon resistors can be used if required as external elements in low-noise, low-frequency integrated circuits as long as they carry no direct current.
If the external resistors for such circuits must carry direct current, however, metal film resistors that have no flicker noise should be used.
The final characteristic of flicker noise that is of interest is its amplitude distribution, which is often non-Gaussian.
2.1.4 Burst Noise
Burst noise is another type of low-frequency noise found in some integrated circuits and discrete transistors. The source of this noise is not fully understood, although it has been shown to be related to the presence of heavy-metal ion contamination. Gold-doped device show very high levels of burst noise.
Burst noise is so named because an oscilloscope trace of this type Log scale f
1/f
Log scale
of noise shows burst of noise on a number (two or more) of discrete levels. The repetition rate of the noise pulses is usually in the audio frequency range (a few kilohertz or less) and produces a popping sound when played through a loudspeaker. This has led to the name popcorn noise for this phenomenon.
The spectral density of burst noise can be shown to be the form
(2.7) that is characteristic of burst noise. At higher frequencies the noise spectrum falls as 1/f 2. Burst noise processes often occur with multiple time constants, and this gives rise to multiple humps in the spectrum.
Also flicker noise in invariably present as well so that the composite low-frequency noise spectrum often appear as in Fig. 2.5. As with flicker noise, factor K2 for burst noise varies considerably and must be determined. The amplitude distribution of the noise is also non-Gaussian.
Fig. 2.4 Burst Noise spectral density versus frequency f
Log scale 1/f
Log scale
fc 1/f 2
Fig. 2.5 Spectral density of combined multiple burst noise sources and flicker noise.
2.1.5 Avalanche Noise
Avalanche noise is a form of noise produced by Zener or avalanche breakdown in a pn junction. In avalanche breakdown, holes and electrons in the depletion region of a reverse-biased pn junction acquire sufficient energy to create hole-electron pairs by colliding with silicon atoms. This process is cumulative, resulting in the production of a random series of large noise spikes. The noise is always associated with a direct-current flow, and the noise produced is much greater than shot noise in the same current, as given by (2.2). This is because a single carrier can start an avalanching process that results in the production of a current burst containing many carriers moving together. The total noise is the sum of a number of random bursts of this type.
The most common situation where avalanche noise is a problem occurs when Zener diodes are used in the circuit. These devices display avalanche noise and are generally avoided in low-noise circuits. If Zener diodes are present, the noise representation of Fig. 2.6 can be used, where the noise is represented by a series voltage generator v2. The dc voltage Vz is the breakdown voltage of the diode, and the series resistance R is
f Log scale
Log scale Burst noise humps
typically 10 to 100Ω. The magnitude of is difficult to predict as it depends on the device structure and the uniformity of the silicon crystal, but a typical measured value is V2/Hz at a dc Zener current of 0.5 mA. Note that this is equivalent to the thermal noise voltage in a 600-kΩ resistor and completely overwhelms thermal noise in R. The spectral density of the noise is approximately flat, but the amplitude distribution is generally non-Gaussian.
Fig. 2.6 Equivalent circuit of a Zener diode including noise
2.2 The Principal Concepts of Low Noise Amplifier
In a classical radio receiver, low noise amplifier is the most significant component, owing to it dominates the sensitivity of overall system [12]. The principal concepts of designing low noise amplifier are to compromise among input impedance, noise figure, power gain, power consumptions and linearity [11]. However, there are usually some tradeoffs among them, and there is almost not one circuit achieving all goals simultaneously, especially in ultra-wide band design. Besides, different process technology will influence the difficulty of goals’
achievement. InP-BASED high electron-mobility transistor (HEMTs) provides an outstanding low-noise performance and superior high-frequency performance [18]. Moreover, HEMTs have excellent low-temperature performance, and do not have the carrier freeze-out effect even at temperature as low as 15K. However, compared with silicon technology, HEMT technology is very expensive. It is not suitable for commercial products. CMOS technology offers advantages such as low cost, mature process, good thermal conductivity, and large scale integration. However, CMOS technology suffers from high noise figure.
There are a huge number of papers published regarding low noise amplifier design. They were applying various structures for different applications. The resistive feedback amplifier [19] could easily achieve input impedance matching, yet it suffers from noise figure deterioration problem. Moreover, it usually limits input match at higher frequency due to the parasitic input capacitance [20]. The series feedback with inductive source degeneration [21-22] offers good input impedance with sufficient low noise figure, yet it is laborious for ultra-wide band design. The low noise amplifier employing an input three-section band-pass Chbyshev filter [20] can acquire wide band input impedance and low noise figure as well. However, the capacitance Cp between the gate and the source of the input device should be chosen considering the compromise between the size of Ls and the available power gain, while large Cp leads to the gain reduction due to the degradation of the effective cutoff frequency.
Below are three popular low noise amplifier structures which are widespread used in numerous products.
2.2.1 Classical Noise Matching Technique
Classical Noise Matching (CNM) Technique is used to acquiring minimum NF, Fmin, by presenting the optimum noise impedance Zopt to the given amplifier. We usually implement this technique by placing a matching circuit between the source and input of the amplifier. By applying this technique, the LNA can be designed to achieve an NF equal to Fmin of the transistor, the lowest NF that can be obtained with the given technology. However, due to the inherent mismatch between Zopt and Z*in (where Z*in is the complex conjugate of the amplifier input impedance), the amplifier can experience a significant gain mismatch at the input.
Therefore, the CNM technique typically requires compromise between the gain and noise performance.
Fig. 2.7(a) shows a cascade-type LNA topology, which is one of the most popular topology due to its wide bandwidth, high gain, and high reverse isolation. In the given example, the selection of the cascade topology simplifies the analysis, and the gate-drain capacitance can be neglected.
Fig. 2.7(b) shows the simplified small-signal equivalent circuit of the cascade amplifier for the noise analysis including the intrinsic transistor noise model. In Fig. 2.7(b), the effects of the common-gate transistor M2 on the noise and frequency response are neglected, as well as the parasitic resistances of gate, body, source, and drain terminal.
In Fig 2.7(b), represents the mean-squared channel thermal noise current, which is given by
(2.8)
Where is the drain-source conductance at zero drain-source voltage VDS, k is the Boltzmann constant, T is the absolute temperature, and xx is the bandwidth, respectively. The parameter γ has a value of unity at
zero VDS and 2/3 in saturation mode operation with long channel devices.
The value of γ increase at high VGS and VDS and can be more than two in short-channel devices.
Fig. 2.7 (a)Schematic of a cascade LNA topology adopted to apply the CNM technique. (b) Its small-signal equivalent circuit.
The fluctuating channel potential due to channel noise current shown in (2.8) couples capacitively into the gate terminal, leading to a noisy gate current. The mean-squared gate-induced noise current is given by
(2.9)
where
(2.10)
In (2.9), δis a constant with value of 4/3 in long-channel device,
and Cgs represents the gate-source capacitance of the input transistor. Like γ, the value of δ also increases in short-channel devices and at high VGS and VDS . Since the gate-induced noise current has a correlation with the channel noise current, a correlation coefficient is defined as follows:
(2.11)
After some lengthy algebraic derivations, the noise parameters for the cascode amplifier shown in Fig. 2.7(1) can be expressed as
(2.12)
(2.13)
(2.14)
where represents the noise resistance, is the optimum noise admittance, and is the minimum noise factor, respectively.
Note that, from Fig. 2.7(b), the input admittance is purely capacitive, i.e., . By comparing the complex conjugate of with (2.13), it can be seen that the optimum source admittance for input matching is inherently different from that of the noise matching in both real and imaginary part. Thus, with the given example, one cannot obtain input matching and minimum NF simultaneously. This is the main limitation of the CNM technique when applied to the LNA topology shown in Fig. 2.7(a).
2.2.2 Simultaneous Noise and Input Matching Technique
While designing low noise amplifier, feedback techniques are always implemented in order to shift the optimum noise impedance Zopt to the desired point. Shunt feedback has been applied for wide-band [23]
and better input/output matching. Series feedback has been preferred to
obtain SNIM without the degradation of the NF. The series feedback with inductive source degeneration, which is applied to the common-source or cascode topology, is especially widely used for narrow-band applications.
Fig. 2.8(a) and (b) shows a cascade LNA with inductive source degeneration and the simplified small-signal equivalent circuit.
Fig. 2.8 (a) Schematic of a cascode LNA topology adopted to apply the SNIM technique. (b) Its small-signal equivalent circuit.
In Fig. 2.8(b), the same simplifications are applied as in Fig. 2.7(b).
The following are the ways to obtain the noise parameter expressions of a MOSFET with series feedback: noise transformation formula using noise parameters, using the noise matrix, or Kirchoff’s current las/Kirchoff’s voltage law (KCL/KVL) with noise current sources. As in (2.12)-(2.14), the noise parameters seen in the gate of the circuit shown in Fig. 2.8(b)
can be obtained. The derivation is somewhat tedious, but the result is simple enough to provide useful insights. The noise factor and noise parameters can be given by
(2.15) (2.16) (2.17) (2.18)
In (2.16)-(2.18), the noise parameters with superscripted zeros are those of the cascode amplifier with no degeneration [see (2.12)-(2.14)].
Note that (2.17) is expressed in impedance, as it is simpler in this case, and is given by
(2.19)
Note that, from (2.16)-(2.18), only Zopt is shifted and there is no change in Rn and Fmin. Also, note that (2.16)-(2.18) are valid for any arbitrary matching circuits, as well as the source impedance Rs in Fig. 2.8.
In addition, as shown in Fig. 2.8(b), the input impedance Zin of the given LNA can be expressed as
(2.20)
As can be seen from (2.20), the source degeneration generates the
real part at the input impedance. This is important because there is no real
real part at the input impedance. This is important because there is no real