Chapter 3 The Implementation of 4GHz to 20GHz Designing Low
3.8 Conclusion For Wide-Band LNA
Fig. 3.19 The measured forward transmission coefficient (S21) and
reverse transmission coefficient (S12)
The measured forward transmission coefficient (S21) and reverse transmission coefficient (S12) are shown in Fig. 3.19. Similarly, due to the restriction of instruments, we fed the port 1 of instrument into the output of device and port 2 into the input of device. Therefore, the S21 on Fig.
3.19 actually is reverse transmission coefficient (S12), and S12 is the forward transmission coefficient (S21). The measured forward transmission coefficient (S21) is around -22dB to -16dB. Compared with simulated result, there are a lot of differences.
3.8 Conclusion For Wide-Band LNA
The chip malfunctions, and we have some ideas which might be the root causes.
A. The issues on layout
In this project most components are placed on metal 6, and metal 2 to metal 5 are placed on ground plane. Owing to there are 0.8μm gap
between metal 5 and metal 6, it will result in much effects of parasitic capacitor under many transmission lines which are utilized to imitate inductors. It makes the character of whole circuit greatly change.
Therefore most places on metal 2 to metal 5 should be reserved.
B. The issues on the settings of ADS
While simulating on ADS, some parameters about substrate are not correctly set, such as the height of substrate. Owing to metal 5 is placed on ground plane, the correct height of substrate is 0.8μm. Besides, the dielectric constant is set wrong, too. Though, it won’t greatly change input reflection coefficient, it will make power gain not to be so flat any more under intended frequency range. After changing wrong parameters to correct ones and simulating again, we found the simulated results are very like to measured results. The simulated results are shown Fig. 3.20.
C. The issues on components placement
In our design, we utilize many components provided by wafer company. The places under inductor, between metal 2 to metal 5, shouldn’t be placed ground plane. It doesn’t make the characters of components consist with the ones provided by wafer company. Besides, after completing simulation on ADS, it is essential to run Momentum to verify the characters of inductors and transmission lines. The works on running Momentum are omitted.
5 10 15 20 25 30 35
Fig. 3.20 The simulated input reflection coefficient (S11) and output reflection coefficient (S12) after changing wrong parameters to correct
ones
D. The issues on transmission lines
The places under inductor, between metal 2 to metal 5, shouldn’t be placed ground plane, and we should place ground place on right metal refer to the height of substrate. Besides, we should run Momentum to verify the characters of transmission lines.
E. We should consider testing environments before designing circuit
While designing circuit, we place two 300μm bond wires at input and output terminals. However, while measuring, there is no bond wire in the circuit. Though it won’t change the character a lot, we should consider testing environments before designing circuit.
Chapter 4 The Design of Novel Mixer
4.1 Induction
The intermodulation performance of a receiver front end is often limited by that of the mixer. This is because the mixer performance is usually worse than that of the other stages, and the mixer must handle the largest signal levels. Consequently, in most low-noise capability can do much to improve dynamic range.
The most commonly used mixers in microwave systems employ Schottky-barrier diodes as the mixing elements. These are usually used in balanced structures to separate the RF and local oscillator (LO) signals, to improve large-signal capability, and to reject certain even-order spurious responses and intermodulation products. Because the Schottky diode is very strongly nonlinear, diode mixers have at best mediocre intermodulation susceptibility.
Nowadays the most popular used topology of mixer is Gilbert Cell.
Many papers discussing Gilbert Cell are published on international journals. In the chapter, we propose a novel mixer which is never published in worldwide international papers.
4.2 Operating Principle
Mixers are conventionally realized by applying a large LO signal and a small RF signal to a nonlinear device, usually a Schottky-barrier diode. The LO modulates the junction conductance at the LO frequency, allowing frequency conversion. In principle, this conductance could be
realized via a time-varying linear conductance, rather than a nonlinear one, resulting in a mixer without intermodulation. A simple examples of such a time-varying linear element, which is capable of intermodulation-free mixing, is an ideal switch, operated at the LO frequency, in series with a small resistor.
Fig. 4.1 shows a conventional very low intermodulation mixer(VLIM). To realize a mixer, the MOS is operated in common-source configuration, the LO is applied to the gate, with proper DC bias, and the RF is applied to the drain. The IF is filtered from the drain. The relatively large value of Cgd would couple the RF and LO circuits to an unacceptable degree, so for a single-device mixer, RF and LO filters must be used. It is important that the LO voltage not be coupled to the drain terminal; if it is, the drain voltage will traverse the more strongly nonlinear portion of the V/I curve, increasing the IM level.
The RF filter should therefore be designed to short-circuit the drain at the LO frequency. The design goal for the LO filter is not so clear. If RF voltage is coupled to the gate, it is conceivable that intermodulation could be increased because of the nonlinearities in Gm. If the gate is shorted at the RF frequency, no RF voltage appears on the gate, so there is no possibility of IM generation in this way. However, open-circuiting the gate effectively halves the capacitance in parallel with the channel resistance, so conversion loss should be lower. In the mixer described here, the LO filter was designed to short-circuit the RF at the gate.
Fig. 4.1 Conventional very low intermodulation mixer
m1freq=
dBm(Vif)=-58.7631.500GHz
5 10 15 20 25 30 35 40 45
0 50
-130 -110 -90 -70 -50 -30
-150 -10
freq, GHz
dBm(Vif)
m1
Fig. 4.2 The simulated spectrum of conventional VLIM when FLO is 8.8GHz and FRF is 10.3GHz without LO and RF filters
Conventional VLIM has many advantages. The most one is simple, and the others are no DC power consumption, no 1-dB compression point since it is a passive circuit. However, the LO and RF filters are essential for reducing intermodulation, such as shown in Fig. 4.2.
Fig. 4.3 shows a novel mixer, double balance very low
intermodulation mixer. Compared with conventional VLIM, its advantage is that LO and RF filters are not needed. The LO and RF signals would be eliminated by the symmetrical architecture.
Fig. 4.3 Double balance very low intermodulation mixer
4.3 Simulated Results
Fig. 4.5 Conversion gain v.s. RF frequency when IF is 150MHz
-70 -60 -50 -40 -30 -20 -10 0
-80 10
-60 -40 -20 0 20
-80 40
Power_RF
ConverGain
Fig. 4.6 Conversion gain v.s. RF power when IF is 150MHz
-70 -60 -50 -40 -30 -20 -10 0
-80 10
-60 -40 -20 0
-80 20
Power_RF
PowerIF_real
Fig. 4.7 IF power v.s. RF power when IF is 150MHz
4.4 Conclusions For Double Balance VLIM mixer
A novel architecture of mixer is proposed. Its advantages are no DC power consumption, no RF and LO filters, and no 1-dB compression point since it is a passive mixer.
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