Chapter 4 A 3 ~ 8GHz Direct Conversion Broadband
4.3.3 On Board PCB Layout Consideration
Because of our broadband mixer design is at high frequency range, we must carefully layout our PCB board. The PCB layout diagram is shown Fig. 4.23. First, the transmission line’s distance between the series inductor and series capacitor (series resonance) can’t be long. In this layout, we just reserve two components’ pad. This is because long distance transmis would influence our input broadband matching. Second, the parallel inductor and capacitor (parallel resonance) would be layout into the signal path, otherwise, it could causes the loading effect. Then, we put the bypass capacitor 0.1uF near the tank at the bias voltage Vg. It let the ac signal ground. The total length of transmission line between two series resonance can’t exceed 5mm.
ference of Anaren Balun 30057
sion line
Fig. 4.23 On board PCB layout diagram
.3.4 Measurement and Simulation Result
asured S-parameter is plotted in
measured result. The solid plot is the simulation result. This mixer’s andwidth is about 3GHz from 2 to 5 GHz. The IIP3 measured result is shown in Fig.
.26 and Fig. 4.27. The simulation noise figure is shown in Fig. 4.27. The minimum
4
Measurement is used on-board probing. The me
Fig. 4.24, together with simulation results for comparison. The circuit plot is the measured result. The solid line is the simulation result. The triangle plot is the measured result with no balun connection. The low frequency is matched the simulation result, but high frequency is not. The cause is the transmission line performance is not good at high frequency. The board we use is RO4003C. The Fig. 4.28 is the measured result of various transmission line width of RO4003C. In these curves, we can observe the transmission line’s return loss is poor in 3 ~ 6 GHz.
The measured maximum conversion gain is 7.4dB at 3-MHz IF band. The Fig.
4.25 is plotted the measured result, together with simulation result for comparison. The circle plot is the
b 4
noise figure is 7.1dB. The total power of this broadband mixer is 11.8mW with a power supply 1.6V. The table 4.3 is the comparison of simulation and measured result.
Simulation result
Fig. 4.24 S11 Simulation and measured result of broadband mixer
simulation result
Fig. 4.25 Conversion gain simulation and measured result of broadband mixer
IIP3=1.23dBm 3GHz two tone test
-18 -13 -8 -3
-23 2
-70 -60 -50 -40 -30 -20 -10 0
-80 10
Input Power(dBm)
Output Power(dBm)
Fig. 4.26 Linearity measured result at 3GHz of 3~8GHz mixer
4 5 6 7
3 8
0 5
-5 10
Freq (GHz)
IIP3 (dBm)
Fig. 4.27 Linearity measured result versus frequency of broadband mixer
2 4 6 8
0 10
10
5 15
Freq (GHz)
Noise Figure (dB)
Fig. 4.28 Simulation result of Noise Figure of broadband mixer
W=25 mil W=26 mil W=27 mil W=28 mil
W
2 4 6 8
0 10
-40 -30 -20 -10
-50 0
Freq (GHz)
dB(S(1,1))
Fig. 4.29 Various width of transmission line of RO4003C
TABLE 4.3 Comparison of simulation and measured performance Simulation Result Measured Result
Bandwidth 3 ~ 8 GHz 2 ~ 5 GHz
According to previous section, we observe the worse return loss and conversion aybe the transmission line and bond-wire’s variation. Then, we use ADS Momentum to simulation the transmission line of RO4003C’s board effect.
The Fig. 4.30 and 4.31 are the simulation results of considering the board effect and bond-wire variation. The cross plot is the bond-wire inductor which value is 1nH. The triangle plot is the bond-wire inductor which value is 0.5nH. The diamond plot is the bond-wire inductor which value is 1.5nH. In Fig. 4.30, the measured result is similar to the bond-wire inductor which value is 0.5nH. Relatively, in Fig. 4.31, the conversion gain is lower than simulation result but not similar to measured result. So, replaced the TSMC_v3 model to new TSMC_1.2A model and bond-wire inductor fix
to 0.5nH, the sim 4.32, the
11 is alike between previous and current version of TSMC model but the conversion ain is serious degrade at high frequency. In Fig. 4.34, a capacitor (Cp=100fF and 00fF) is included between the gate and source of the transistor in TSMC_v3 model.
ss and Troubleshooting
gain whose reason m
ulation result is shown in Fig. 4.32 and Fig. 4.33. In Fig.
S g 4
We find out odel’s
conversion gain is si ilar to parasit 00f the
gate and source of the transistor in old m g 4.35 is sh version
gain with and ide source. The rence is
around 1dB. So, we onclude the old m t accurate at h ncy. The noise figure with and without board simula wn in Fig. 4.36
a phenomenon in Fig. 4.33 and Fig. 4.34. The new TSMC m m ic capacitor about 4 F included between
odel. The Fi own the con PMOS current source al current gain diffe
can c odel is no igh freque
Fig. 4.30 S11 Momentum Boardsim with various bond-wire values
Momentum_Boardsim_Lg=1nH
Fig. 4.31 Conversion gain Momentum Boardsim with various bond-wire values
Momentum_Boardsim_TSMC_1.2A(new)
Fig. 4.32 S11 Momentum Boardsim with two TSMC models
Momentum_Boardsim_TSMC_1.2A(new)
Fig. 4.33 Conversion Gain Momentum Boardsim with two TSMC models
Momentum_Boardsim_Cp=100fF
Fig. 4.34 Conversion Gain Momentum Boardsim with various parasitic capacitors
Measured result
Fig. 4.35 Conversion Gain with and without PMOS current source
Simulation without board effect Simulation with board effect
2 4 6 8
0 10
10 15
5 20
Freq (GHz)
NFdsb
Fig. 4.36 Noise figure with and without Momentum Boardsim
In this work, we design a 3~8GHz direct conversion broadband mixer of the second stage of ultra wideband receiver front end. This circuit uses the chebyshev filter for input matching in order to reduce extra power consumption. The inductive peaking also uses in this circuit in order to compensation the high frequency transconductance. Due to the RO4003C board’s transmission line effect and inaccuracy TSMC model, it makes the return loss poor and conversion gain will be degrade at high frequency as we discuss in section 4.3.5. The measured maximum conversion gain is 7.4dB at 3-MHz IF band. The average measured IIP3 is 3.4dB. The bandwidth is 3GHz from 2 to 5GHz. The input return loss is less than 4dB at 3 to 8GHz.
4.4 Conclusion
Chapter 5
Summary and Future Works
5.1 Summary
In the chapter 2, we will introduce the fundamentals of conventional low noise amplifiers and mixer. And some theoretical MOSFET noise model and noise theory are presented. LNA noise analysis also introduce in this chapter. Furthermore, many important design parameters and direct conversion receiver front end would be presented in this chapter.
In the chapter 3, we design an ultra wideband low noise amplifier for the receiver used the standard TSMC 0.18µm CMOS process. This circuit uses the feedback resistor to achieve the input, output broadband matching. But actually, the transistor’s parasitic capacitor (Cgs) will serious degrade performance at high frequency. So, we add inductive degeneration matching method to enhance bandwidth. And between two stages, we add an inductor to compensation the high frequency transconductance. The measured result shows that S11, S22 and S12 parameters are similar to simulation result. The measured 3dB frequency range is from 0.1 to 6.6GHz. The measured maximum power gain is 6.2dB. The average IIP3 is -1.6dB. The noise figure minimum is 6.5dB.
In the chapter 4, we design a 3~8 GHz direct conversion broadband mixer of the second stage of ultra wideband receiver front end. This circuit uses the chebyshev filter for input broadband matching. Its advantage has easy to achieve high gain, broadband matching and doesn’t consume extra power consumption but it needs large die size and accurate inductor model. The inductive peaking also uses in this circuit in path of UWB system which is
order to compensation the high frequency transconductance. Due to the board’s transmission line effect, it makes the return loss poor at high frequency. The measured
s .4dB. The bandwidth is 3GHz form 2 to 5GHz. The input return loss is less than 4dB
ency band. But actually, the measured results present poor gain
maximum conversion gain is 7.4dB at 3-MHz IF band. The average measured IIP3 i 3
at 3 to 8GHz.
5.2 Future Works
The ultra-wideband LNA and mixer using frequency compensation can achieve flatness gain at UWB frequ
and noise figure in comparison to our simulation results. This could be the lower transconductance gain and inaccuracy model. It is presented in section 4.3.5. In next work, we can replace the TSMC accuracy model then the measured data will be close to the simulation result. Furthermore, we can merge LNA and Mixer with an broadband active balun.
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簡歷
國立交通大學電子工程所碩士班 ( 90 年 9 月 ~ 94 年 5 月) 姓 名: 林志修
學 歷:
國立台北科技大學電子工程學系 ( 85 年 9 月 ~ 87 年 6 月)