Chapter 3 Ultra-wideband Low Noise Amplifier
3.4 Conclusion
In this work, we design an ultra wideband low noise amplifier for the receiver path of UWB system which is used the standard TSMC 0.18µm CMOS process. This circuit uses the feedback resistor and inductive degeneration to achieve the input, output broadband matching. Between two stages, we add an inductor to compensation the high frequency transconductance. The measured result shows that S11, S22 and S12 parameters are similar to simulation result. It shows the L-degeneration method is useful at this work. The measured 3dB frequency range is from 0.1 to 6.6GHz. The measured maximum power gain is 6.2dB. The average IIP3 is -1.6dB. The noise figure minimum is 6.5dB. From Table 3.1, the power consumption of our work is lower than other distributed amplifier [8,9,10] and die size is also small than others.
Chapter 4
A 3 ~ 8 GHz Direct Conversion Broadband Mixer
This chapter describes circuit design principle of a 3~8GHz direct conversion broadband mixer. And this chip fabricated by TSMC 0.18μm RF CMOS technology.
4.1 Introduction
Due to data communication capacity increase progressively, the wide bandwidth and low power are the urgent requirements.
Direct conversion receiver (DCR) is most popular architecture in recently years.
The frequency of RF signal is the same with the frequency of LO (local oscillator) signal, and down converts to around DC frequency. The homodyne receiver (DCR) offers two important advantages over a heterodyne receiver. First, there are no image problems, so image rejection filter is not required. Second, the SAW filter and subsequent down-conversion stages are replaced with low pass filter and baseband amplifiers that are amenable to monolithic integration. In general, the homodyne receiver just needs small chip area.
In this chapter, we employ chebyshev filter to design a 3~8GHz direct conversion mixer. Its advantage has easy to achieve high gain, broadband matching and doesn’t consume extra power but it needs large die size and accurate inductor model. Therefore, this mixer can receive signals from 3~8GHz and down convert to baseband signal directly. The extra current source is used to decrease the current of switch MOS, and increase the lower stage’s transconductance gain. An inductor adds to the inter-stage between the switch and transconductance stage whose the high frequency gain can be improved.
4.2 Principle of the Circuit Design
Fig. 4.1 Schematic of 3~8GHz direct conversion broadband mixer
In this section, the circuit design principle of 3~8GHz direct conversion mixer is presented. This circuit of mixer uses chebyshev filter design to achieve broadband matching. In section 4.2.1 will introduce the broadband matching. Inductor adds to the inter-stage between the switch and transconductance stage then the high frequency gain can be improved. In section 4.2.2 will introduce the inductive peaking function. In order to down covert the RF signal, the mixing stage is needed. In section 4.2.3 will introduce the mixing stage.
4.2.1 Input Broadband Matching
Fig. 4.2 shows the input impedance matching of this circuit. Input matching uses the chebyshev filter design. The filter design technique has two kinds of methods. One
is the image parameter method, and another one is insertion loss method [11]. In this work, we use insertion loss method to design filter. Insertion loss uses network synthesis techniques to design filters with a completely specified frequency response.
The design is simplified by beginning with low pass filter prototypes that are normalized in terms of impedance and frequency. Transformations are then applied to convert the prototype designs to the desired frequency range and impedance level. So, the insertion loss method is the most common technique to design filter. The Fig. 4.3 shows the filter design procedure of insertion loss method.
Fig. 4.2 The input impedance matching
Fig. 4.3 Insertion loss method design procedure
4.2.1.1 Low Pass Filter Prototype
In the insertion loss method a filter response is defined by its insertion loss, or power loss ratio, PLR:
LR 2
Power available from source 1
P = Power delivered to load =1 | ( ) |− Γω (4-1)
where | ( ) | = ( )Γω 2 Γω ⋅ Γ*( )= ( ) (- )= | (- ) |ω Γω ⋅ Γ ω Γ ω 2, so
Substituting this form in equation (4-1), we get
(4-3)
hus, for a filter to be physically realizable its power loss ratio must be of the form in
2
equation (4-3). Notice that specifying the power loss ratio simultaneously constrains the reflection coefficient, ( )Γω . We now discuss two practical filter responses below.
<1>Butterworth filter:
Butterworth response provides the flattest possible passband response for a given filter complexity, or order. For a low pass filter, it is specified by
2 2N
P =1+k (LR )
c
ω
ω (4-4)
where N is the order of the filter
then a sharper cutoff will result, although the passband response will have ripples of amplitude , as shown in Fig. 4.4, since TN(x) oscillates between ±1 for |x| 1.
Thus, determines the passband ripple level. For large x,
1 k+ 2 ≤
k 2 T (x) 1/2(2x)N N
Fig. 4.4 Butterworth and Chebyshev Low pass filter responses (N=3)
From the power loss ratio equation of butterworth and chebyshev filter, then we could derive the normalized element values of L and C of low pass filter prototype.
The element values for the ladder type circuits of Fig. 4.5 and the normalize values of L and C are presented in Table 4.1 and Table 4.2.
Fig. 4.5 Ladder circuits for Low pass filter prototypes and their element definitions
Table 4.1 Element values for Butterworth Low Pass Filter prototypes(g0=1,ωc=1)
N (order) g1 g2 g3 g4
1 2.0 1.0
2 1.4142 1.4142 1.0
3 1.0 2.0 1.0 1.0
Table 4.2 Element values for Chebyshev Low Pass Filter prototypes(g0=1,ωc=1) Ripple=0.5dB
N (order) g1 g2 g3 g4
1 0.6986 1.0
2 1.4029 0.7071 1.9841
3 1.5963 1.0967 1.5963 1.0
4.2.1.2 Impedance and Frequency Scaling
Impedance scaling: In the prototype design, the source and load resistances are
un ).
A ltiplying the impedances of the
prototype d . Then, if we let prim note im nce scale antities, we have the new filter compone given
(4-6) ity (expect for chebyshev filters with even N, which have nonunity load resistance
source resistance of R0 can be obtained by mu
esign by R0 es de peda d qu where L, C, and RL are the component values for the original prototype
Frequency scaling for low pass filters: To change the cutoff frequency of a low pass prototype from unity to ωc requires that we scale the frequency dependence of the filter by the factor 1/ωc, which is accomplished by replacing ω by ω/ωc:
c
ω ω
←ω (4-10) then the new power loss ratio will be
LR LR
c
P' ( )=P (ω ω )
ω (4-11) where ωc is the new cutoff frequency; cutoff occurs when ω/ωc = 1, or ω=ωc.
Therefore, the new element values are determined by applying the substitution of (4-10) to the series reactances, jωLk, and shunt susceptances, jωCk, of the prototype filter. Thus,
y scaling are required, thus the new value is
= ⇒ (4-13)
responses illustrated in Fig. 4.6. If ω1 and ω2 denote the edges of the passband, then a bandpass response can be obtained using th
4.2.1.3 Bandp
r designs can also be transformed to have the bandpass
e following frequency substitution:
0 0 1 0
the fractional bandwidth of the passband. The center frequency, ω0, could be chosen ω2, but the equations are simpler if it is chosen as
the low pass response of Fig. 4.5(a) as follows:
When is
as the arithmetic mean of ω1 and the geometric mean:
Then the transformation of (4-16) maps the bandpass characteristics of Fig. 4.5(b) to
0= 1 2
Fig. 4.6 (a) Low pass filter prototype
(b) Transformation to bandpass filter frequency response frequency response for ωc=1
The new filter elements are determined by using (4-16) in the expressions for the series reactance and shunt susceptances. Thus,
k 0 k k which shows that a series inductor, Lk , is transformed to a series LC circuit with element values,
ansformed to a shunt LC circuit with element values, which shows that a shunt capacitor, Ck , is tr
k
The low pass filter elements are thus converted to series resonant circuits (low impedance
impedance at resonance) in the shunt arms. Notice that both series and parallel resonator elements have a resonant frequency of ω0. The Fig. 4.7 shows the condition. The Fig. 4.8 is the
(
at resonance) in the series arms, and to parallel resonant circuits (high
complete circuit of low pass filter transformed to bandpass filter.
Fig. 4.7 Components transformation of low pass filter transfer to bandpass filter (a) Series inductor transformed to series LC
(b) Parallel capacitor transformed to shunt LC
Fig. 4.8 Complete circuit of bandpass filter tansformation
4.
end mixer which help us
to understan ixer is the same as the
single end. In this double balanced gilbert mixer, we put an inductor (Lp1) between
2.2 Inductive-peaking Function
The Fig. 4.9 is the inductive peaking function of single d this function [12,13]. The differential pair m
the transconductance stage and mixing stage in order to enhance the high frequency gain. The equivalent circuit is shown in Fig. 4.10. The capacitor (Cp) and resistor (R) is the equivalent model that looking into the mixing stage. The capacitor (CL) is the equivalent element which looking into the drain node of transconductance stage. As the Fig. 4.10(a), we can derive
out in
L p
V =I R
sR(C +C )+1
⋅ (4-26)
where
L p
R (C +C )⋅
(4-27) is shown the pole location would be decrease the signal at high frequency.
As the Fig. 4.10(b) shown, we can derive relationship of current and voltage with inductive peaking function.
= -1
ω ( pole ) (4-27)
out
p L p1 L p1 p L
V =I ⋅ R (4-28)
We can adjust Lp1 to enhance transconductance gain (Gm) at high frequency. The Fig.
4.10 shows the result that with
in s RC C L +s C L +sR(C +C )+13 2
and without inductive peaking.
Fig. 4.9 The inductive peaking function
Fig. 4.10 (a) without inductive peaking (b) with inductive peaking
L-peaking non L-peaking
2 4 6 8
0 10
0.01 0.02 0.03 0.04
0.00 0.05
Freq (GHz)
Gm (A/V)
Fig. 4.11 Gm curve with and without inductive peaking
4.2.3 Switching Stage of the 3~8 GHz Direct Conversion Mixer
The Fig. 4.12 is the mixing stage of this mixer. The main noise source of the mixer comes from the flicker noise. The (4-29) is the noise current representation.
m2
2 2
n 2 T
ox
K g K
i f
f WLC f ω
= ⋅ ⋅ ∆ ≈ ⋅ ⋅ ⋅ ∆A f (4-29)
From the equation (4-29), we would observe two conditions to reduce noise. One is increase the transistor area size, and another one is decrease the transistor’s gm. The smaller gm means the less current flow of the transistor. The bias current of the transistor set to about 500μA which is around the saturation region. It can improve the noise figure that bias at around the saturation region. The current source is used to enhance the voltage conversion gain and d rrent flow of the transistor of M3-M6.
ecrease the cu
Fig. 4.12 Switching stage of the 3~8GHz direct conversion mixer
4.3 Chip Implementation and Measured Result
4.3.1 Microphotograph of Chip
A microphotograph of the 3~8GHz direct conversion broadband mixer circuit is shown in Fig. 4.13. The circuit is fabricated in the TSMC 0.18µm CMOS technology.
The die area including bonding pads is 0.96 mm by 0.985 mm.
Fig. 4.13 Microphotograph of 3~8GHz Direct Conversion Broadband Mixer
Gnd
Gnd RF-
L RF+
s2 M2 M1 Ls1
Ld1
Ld2
Vdd Vdd
Rd2 M6 Rd1 M3
Vdd Vb2 Vb1 Vdd
LO+
LO-IF+
IF-4.3.2 Photograph of On Board Measurement
This 3~8Ghz broadband mixer is used by on board probing. The photograph of on board probing is shown as Fig. 4.14. The output stage uses an OP Amplifier (AD9631) as output buffer. The measurement setup is shown Fig. 4.15. The RF port and LO port that we design is differential, so we need to connect a balun to transfer signal. This balun type we choose is 30056 and 30057 as in . The m ent characteristic of Anaren 30056 balun is shown from Fig. 4.17 to Fig.
4.19. The Fig. 4.17 is the return loss of balun. The Fig. 4.18 is the gain difference curve. The Fig. 4.19 is the phase difference curve. The Anaren 30057’s characteristic is shown from Fig. 4.20 to Fig. 4.22.
Anaren shown Fig. 4.16
easurem
LO Port
IF Port
Fig. 4.14 Photograph of on board measurement of broadband mixer
RF Port
Fig. 4.15 Block diagram of measurement setup
Fig. 4.16 Photograph of Anaren Balun 30056 and 30057
Port 1 Port 2 Port 3
1 2 3 4 5
0 6
-40 -30 -20 -10
-50 0
Freq (GHz)
dB(S11, S22, S33)
Fig. 4.17 Return loss of Anaren Balun 30056
In phase diff phase
1
3 2
3 2
1
diff phase In phase
Gain Diff
Fig. 4.18 Gain difference of Anaren Balun 30056
m5 freq=2.000GHz
Fig. 4.19 Phase difference of Anaren Balun 30Port 1
Fig. 4.20 Return loss of Anaren Balun 30057
Gain Diff
Fig. 4.21 Gain difference of Anaren Balun 30057
m5 freq=4.000GHz
4.3.3 On Board PCB Layout Consideration
Because of our broadband mixer design is at high frequency range, we must carefully layout our PCB board. The PCB layout diagram is shown Fig. 4.23. First, the transmission line’s distance between the series inductor and series capacitor (series resonance) can’t be long. In this layout, we just reserve two components’ pad. This is because long distance transmis would influence our input broadband matching. Second, the parallel inductor and capacitor (parallel resonance) would be layout into the signal path, otherwise, it could causes the loading effect. Then, we put the bypass capacitor 0.1uF near the tank at the bias voltage Vg. It let the ac signal ground. The total length of transmission line between two series resonance can’t exceed 5mm.
ference of Anaren Balun 30057
sion line
Fig. 4.23 On board PCB layout diagram
.3.4 Measurement and Simulation Result
asured S-parameter is plotted in
measured result. The solid plot is the simulation result. This mixer’s andwidth is about 3GHz from 2 to 5 GHz. The IIP3 measured result is shown in Fig.
.26 and Fig. 4.27. The simulation noise figure is shown in Fig. 4.27. The minimum
4
Measurement is used on-board probing. The me
Fig. 4.24, together with simulation results for comparison. The circuit plot is the measured result. The solid line is the simulation result. The triangle plot is the measured result with no balun connection. The low frequency is matched the simulation result, but high frequency is not. The cause is the transmission line performance is not good at high frequency. The board we use is RO4003C. The Fig. 4.28 is the measured result of various transmission line width of RO4003C. In these curves, we can observe the transmission line’s return loss is poor in 3 ~ 6 GHz.
The measured maximum conversion gain is 7.4dB at 3-MHz IF band. The Fig.
4.25 is plotted the measured result, together with simulation result for comparison. The circle plot is the
b 4
noise figure is 7.1dB. The total power of this broadband mixer is 11.8mW with a power supply 1.6V. The table 4.3 is the comparison of simulation and measured result.
Simulation result
Fig. 4.24 S11 Simulation and measured result of broadband mixer
simulation result
Fig. 4.25 Conversion gain simulation and measured result of broadband mixer
IIP3=1.23dBm 3GHz two tone test
-18 -13 -8 -3
-23 2
-70 -60 -50 -40 -30 -20 -10 0
-80 10
Input Power(dBm)
Output Power(dBm)
Fig. 4.26 Linearity measured result at 3GHz of 3~8GHz mixer
4 5 6 7
3 8
0 5
-5 10
Freq (GHz)
IIP3 (dBm)
Fig. 4.27 Linearity measured result versus frequency of broadband mixer
2 4 6 8
0 10
10
5 15
Freq (GHz)
Noise Figure (dB)
Fig. 4.28 Simulation result of Noise Figure of broadband mixer
W=25 mil W=26 mil W=27 mil W=28 mil
W
2 4 6 8
0 10
-40 -30 -20 -10
-50 0
Freq (GHz)
dB(S(1,1))
Fig. 4.29 Various width of transmission line of RO4003C
TABLE 4.3 Comparison of simulation and measured performance Simulation Result Measured Result
Bandwidth 3 ~ 8 GHz 2 ~ 5 GHz
According to previous section, we observe the worse return loss and conversion aybe the transmission line and bond-wire’s variation. Then, we use ADS Momentum to simulation the transmission line of RO4003C’s board effect.
The Fig. 4.30 and 4.31 are the simulation results of considering the board effect and bond-wire variation. The cross plot is the bond-wire inductor which value is 1nH. The triangle plot is the bond-wire inductor which value is 0.5nH. The diamond plot is the bond-wire inductor which value is 1.5nH. In Fig. 4.30, the measured result is similar to the bond-wire inductor which value is 0.5nH. Relatively, in Fig. 4.31, the conversion gain is lower than simulation result but not similar to measured result. So, replaced the TSMC_v3 model to new TSMC_1.2A model and bond-wire inductor fix
to 0.5nH, the sim 4.32, the
11 is alike between previous and current version of TSMC model but the conversion ain is serious degrade at high frequency. In Fig. 4.34, a capacitor (Cp=100fF and 00fF) is included between the gate and source of the transistor in TSMC_v3 model.
ss and Troubleshooting
gain whose reason m
ulation result is shown in Fig. 4.32 and Fig. 4.33. In Fig.
S g 4
We find out odel’s
conversion gain is si ilar to parasit 00f the
gate and source of the transistor in old m g 4.35 is sh version
gain with and ide source. The rence is
around 1dB. So, we onclude the old m t accurate at h ncy. The noise figure with and without board simula wn in Fig. 4.36
a phenomenon in Fig. 4.33 and Fig. 4.34. The new TSMC m m ic capacitor about 4 F included between
odel. The Fi own the con PMOS current source al current gain diffe
can c odel is no igh freque
Fig. 4.30 S11 Momentum Boardsim with various bond-wire values
Momentum_Boardsim_Lg=1nH
Fig. 4.31 Conversion gain Momentum Boardsim with various bond-wire values
Momentum_Boardsim_TSMC_1.2A(new)
Fig. 4.32 S11 Momentum Boardsim with two TSMC models
Momentum_Boardsim_TSMC_1.2A(new)
Fig. 4.33 Conversion Gain Momentum Boardsim with two TSMC models
Momentum_Boardsim_Cp=100fF
Fig. 4.34 Conversion Gain Momentum Boardsim with various parasitic capacitors
Measured result
Fig. 4.35 Conversion Gain with and without PMOS current source
Simulation without board effect Simulation with board effect
2 4 6 8
0 10
10 15
5 20
Freq (GHz)
NFdsb
Fig. 4.36 Noise figure with and without Momentum Boardsim
In this work, we design a 3~8GHz direct conversion broadband mixer of the second stage of ultra wideband receiver front end. This circuit uses the chebyshev filter for input matching in order to reduce extra power consumption. The inductive peaking also uses in this circuit in order to compensation the high frequency transconductance. Due to the RO4003C board’s transmission line effect and inaccuracy TSMC model, it makes the return loss poor and conversion gain will be degrade at high frequency as we discuss in section 4.3.5. The measured maximum conversion gain is 7.4dB at 3-MHz IF band. The average measured IIP3 is 3.4dB. The bandwidth is 3GHz from 2 to 5GHz. The input return loss is less than 4dB at 3 to 8GHz.
4.4 Conclusion
Chapter 5
Summary and Future Works
5.1 Summary
In the chapter 2, we will introduce the fundamentals of conventional low noise amplifiers and mixer. And some theoretical MOSFET noise model and noise theory are presented. LNA noise analysis also introduce in this chapter. Furthermore, many important design parameters and direct conversion receiver front end would be presented in this chapter.
In the chapter 3, we design an ultra wideband low noise amplifier for the receiver used the standard TSMC 0.18µm CMOS process. This circuit uses the feedback resistor to achieve the input, output broadband matching. But actually, the transistor’s parasitic capacitor (Cgs) will serious degrade performance at high frequency. So, we add inductive degeneration matching method to enhance bandwidth. And between two stages, we add an inductor to compensation the high frequency transconductance. The measured result shows that S11, S22 and S12 parameters are similar to simulation result. The measured 3dB frequency range is from 0.1 to 6.6GHz. The measured maximum power gain is 6.2dB. The average IIP3 is -1.6dB. The noise figure minimum is 6.5dB.
In the chapter 4, we design a 3~8 GHz direct conversion broadband mixer of the second stage of ultra wideband receiver front end. This circuit uses the chebyshev filter for input broadband matching. Its advantage has easy to achieve high gain, broadband matching and doesn’t consume extra power consumption but it needs large die size and accurate inductor model. The inductive peaking also uses in this circuit in path of UWB system which is
order to compensation the high frequency transconductance. Due to the board’s transmission line effect, it makes the return loss poor at high frequency. The measured
s .4dB. The bandwidth is 3GHz form 2 to 5GHz. The input return loss is less than 4dB
ency band. But actually, the measured results present poor gain
maximum conversion gain is 7.4dB at 3-MHz IF band. The average measured IIP3 i 3
at 3 to 8GHz.
5.2 Future Works
The ultra-wideband LNA and mixer using frequency compensation can achieve flatness gain at UWB frequ
and noise figure in comparison to our simulation results. This could be the lower transconductance gain and inaccuracy model. It is presented in section 4.3.5. In next work, we can replace the TSMC accuracy model then the measured data will be close to the simulation result. Furthermore, we can merge LNA and Mixer with an broadband active balun.
REFERENCES
Available:
[1] IEEE 802.15 WPAN High Tate Alternative PHY Task Group 3a(TG3a)[Online].
http://www.ieee802.org/15/pub/TG3a.html
Paul R. Gray, Paul J. Hurst, Stephen H. Lewis, Robert G. Meyer, “Analysis and
Paul R. Gray, Paul J. Hurst, Stephen H. Lewis, Robert G. Meyer, “Analysis and