International Standard IEC 61000-3-2 - A SURVEY of PREVIOUS SINGLE-STAGE PFC CIRCUITS

CHAPTER 2. A SURVEY of PREVIOUS SINGLE-STAGE PFC CIRCUITS

2.5 International Standard IEC 61000-3-2

While a power sets are design, the power quality must be controlled. Thus, in the international society, some standards for measuring the power quality are published. One of the important standards is the International Standard IEC 61000-3-2. The outline of it is briefly mentioned as follows.

For the purpose of harmonic current limitation, equipment is classified as follows:

Class A: - Balanced three-phase equipment;

- Household appliances, excluding equipment identified as class D;

- Tools, excluding portable tools;

- Dimmers for incandescent lamps;

- Audio equipment.

Class B: Portable tools, arc welding equipment, including dimming device.

Class C: Lighting equipment, including dimming device.

Class D: Equipment having a specified power according to 6.2.2 (the item shown in IEC 61000-3-2) less than or equal to 600 W of the following types:

- Personal computers and personal computer monitors;

- Television receivers.

The converters proposed in this thesis shall be of the power sets applied in modern electronic products, such as personal computers, computer peripherals, and television receivers. All these electronic products’ input power is less than 600W. Therefore, the experimental results will be criticized by employing the standard class D, which gives the current limits for eleven major harmonics as shown in Table 2.1.

Table 2.1 Limits for class D equipment Harmonic order n Maximum permissible harmonic

current per watt mA/W

Maximum permissible harmonic current A

3 5 7 9 11

13 ≦ n ≦ 39(odd harmonics only)

3.4 1.9 1.0 0.5 0.35 3.85/n

2.3 1.14 0.77 0.4 0.33

0.21, 15/n when n≧15

CHAPTER 3 THE FLYBACK CONVERTER USING THE PROPOSED INPUT CURRENT SHAPER

Figure 3.1 shows the proposed new flyback ac/dc converter with harmonic current correction function and tight output regulation. The circuit is a single-stage single-switch ac/dc converter, which comprises single switch S1, an input filter Lf, Cf and C1, a bulk capacitor C2, a soft-switching inductor L1 and a transformer with two primary windings N1

and N2. The winding N1, inductor L1, diode D1 and D2 form an input current shaper cell.

Besides, the winding N1, inductor L1, diode D1 and D2, switch S1, and bulk capacitor C2

comprise a boost circuit. The winding N2 and N3, bulk capacitor C2, switch S1, diode D3 and output capacitor C3 form a flyback converter.

The function of inductor L1 in the proposed circuit is different from that of the boost inductor presented in the converters of [2]-[10]. Actually, L1 provides partial soft switching functions for diodes D1 and D2, to suppress the harmonic current by increasing the conduction time for iac from time t1 to t5 in Fig. 3.2, and to reduce the voltage VC2 across the bulk capacitor.

R_L

Fig. 3.1 Proposed flyback AC/DC converter with ICS

The inductor L1 has a soft-switching function on D1 and D2, as mentioned in [12]. Figure 3.3 shows that when S1 turns off, D2 changes to reverse bias and proceed soft off since the current iN1 has gradually reduced to zero at the reverse bias time t1,M3 or t1,M2 . Therefore, to overcome the problem of the reverse recovery effect of D2, a suitable inductance of L1 must be selected. Contrarily, D1 softly turns on when the current iN1 gradually increases from zero at time t2,M2/t2,M3.

The winding N1 provides the voltage-boost function for bulk capacitor C2 during the period from t1 to t5, as illustrated in Fig. 3.2. During this period, when S1 turns off, D1 turns on and the charge current flows from the power line source to C2 through N1, L1, and D1. At this moment, the residue magnetic energy stored in the transformer will also induce current iN3 as a falling ramp waveform, as illustrated in Fig. 3.3. Furthermore, the increasing current iN1

keeps storing the magnetic energy in L1. The magnetic energy stored in L1 passes to winding N2 through winding N1, and induces a portion of current iN2 when S1 is turned on.

The turns-ratio, n1/n2, of the transformer can determine not only the starting conduction angle of the line current but also the voltage across a bulk capacitor C2. Furthermore, the inductance and volume of L1 are significantly smaller than the primary windings N1 or N2 of the transformer.

The control circuit can be designed by using a fixed-frequency simple voltage-mode control or a conventional peak-current control. The experiment results have demonstrated that even using a simple control method, the line current of the proposed AC/DC converter can comply with the standard IEC 61000-3-2, and the converter also provides fast load dynamic response.

3.1 Basic Operation Theories

The fundamental operating principle of the proposed converter is to store the magnetic energy in windings N2 when switch S1 turns on, and then to deliver it to bulk capacitor C2 and secondary winding N3 when switch S1 turns off. The entire operation principle of the circuit can be explained in three operation modes. Figure 3.2 shows six operation modes in a line cycle. Only three modes are left after combining the similar modes, namely M1/M6, M2/M5

and M3/M4. Figure 3.3 shows the main current and voltage waveforms in every mode.

Fig. 3.2 Operation modes in proposed flyback converter

t_0,M1 t_1,M1

Fig. 3.3 Current and voltage waveforms in a switching cycle in M1~M3.

3.1.1 Operation Modes M1 (t0 ,t1) and M6 (t5, T/2)

This mode holds when 0<|Vac|<Vc2 -V_o×(n1/n₃). Currents |iac| and iN1 have not yet been induced. The converter operates as a conventional flyback converter. Figure 3.4 shows the current conducting path in mode M1/M6 with S1 turned on and off. Figure 3.4 shows that the transformer does not sink the current from the power line. Rather, the converter sinks the current from the bulk capacitor C2. VC2 shows the voltage across on C2 and is approximated to a constant value during a line cycle in the steady state and can be obtained as follows:

The voltage-second balance criterion is applied to the flyback transformer, and thus the total voltage-second should be zero in one time period in steady state. Additionally, another required assumption is that the flyback transformer operates in the CCM mode such that

From equation (3.1) and (3.2) the boundary time of M1 can be obtained by

Integrating the winding inductor voltages of VN2 and VN3 over the duty on and off periods yields

⎪⎩

winding N2 and S1. The conducting paths are as shown in Figs. 3.5(a) and (b) since the induced voltage VN1 exceeding |Vac|, iN1 is none zero current initially, as displayed in Fig. 3.5(a) and reduces to zero linearly and then continues to be zero, as shown in Fig. 3.5(b).

When S1 turns off, VN1 causes iN1 to flow via winding N1, L1 and D1 and to charge the bulk capacitor C2. Simultaneously, D3 turns to on state and delivers the magnetic power to the output circuit, as shown in Fig. 3.5(c). The conduction of D3 causes the output capacitor connected to two terminals of winding N3. Therefore, the output current iN3 linearly reduces during the duty off period. In this operation, iN3 is nonzero even at the end of the duty-off period. Consequently, in mode M2 the current iN3 of winding N3 operates in the continuous current mode, denoted by CCM. Based on the CCM of iN3 and voltage-second balance, the duty ratio D is the same as that in mode M1/M6.

Integrating the voltages of the winding inductors of VN1, VN2, and VN3 yields the following winding currents:

⎪⎪

where iN2,M2(t1,M2), the magnetic current appearing in winding N2, comes from three sources, )

during the duty on period. The resulting formula is obtained as

i_S1

)

3.1.3 Boundary condition between CCM and DCM

The boundary between CCM and DCM occurs just as iN3 reaches zero at the end of the switching period. Since the capacitance of C2 is large, the value of the voltage VC2 is almost kept constant. Throughout the M2 period, when S1 is in the off period, the N1 current generated by the line power increases, and accelerates the decrease of iN3 according to Ampere’s law. If the duty ratio remains unchanged in Mode M2, then the current iN1(t3,M2) at the end of M2 equals the current iN3(t2,_M1)×(n3 /n1) at the end of mode M1. This approximation yields

3.1.4 Operation modes M3(t2,t3)and M4(t3,t4)

This mode holds when VBD < |Vac|<Vm. The large current iN1 increases the rate of decay of iN3. The current iN3 falls to discontinuous current mode (DCM) in this operational mode.

Figure 3.6 shows four different current flow paths in a switching cycle. The energy stored in winding N2 during the duty on period of S1 are distributed to the winding N1 and winding N3

Fig. 3.6 Current loops in mode M3: (a) t0,M3 ≤ t < t1,M3, S1 turns on (b) t1,M3 ≤ t <

t2,M3, S1 turns on (c) t2,M3 ≤ t < t3,M3, S1 turns off (d) t3,M3 ≤ t < t4,M3, S1 turns off

The winding currents and voltages of inductor L1 and transformer are given as follows:

⎪⎪

where VL1 is calculated via the law of the conservation of voltage-second for L1 in a switching cycle.

Employing equations (3.15)-(3.18) with the voltage-second balance theorem to winding N2 in mode M3 in steady stat gives the following equations.

Simplifying the above equation, the duty ratio D can be obtained by,

)

3.2 Analysis of Converter Operation 3.2.1 Primary current iN1 and Duty ratio D

In the converter circuit the filter capacitance C1 is designed as a low pass filter to bypass the switching signal to ground and to pass the line power signal to the converter.

Consequently, the primary component of |iac|approximates to the primary component of iN1. From equations (3.7) and (3.12), iN1 and VC1 display a linear relation when the angle of the line power signal exceeds the conduction angle. Therefore, iac can be controlled to linearly vary with Vac during the conduction periods.

The secondary winding N3 of the transformer operates in CCM during modes M₁ and M2

and in DCM during mode M3. The calculation of duty ratio can be simply obtained from equation (3.2), when winding N3 operates in modes M1 and M2. However, the duty ratio becomes more complicate as given in equation (3.19) for mode M3, since the current iN3 enters DCM. The value of duty ratio in mode M3 is smaller than in modes M1 and M2. Furthermore, the duty ratio will be smallest when the peak Vac presents, since ^D^~ increases and ^F^~ decreases when |Vac| increases from zero to the peak value.

3.2.2 Starting conduction angle

The value ω(t₁−t₀) is called a starting conduction angle (SCA), as shown in Fig. 3.7. A smaller SCA leads to higher power factor and lower THD. Equation (3.3) shows that the SCA increases with increasing product of Vo/Vm and n2/n3. This phenomenon implies that power factor or THD decreases with increasing Vo/Vm or n2/n3. Figure 3.8 shows the relationship between Vo/Vm and SCA under various duty ratios D and two different winding ratios of n1/n2.

3.2.3 Voltage across bulk capacitor

Equation (3.1) shows that the voltage across bulk capacitor, VC2, varies with Vm, SCA, Vo, and n1/n3 but does not vary with the output load. In most applications, all the design calculations are always based on the given values of Vm and Vo. Thus, the voltage VC2 can be determined by selecting the preferred n1/n3, SCA, or n2/n3. In practical applications, the

0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6

0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8

0.9 n2/n3=2

Starting conduction angle(rad)

Vo/Vm

D=0.17 D=0.24 D=0.31 D=0.38

D=0.45

D=0.1

n1/n2=0.9 n1/n2=1.2

Fig. 3.7 Starting conduction angle

voltage VC2 is kept under 450v for commercial considerations. Figure 3.9 provides designers with a convenient graphical design aid for obtaining the eclectic selections of n1/n3, SCA, and VC2 for certain line voltage ranges.

According to Fig. 3.2, the current iN1 is zero in the mode M1 because the sum of VC1 and VN1 is smaller than VC2 when S1 is in the off state. The diode D1 continues off until the sum of VC1 and VN1 exceeds VC2. Therefore, SCA decreases with decreasing VC2. Two methods can be used to reduce VC2. One method uses a smaller number of winding turns for n1, and the other uses a larger inductance in L1. The larger inductance L1 can resist iN1 to charge VC2, thus achieving lower VC2.

According to equation (3.2), n2/n3 is the dominant parameter to influence the value of VC2. L1 and n1 are the dominant parameters to influence the harmonic current and line current waveform. The turn number n1 can be larger than n2 or smaller than n2. If n1 is smaller than n2, it will not gain any benefit and the harmonics current will become a poor value. Figure 3.11 and 3.12 show the simulation results in different turn ration n1/n2, and the lower value of n1/n2

will get higher harmonics current. Figure 3.13 also display the harmonics current in different turn rations.

0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6

0 100 200 300 400 500

600 Vc2 & SCA,n1/n3 @Vo=48V

Starting conduction angle(rad)

Vc2(v)

0.51.0 1.52.02.5 3.0 0.51.0 1.5

2.02.5 Vac=265V 3.0

Vac=85V n1/n3=

n1/n3=

Fig. 3.9 Curve of starting conduction angle,VC2, and n1/n3 at Vo=48v.

0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.5 -0.05

0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4

L1/LN2 & Duty Cycle(N2/N3=2,Vo/Vm=.42)

Duty Cycle

max L1/LN2

n1/n2=0.5

n1/n2=0.85 n1/n2=1.2

n1/n2=1.55

n1/n2=1.7

Fig. 3.10 Max L1/LN2 and Duty cycle

Fig. 3.11 Waveforms of iac&Vac at n1:n2:n3=1.2:1:0.5, L1=33uH

Fig. 3. 12 Waveforms of iac&Vac at n1:n2:n3=0.8:1:0.5, L1=33uH

0 0.5

1 1.5

2 2.5

3 5 7 9 11 13 15 17 19 21 IEC61000-3-2 n1>n2

n1<n2

Harmonic numbers

0 0.5

1 1.5

2 2.5

3 5 7 9 11 13 15 17 19 21 IEC61000-3-2 n1/n2=1. 2 n1/n2=0. 8

Harmonic numbers i

_ac

(A)

0 0.5

1 1.5

2 2.5

3 5 7 9 11 13 15 17 19 21 IEC61000-3-2 n1>n2

n1<n2

0 0.5

1 1.5

2 2.5

3 5 7 9 11 13 15 17 19 21 IEC61000-3-2 n1>n2

n1<n2

Harmonic numbers

0 0.5

1 1.5

2 2.5

3 5 7 9 11 13 15 17 19 21 IEC61000-3-2 n1/n2=1. 2 n1/n2=0. 8

Harmonic numbers i

_ac

(A)

Fig. 3.13 Harmonic current at n1/n2=1.2 or n1/n2=0.8

3.2.4 Inductor L1

The inductor L1 is designed to provide the partially soft-switching function for diodes D1

and D2. When the current iN1 decreases, inductor L1 causes the current iN1 to linearly decrease to zero, and the diode D2 turns off without any switching-loss. Furthermore, the inductor L1

causes iN1 to increase linearly from zero, and the diode D1 turns on without switching-loss. To guarantee the partial soft switching, functions above, current iN1 must reduce to zero before switch S1 turns off.

Employing the voltage-second balance theorem in L1 for one switching cycle, equations (3.15)-(3.18) in mode M3 yields

) )(

( )

)(

(V_N₁−V_C₁ t₁_,_M₃−t₀_,_M₃ +V_L₁_,₃₋₄ = V_C₂ +V_N₁−V_C₁ t₃_,_M₃ −t₂_,_M₃ , (3.20) where t1,M3 - t0,M3=DTs, Ts is the switching period, and VC1=Vm.

Substitution of equation (3.20) gives the maximum L1 to guarantee the partial soft switching. Thus, L1 can be obtained from

2 3 1

2 1 1 2 2

3 2 3 1

3 1 3

1 2 2

2 3

2 3 2 1

1 2

C m o C

m o

C o m

V V n V

n n n n n

n n D n n n n D n

n D n V

V V

V n V

V n

n L n L

−

⋅

⎥−

⎦

− ⎤ + +

⋅

⎟⎟+

⎠

− ⎞

⋅

⎢⎣

⎡

⎜⎜⎝

⎛ ⋅ +

⋅

−

⋅ +

⋅

≤

(3.21)

Fig. 3.10 provides the designer a selection aid of L1 to guarantee the partially soft-switching function working to D1 and D2.

3.3 Design Procedures

The method for designing the circuit of control loop and determining the voltage stresses of components voltage for the proposed converter is similar to that for designing the conventional flyback converter. However, the transformer design needs more calculations and considerations. The design method for transformer is shown as following.

1) Windings turns ratio n1/n2/n3: The turn ratio n2/n3 can be obtained from equations

(3.1)-(3.3) by the substitutions of the given Vm,min, Vm,max, Vo, Dmax, and ωt₁_,_min, where Vm,min and Vm,max is the amplitude of minimum line voltage and maximum line voltage respectively, Vo is typical output voltage, Dmax is the maximum duty ratio, and 0.4≦

Dmax≦0.45. The corner angle ω^t₁_,_min,

0< tω ₁_,_min ≤π4 , can be obtained as long as Vm,min is chosen. The detailed steps for obtaining the turn ratio n2/n3 is depicted as follows:

(i) Let VC2 be limited under 420V at Vin=265V.

(ii) Assume that VC2 is proportional to Vin. Then VC2≒85×(420/265)=134.7V at Vin=85V.

(iii) Let Vo=48V, Dmax=0.4. Then substitution to equation (3.2) gives 48=134.7×(n3/n2)×[0.4/(1-0.4)], n3/n2≒0.5,

(iv) Substituting n3/n2=0.5 to equation (3.1) yields 0.9≦n1/n3.

(v) Let ωt₁_,_min=0.24, Vm,min≒120V, Vo=48V, Dmax=0.4, and n2/n3=2/1. Then the substitutions of all the data to equation (3.3) gives n1/n2≒1.2.

2) Inductance LN2, LN3: The output ripple voltage varies with the inductance value of Lm. Therefore, it is suggested to count the output ripple voltage to calculate Lm. The ripple current of output port is shown as following,

L t V i= ⋅∆ /

∆ .

Consider the steady state situation. The average output current is

)

and the output ripple current is

)

) 1 ( 2

) 1 ) (

( ²

3 I k

D k T V L

O s

N −

× +

−

= .

and

2 3 3 2

2 ( )

n L n L_N = _N ×

3) Series inductance L1: The inductance L1 can be yielded by putting above parameters in equation (3.21).

4) To confirm dB<Bmax : The maximum change value of magnetic flux density has to limit under maximum magnetic flux density for the selected magnetic material.

Given L_N₂⋅di_N₂ =N₂⋅dB⋅Ae or

Ae N

di dB L^N ^N

⋅

= ⋅

2 2

2 , where diN2 can be calculated by substituting DTs for (t-t0,M3) in equation (3.13), Ae is the effective area of the selected magnetic core.

) ] [(

2 2 2

1 1 1

max , 2 2

N C m

N N

N L

V n

L n V DTs V

di − +

≡ ; _max

2 2

2 B

Ae N

di L_N _N

⋅ <

⋅ .

3.4 Experimental Results

An experimental prototype has been established to demonstrate the circuit operation and the analysis results presented above. The experimental circuit can operate in 85V~265V/ac input voltage range and generate an output voltage of 48v/dc and an output power of 96W.

The turn ratio of n₁/n₂/n₃ is 2.38/2/1 and the inductance ratio of L₁/L_N1 is 0.17, where L1=30μ H and transformer core PQ32/20 is used. The transformer core employed in previous similar converter should be EER35 in [6]-[7]. Although some previous similar converters have similar transformer core size to the proposed converter for similar output power and switching frequency, the values of the boost inductors, 58μH~240μH in [6] or 1.4mH in [7], are several times greater than the value of L1 in the proposed converter. The sizes of the boost inductors employed in [6]-[7] thus are several times greater than that of L1 when flowing through a similar line current. The converters in [8] or [9] use similar smaller boost inductor 30μH, but the line current harmonic distribution is higher than that produced by the proposed converter. Table 3.1 shows that the detailed harmonic distributions of the experimental circuit using 230V line voltages are significantly below the levels required by class D.

, 1A/div

40ms/div

Fig. 3.15 Dynamic response waveforms for output voltage Vo, line current iac , output current Io. Ch1->Vo, Ch2-> iac, Ch3->

Io=0.75A/1.5A.

Fig. 3.14 Line current and line voltage waveforms, Vac=230V, Output=48v/2A.

Vac, 100V/div

Iac, 1A/div

Figure 3.14 shows the line current in a line-cycle, revealing that its harmonic distribution complies with a standard of IEC 61000-3-2. Figure 3.15 shows the dynamic response from a 3/8 to 3/4 full load in 110v/ac input voltage. The output voltage of prototype shows a fast response and stable regulation. Figure 3.16 shows the voltages across bulk capacitor for different loads from 85V to 265V line voltage. The experimental results demonstrate that the bulk capacitor voltage was ranged between 60~70% higher than the line voltage. Furthermore, a lower percentage can be achieved by carefully selecting the winding ratio. The voltage of the bulk capacitor is shown to depend on both Vac and turn ratio n2/n3, but irrelative to load current.

Figures 3.17 and 3.18 show the test system and prototype of proposed converter respectively. The test system contains Power Analyzer/Chroma 6630, AC power source, Electronic Load and oscilloscope.

0 100 200 300 400 500

85V 110V 150V 220V 265V

0.5A 1A 2A

Fig. 3.16 Voltage rating of bulk capacitor and line voltage Vac (rms)

VC2 (V)

Fig. 3.17 Test system picture

Fig. 3.18 The prototype of the proposed converter and test results at Vac=110V and 48V/2A output.

3.5 Extension Circuit

The voltage stress across main switch in primary side will be over 500V when the proposed flyback converter is applied in wide range input voltage up to 265V. The high voltage stress will cause two drawbacks, expensive cost in MOSFET and high switching loss in the same MOSFET. Therefore, this section will introduce an extension circuit to prevent the switch from the high voltage stress.

The voltage stress on main switch can be expressed as

3 Based on the prototype’s data, the maximum voltage stress Vds is 536V where VC2=440V, Vo=48V, and n2/n3=2.

Figure 3.19 is the extension circuit, twin-transistors type flyback converter, based on the proposed flyback circuit. The circuit adds a switch S2 and the switching operation is synchronous to S1. Except the additional switch, there is no difference in comparing to the proposed flyback converter. The additional transistor can share a half voltage-stress in the single transistor flyback converter. Therefore, the high voltage stress issue can be solved via the proposed twin-transistors flyback converter.

R_L

Fig. 3.19 The proposed twin-transistors flyback converter

CHAPTER 4 THE FORWARD CONVERTER USING THE PROPOSED INPUT CURRENT SHAPER

Figure 4.1 illustrates the proposed forward AC/DC converter with input current shaper and fast output regulation. The proposed circuit is a single-switch single-stage AC/DC converter, which comprises a single switch S1, an input filter C1, bulk capacitor C2, soft-switching inductor L1, and a transformer with two primary windings N1&N2. The winding N1, inductor L1 and diode D1&D2 form an input current shaper. The winding N1, inductor L1, diode D1&D2, switch S1, and bulk capacitor C2 form a boost circuit. Moreover, the windings N2&N3, a bulk capacitor C2, switch S1, diode D3, D4, inductor L2, and output capacitor C3 form a forward converter. The circuit connection of the reset winding N1 differs from that in the classical forward converter. In the proposed design, the reset winding N1 has two functions, to recycle the magnetic current generated by the winding N2, and also to form a magnetic feedback for shaping line current. Besides,

N₁N₂ N₃

Fig. 4.1 Proposed forward AC/DC converter with ICS

A turn ratio of n1/n2 can determine not only the corner angle of the line current, but also the voltage across a bulk capacitor C2. More detailed effects of turn ratio of n1/n2 are discussed in the following section. The inductor L1 provides a soft-switching function for diodes D1 and D2. When the current iN1 of L1 linearly reduces to zero illustrated in Fig. 6, D1

(in mode M1) or D2 (in mode M2) turn off switching loss. Additionally, the inductance and volume of L1 are significant, and are smaller than the primary windings N1 or N2 of the transformer.

The control circuit can be designed using either a simple fixed-frequency voltage mode control or a conventional peak-current mode control. The experimental results have

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