Fourth-Order Cascaded
61 Modulator Using
Tri-Level Quantization and Bandpass Noise Shaping
for Broadband Telecommunication Applications
Teng-Hung Chang, Student Member, IEEE, and Lan-Rong Dung, Member, IEEE
Abstract—This paper presents a 14-bit cascaded sigma-delta modulator for broadband telecommunication applications. The modulator is a 2-1-1 cascaded architecture that employs a res-onator-based topology in the first stage, three tri-level quantizers, and two different pairs of reference voltages. As shown in the ex-perimental result, for a 2.5-MHz signal bandwidth, the modulator achieves a dynamic range of 86 dB and a peak signal-to-noise and distortion ratio of 78.5 dB with an oversampling ratio of 16. The proposed modulator including reference voltage buffers and bandgap circuitry dissipates 62.5 mW from a 2.5-V supply. The active area is 1.2-mm2in a 0.25- m CMOS technology.
Index Terms—Analog-to-digital conversion, broadband telecom-munication, MASH, resonator-based topology, sigma-delta (61)modulation.
I. INTRODUCTION
T
HE growing trend in broadband telecommunication sys-tems, such as asymmetric digital subscriber line (ADSL), is energizing the demand for low-cost low-power high-perfor-mance analog-to-digital converters (ADCs). The newly released ADSL standard ADSL2+, for instance, requires a 14-bit ADC with a signal bandwidth over 2.2 MHz while considering power dissipation and silicon cost. How one chooses the ADC archi-tecture for optimizing the trade-off among power, resolution and bandwidth becomes a key issue.Pipelined and sigma-delta converters, among the ex-isting ADC architectures, are most suitable candidates to meet the requirements of such applications. Generally speaking, pipeline ADCs can achieve wide bandwidth, but have limited dynamic range, high power dissipation, and large silicon area. On the other hand, the state-of-the-art ADCs can achieve high dynamic range and high bandwidth with noise shaping and oversampling techniques. Papers [1], [2] have presented single-bit cascaded modulators for ADSL application, which achieved 15-bit resolution and 1.1-MHz signal band-width with an oversampling ratio (OSR) of 24. To further extend the signal bandwidth and reduce the OSR, several papers [3]–[7] have presented modulators with high-order multibit topologies. As shown in their results, they can achieve Manuscript received April 1, 2006; revised April 27, 2007. First published February 2, 2008; last published July 10, 2008 (projected). This work was supported in part by the National Science Council, R.O.C., under Grant NSC 95-2221-E-009-337-MY3. This paper was recommended by Associate Editor I. Bell.
The authors are with the Department of Electrical and Control Engineering, National Chiao Tung University, Hsinchu 30010, Taiwan, R.O.C. (e-mail: [email protected]).
Digital Object Identifier 10.1109/TCSI.2008.916450
13-bit (or above) and up to 2 MHz of signal bandwidth when the OSRs range from 8 to 24. However, the use of the multibit quantizer requires data weighted averaging (DWA) algorithms to solve the nonlinear problem of multibit digital-to-analog converters (DACs). The DWA circuits usually consume extra power (say 30 40 mW with 2.5-V supply) [6], [7] and cost additional silicon area. Paper [8], [9] instead of using DWA, uses the cascaded architecture with the single-bit quantizer in the first stage and multibit quantizer in the last stage to relieve the linearity requirement of feedback DACs. Ideally, they can achieve high dynamic range for low-OSR design. Yet, in practical, the leakage quantization noise of such architec-tures may limit achievable dynamic range, and the multibit quantizer of the last stage may increase circuit complexity and power dissipation. Recently, paper [10] presents a single-loop fifth-order modulator with a tri-level quantizer, which is pop-ular in pipelined ADCs, to achieve high dynamic range and consume low power for ADSL applications. Accordingly, we intend to take the advantage of the tri-level quantization and present a new topology for cascaded modulators to perform good trade-offs among power consumption, dynamic range and signal bandwidth.
Targeting on the specification of broadband telecommunica-tion system, say ADSL2+, three architectural approaches are applied for the proposed cascaded modulator. First of all, the structure of the proposed cascaded modulator is 2-1-1 where uses the tri-level quantizer and DAC in each stage. Comparing with single-bit quantizer, the tri-level quantizer can reduce the quantization noise and result in larger inter-stage gain and, hence, larger dynamic range. Since the tri-level DAC suppresses the nonlinearity of the DAC to the required performance, the proposed modulator does not need DWA algorithm. Secondly, to improve the signal-to-noise ratio (SNR) of the modulator for low-OSR design, we employ a resonator-based topology in the first stage. The resonator-based topology adds in-band zeros in the noise transfer function (NTF) and hence suppresses the quantization noise over the signal band [11]. Furthermore, by adding an input feed-forward path into the first stage, the swing ranges and distortions of integrator outputs are reduced [12], [13]. Although this arrangement raises objections due to the increased difficulty of analog-digital mismatch (i.e., leakage quantization noise), with properly scaling the coefficients of the resonator, the proposed modulator can still achieve more than 80-dB SNR in case the capacitor mismatch is 0.5% and the opamp dc gain is 70 dB. Finally, we employed two dif-ferent pairs of reference voltages for comparators and DACs to enhance the achievable dynamic range; a pair of 0.9 V is used 1549-8328/$25.00 © 2008 IEEE
Fig. 1. Resonator-based second-order 1.5-bit modulator.
in the first stage and another pair of 0.45 V in the succeeded stages. The proposed modulator has been realized in a 0.25- m CMOS technology. The experimental results show that the architecture, for a 2.5-MHz signal bandwidth, can achieve a dynamic range of 86 dB and a peak SNDR of 78.5 dB. It costs an active area of 1.2 mm and consumes 62.5 mW from a 2.5-V supply voltage.
This paper is organized as follows. Section II describes the system-level design and architectural decisions of the proposed modulator with nonideality analysis. Section III provides circuit details for the implementation of proposed modulator. Section IV reports the measured results of the fabricated chip. Finally, Section VI concludes our work in terms of chip fea-tures.
II. PROPOSEDARCHITECTURE
A. Design of Resonator-Based Modulator
The proposed architecture employs a resonator in the second-order stage of modulator. By adding in-band zeros, the res-onator can suppress the in-band quantization noise and improve SNR. Fig. 1 illustrates the proposed resonator-based, second-order modulator. It uses a low-Q resonator-based loop filter that introduces a pair of zeros into NTF. The NTF of the res-onator-based modulator can be expressed as
(1)
where , and . Note that is the
loop gain of the resonator and is the gain of the quantizer. The zeros of NTF create a notch around the edge of signal band and hence suppress the in-band quantization noise. The suppression can improve the SNR of modulators, especially for low-OSR de-sign [11]. The feedforward path from input to the adder followed by quantizer is used to suppress the signal swing and distortion caused by integrator nonidealities [12], [13]. Equation (2) shows the signal transfer function (STF) induced by the feedforward path
(2)
Based on (2), the error signal becomes
(3)
In (3), the term of can be treated as the impact factor of to . Obviously, the smaller the impact factor is, the less sensitive the loop filter is to input signal. When is unity, the does not have the component and the integra-tors of the loop filter will only process the quantization noise . In this case, the distortion caused by integrator nonideal-ities can be significantly reduced [12]. Unfortunately, cannot be unity in practical design because there always exists quan-tization error between input and output of a quantizer. In case of using the single-bit quantizer, is varying with the input signal of quantizer and its value can be much greater than unity. The variation of causes the sensitivity of loop filter to input signal to be high. To take the advantage of making unity, one can use the multibit quantizer to lower the sensitivity as much as possible [13]. Nevertheless, when using the multibit quan-tizer, the modulator requires the DWA algorithm to compensate the nonlinearity of the multibit DAC. Here we replace multibit quantizer with a tri-level one in that the variation of tri-level quantizer can be reduced and the perfomacne is improved as compared with single-bit quantizer. Although the tri-level DAC introduces distortion, the 14-bit linearity can be achieved by careful sizing of transistors composing the OTA and symmet-rical layout without using the DWA circuitry. The next subsec-tion will detail the design of tri-level quantizer.
B. Analysis of Tri-Level Quantization
The tri-level quantization technique has been proved to be able to achieve high dynamic range without using DWA circuitry [14]. In our work, the use of tri-level quantization is twofold: 1) to keep the impact factor of to lower than that of the single-bit quantizer so the performance can meet the requirement of ADSL applications, and 2) to have the nonlinearity of feedback DAC lower than multi-bit quantizers so that the DWA is not a necessity for high dynamic range. To explain why the tri-level quantizer outperform single-bit one, we first define the value of of tri-level quantization for a given threshold voltage as shown as
(4) where and are input and output amplitudes of tri-level quan-tizer. Based on (4), Fig. 2 illustrates the variation of with the input amplitude of tri-level quantizer when V. As mentioned above, the smaller the impact factor is, the less sensi-tive the loop filter is to input signal. In this case, the value of of tri-level quantizer ranges from 0.5 to 2 and the impact factor
Fig. 2. Nonlinear gain of tri-level quantizer. (V = 0:45 V).
Fig. 3. (a) Transfer characteristic. (b) Error function of tri-level DAC. (The figures are excerpted from [14]).
from 0.5 to . However, for the single-bit quantizer, the value of ranges from 0 to infinity and the impact factor from 1 to . Note that the infinity comes from the fact that the single-bit quantizer does not have a zero-level. If we consider the output levels of the quantizers between 1 and 0, the of tri-level and single-bit quantizers with input level of 0.4 are 1.25 and 2.5, respectively. Therefore, the values of impact factor, , of tri-level quantizer are smaller than those of the single-bit quan-tizer.
The characteristic and nonlinearity of a tri-level DAC can be briefly described in Fig. 3(a). The horizontal axis represents the digital input codes while the vertical axis represents the analog output levels, which can vary due to circuit nonidealities. The nonlinearity is then defined by the displacement, , of medium level. With an approximation by using quadratic polynomial as shown in Fig. 3(b), the nonlinear error function of tri-level DAC can be expressed as [14]
(5) According to the analysis of paper [14], the well-sizing of the transistor of OTA and symmetrical layout will result in an of 0.01%, which enables 14-bit linearity.
To verify if the use of tri-level quantizer can reduce the har-monic distortion caused by integrator nonlinearities, we applied single-bit quantizer and tri-level quantizer for the
resonator-Fig. 4. The simulated 3rd harmonic distortion of resonator-based modulator with tri-level and single-bit quantizers. (V = 0:45 V, n = 500 kHz, OSR = 16; fs = 80 MHz).
based modulator. Given V, , and
MHz, Fig. 4 plots the third harmonic distortion results for 500-kHz sinusoid inputs. In the simulation, each integrator has a finite closed-loop pole at 240 MHz. The closed-loop pole fre-quency is three times of the sampling frefre-quency . As shown in Fig. 4, the third harmonic distortion of resonator-based mod-ulator with tri-level quantizer is lower than dB. So, its spu-rious-free dynamic range (SFDR) can be greater than 90 dB. Comparing with the single-bit version, the use of tri-level quan-tizer lowers the sensitivity of the loop filter to input signal and consequently reduces the distortion caused by integrator setting error.
C. Design of Fourth-Order Cascaded Modulators
The mostly popular fourth-order cascaded architectures are MASH 2-2 and MASH 2-1-1. An in-depth study on compar-ison of 2-2 and 2-1-1 cascaded architectures has been presented in [15]. Fig. 5 illustrates the block diagram of two resonator-based cascaded modulators, which are referred as RMASH - - and RMASH - . The first stage of the RMASH - - , as mentioned above, uses the resonator-based mod-ulator and the following two stages are the conventional first-order modulators. The and in Fig. 5(a) denote the digital error cancellation logic, which are used to cancel the first-and second-stage quantization noise of RMASH - - . The transfer functions of and are deduced
(6) With ECL, the NTF of RMASH - - is given by
(7) where is the quantization noise of tri-level quantizer in the third stage and is the inverse of . Obviously, the SNR can be improved by tuning parameters, , and . With the careful selection of the resonator loop gain , the NTF zeros can produce a notch near the edge of the signal band to suppress the quantization noise over the desired signal band.
Fig. 5. Block diagrams of (a) RMASH2-1-1 . (b) RMASH2-2 . According to [16], the NTF zeros are placed at the corner fre-quency of the signal band, and the value of can be chosen by the following expression.
(8) At the OSR of 16, with in-band zeros, the four-order NTF can improve the SNDR by 14 dB.
Furthermore, we applied two pairs of reference voltages (TPRVs) for the proposed modulator; a pair of 0.9 V for the tri-level quantizer of first stage and another pair of 0.45 V for the quantizers in the second and third stages. The high voltage pair can have the first stage operate at high dynamic range while the low voltage pairs can reduce the output swing of the second stage and the power of .
Based on (6), the theoretical quantization noise (TQN) can be expressed as
(9) where is the step size of tri-level quantizer in the third stage. To reduce the TQN of the modulator, we cut down on
para-ments and . Because of , we first
re-duced the output swing of the second stage. The reduction im-plies that the integrator gain of can be increased and hence decreases . Then, we halved the reference voltage of third stage to reduce the step size of tri-level quantizer. Thus, with the proposed TPRVs, our work can have additional 12-dB SNR gain. The use of TPRVs does not affect the NTF and ECL equa-tions. Although the SNR of the modulator is enhanced, this ar-rangement increases the effect of the leakage quantization noise, caused by capacitor mismatching and finite OTA dc gain. For-tunately, the desired SNR still can achieve 90 dB with 0.1% of capacitor mismatching and 70 dB of OTA dc gain. Using the same design methodology, the RMASH - can achieve similar performance to RMASH - - . One can also apply the second-order resonator-based modulator for stages of the
RMASH - .
Fig. 6. TQN plots of RMASH 2-1-1 with TPRVs, MASH2-1-1 without TPRVs, and RMASH2-1-1 without TPRVs.
Given the OSR of 16 and the of 2, Fig. 6 shows the NTFs of the proposed RMASH - - with TPRVs, conventional MASH - - without TPRVs, and RMASH -without TPRVs. We can see the performance gain of 20 dB in the shadow region. Fig. 7 illustrates the SNR versus OSR in comparison of the fourth-order modulators using tri-level quan-tizer. To achieve the SNR of 90-dB (or higher), the single-loop architecture needs to operate at the OSR of 32 while the others need the OSR of 16. The lower OSR implies the lower power dissipation. To optimize the load of the second integrator of first-stage modulator and trade off the dynamic range of the opamp in the following stages, here we adopt the MASH
- - as our design target.
In general, the leakage quantization noise is the major con-cern on the design of a cascaded modulator. The leakage quantization noise is mainly caused by the finite OTA gain and capacitor mismatching of the first-stage integrators. In the RMASH - - , the capacitor mismatching is especially critical because the SNR improvement mainly relies on the in-band zeros induced by the first-stage resonator. Therefore,
Fig. 7. Peak SNDR versus OSR plots for four different modulators.
Fig. 8. Circuit implementation of the first integrator in the first stage of RMASH2-1-1 .
we have to select coefficients of the loop filter in the first stage carefully to achieve the desired SNR. Fig. 8 shows a circuit implementation of the first integrator where the
and . Given the notch and sampling frequencies of 2.25 and 80 MHz, the resonator loop gain , based on (8), approximates to 0.03125. In general, the and should be as large as possible to keep the thermal noise low, and thus has to be relatively small for the given . Conse-quently, the difference between and may become large and cause a large ratio mismatching [17]. For instance,
if , and pF, and will be
0.0625 and 93.75 fF, respectively. The value of is much larger than that of . To reduce the mismatch between and while keep transfer function unchaged, we scaled the
, and to 0.5, 0.25, and 0.25, respectively. In this case, the becomes 375 fF and the mismatch can be reduced. However, the integrating capacitor, , should be scaled up to 3 pF to keep the thermal noise contribution unchanged. This increases chip area. Table I shows the coefficient sets of the RMASH - - used in this work, and the proposed RMASH - - can achieve a peak SNR of 92 dB and a dynamic range of 95 dB, respectively.
TABLE I
COEFFICIENTS OFRMASH2-1-1 FOROSR = 16
Fig. 9. Plots of simulated SNDR versus OTA dc gain.(OSR = 16).
D. Integrator Nonidealities
In switched-capacitor (SC) cascaded modulators, the capac-itor mismatching and finite OTA gain are two main static non-idealities. These nonidealities can lead to imperfect cancella-tion of the first- and second-stage quantizacancella-tion noises at mod-ulator output and thus degrade the performance. In addition, in the RMASH - - , the nonlinearity of tri-level DACs can also introduce distortion to the modulator output, especially for the first-stage DAC. Paper [2] gives the practical model of a non-ideal integrator for behavioral simulation to determine the min-imum requirement of OTA gain. By applying their model for our simulation, Fig. 9 shows that increasing the finite OTA gain above 70 dB will not significantly improve the performance. So, in our work, we set the finite OTA gain to 70 dB. Given 70-dB OTA gain and 0.5% capacitor mismatching, Fig. 10 illustrates that the RMASH - - can achieve a peak SNR of 86 dB.
E. Nonidealities of Tri-Level DAC
The paper [14] has provided a good study and insight for the the nonidealities of tri-level DACs. The first integrator of the tri-level DAC is implemented as shown in Fig. 8. There are three major nonidealities in the tri-level DAC. They are the mismatch of sampling capacitor , the mismatch of integrating capacitor
, and the input-referred OTA offset .
One may claim other nonideality sources, such as charge in-jection, parasitic capacitors, and so on. However, these sources are negligible with careful design and layout schemes. To an-alyze the distortion of the tri-level DAC, we first calculate the
Fig. 10. Plots of simulated SNDR versus capacitor mismatching(OSR = 16). output voltages for three feedback signals, , and
(10) (11)
(12)
where and .
Accordingly, the nonlinearity factor defined in (5) can be cal-culated as
(13)
By assuming and and by
substi-tuting (10), (11), and (12) in (13), we can find that
(14) Since the and are small, the , first term of (14), can be considered negligible. By substituting (14) in (5), the discrete-time error function of tri-level DAC approximates
(15) where , and denote the offset voltage of OTA, reference voltage and output of tri-level quantizer, respectively. As shown in (15), the nonideality of tri-level DAC mainly depends on the offset voltage of OTA for a given reference
Fig. 11. Plots of simulated dynamic range versus OTA offset voltage.(OSR = 16).
voltage. Fig. 11 shows the simulated dynamic range of RMASH - - as a function of offset voltage of OTA. To target the dynamic range of 90 dB, the offset voltage of 5 mV is required. The offset voltage of 5 mV is achievable with carefully sizing and paying attention to layout of the input differential pairs of OTA.
III. CIRCUITIMPLEMENTATION
A. Specifications for Building Blocks
Upon simulating the behavior of the proposed modulator, we determined the specifications of analog building blocks with consideration of power-performance trade-offs. Table II sum-marizes the circuit specifications for the 14-bit 5-MS/s RMASH - - . To evaluate the robustness, we modeled the critical circuit parameters as Gaussian distribution with the standard de-viation of 20% and executed the Monte Carlo analysis by using MATLAB. The critical parameters are dc gain, transconduc-tance, unity gain bandwidth (GBW), output current, and input offset of OTAs, on-resistance of switches, and clock jitter. In addition, the standard deviations of integrator and feedforward gains are 0.5% and 2%, respectively. As shown in Fig. 12, the mean, minimum, and standard deviation of peak SNDR are 84.6 dB, 81.5 dB, and 0.72%, respectively.
B. OTA Circuit
According to Table I, the proposed modulator requires the dc gain of 75 dB and the GBW of 300 MHz for the OTA with a supply voltage of 2.5 V. To meet the requirements, we chose a folded-cascode OTA with additional gain-boosting amplifier. By carefully design and sizing of the gain-boosting amplifiers, the induced nondominant pole can be located at 1 GHz. As men-tioned in Section II, the output swing of the first-stage integra-tors can be reduced by using the tri-level quantizer and the input feedforward path. Hence, for a supply voltage of 2.5 V, the re-quired single-ended output swing of OTA is approximately 1 V. Fig. 13 shows the schematic of the folded-cascode OTA being used for the first-stage integrators. The single-ended output
TABLE II
CIRCUITSPECIFICATIONS FOR14-BIT5 MS/SRMASH2-1-1
Fig. 12. Plots of simulated SNDR versus input level with 30 Monte Carlo anal-ysis runs.
swing of OTA is 1.4 V, which can sufficiently accommodate the required output swing at the integrator outputs, and a dc gain of 75 dB is accomplished over the entire output range. The OTA, including gain-boosting and biasing circuits, dissipates 15 mW from a 2.5-V supply and achieves a GBW of 300 MHz with a capacitive loading of 5 pF, while the phase margin is 75 degree. The total thermal noise contribution over 2.5-MHz signal bandwidth is about 12.6 V. The SC common-mode feedback (CMFB) is used for designed OTA because it does not dissipate the static power. The capacitors used in the CMFB circuitry are properly chosen to maximize the gain bandwidth, and thus avoid the settling error. We also used the similar OTA for other stages, and it dissipates 0.7 times of the power consumption the first-stage OTA consumes.
Fig. 13. Circuit schematic of the OTA.
C. First-Stage Circuit
Fig. 14 illustrates the circuit diagram of the first-stage modu-lator. Since the dynamic range of the modulator is targeted at 90 dB at the sampling rate of 80 MHz, the sampling capac-itor is chosen to be 1.5 pF and, accordingly, the integrating and resonator feedback capacitors are 3 and 0.375 pF, respectively. The closed-loop bandwidth of front-end integrator is about 255 MHz, which is larger than three times the sampling frequency. Because the feedback gain of the tri-level DAC is equal to one, we can use the share-capacitor switching technique to elimi-nate coefficient mismatch. The share-capacitor switching tech-nique is to have the input sampling and feedback DAC share a common sampling capacitor [18]. However, the dependent load on the reference voltage may cause harmonic distortion. In our work, we used a dummy SC network to reduce the distor-tion [19]. The output two-bit code of tri-level quantizer is used to switch , and at the integrating phase.
The summing circuit in front of the quantizer is implemented by using a passive SC network to avoid the use of additional OTA and save the power dissipation. The summed signal can be expressed as
(16) where , and are the capacitors for ward gains. According to behavioral simulation, the feedfor-ward gains are not critical and can tolerate the variation up to 2%. This allows the use of small capacitance to implement the feedforward gains. We set the values of , and to 0.125, 0.25, and 0.5 pF, respectively. Note that the summed signal is scaled down by 1/7 when comparing with the parame-ters of Table I and Fig. 1. In order not to affect the desired per-formance of the modulator, the reference voltages of the quan-tizers must be scaled down by a factor of 1/7 from the nominal value. This also scales down the quantizer step size, and hence increases the requirement of the comparator resolution. In our case, the required step size of the tri-level quantizer is about 150 mV. This requirement is feasible because, in practice, the
Fig. 14. Circuit implementation of the first-stage modulator.
CMOS comparator with preamplifier can provide a resolution better than 50 mV.
D. Tri-Level Quantizer Circuit
The circuit diagram of tri-level quantizer is shown in Fig. 15. As mentioned above, the SC network must scale down the ref-erence voltages, VR+ and VR-, by a factor of 1/7. So, we set the values of the capacitors and to 0.125 pF and 0.75 pF, respectively. In our design, we used a high-speed, high-accu-racy CMOS comparator with preamplifier which is presented by [20]. The clock is used to control the generation of , and . Because of the time-delay of AND gates, the nonover-lapping interval of is limited to 1–2 ns when a sampling rate of 80 MHz.
E. Reference Buffer Circuit
Since the first integrator samples the input signal, it draws a signal-dependent current from the voltage reference and intro-duces harmonic distortion due to the finite output impedance of reference buffers. Our design requires a high-bandwidth refer-ence voltage buffer. The referrefer-ence buffer must have fast settling behavior during the integrating phase . However, such kind of reference buffer is power-consuming, especially for high signal-bandwidth design. In this work, we used a class-A amplifier with an external capacitor to make output impedance low enough to meet desired linearity requirement [21]. The schematic of the class-A amplifier is shown in Fig. 16. The output impedance of the amplifier increases with frequency due to its finite gain-bandwidth product. Because of the large external capacitor, the output impedance remains low in high-frequency band. That is, the transient charge of the reference voltage is almost delivered from the external capacitor. Fig. 16(b) shows the decoupling scheme of TPRVs used in the printed circuit board (PCB). The decoupling capacitors and are important because
Fig. 15. Circuit implementation of tri-level quantizer.
they can reduce high-frequency common-mode noise between the differential reference voltages. By using this topology, the reference buffer in first stage consumes only 5 mW at the cost of using large external decoupling capacitors. The buffers for the reference voltages of the second and third stages are sim-ilar to Fig. 16, but dissipate approximately half of that of the first stage. This arrangement is practical because the settling er-rors of reference voltages in the second and third stages will be suppressed by the high-pass noise shaping of and , respectively.
Fig. 16. (a) The schematic of the voltage buffer amplifier. (b) The off-chip decoupling scheme of reference voltages.
Fig. 17. Chip micrphotograph.
IV. EXPERIMENTALRESULTS
The modulator is designed for a sampling rate of 80 MHz and a fixed OSR of 16; so, the signal bandwidth is 2.5 MHz. The signal bandwidth is slightly larger than the requirement of ADSL2+ (say 2.2 MHz) because we intend to reduce the cost of the decimation filter. Increasing the signal bandwidth from 2.2 to 2.5 MHz makes the required order of decimation filter lower. The modulator was fabricated in a 0.25- m 1P5M CMOS tech-nology with metal–insulator–metal (MIM) capacitors. In the ex-periment, to fairly compare with existing wide-band mod-ulators, we collected the results without counting the decima-tion filter part. The active area of the modulator is 1.2 mm2. The power dissipation of the modulator with I/O pads is 62.5
Fig. 18. Plots of measured SNDR and SNR versus input signal level.
Fig. 19. Measured output PSD of RMASH2-1-1 operating at 80-MHz sampling rate.
mW with a 2.5-V supply while the core of modulator consumes 53 mW. Fig. 17 shows the chip microphotograph in which the experimental modulator includes the clock generator, reference buffer, and bandgap circuitry. To measure the performance data, the chip was mounted onto a four-layer printed circuit board to separate the analog signal from digital signal and hence re-duce the coupling. The input clock is generated from an external 80-MHz low-jitter crystal with independent power supply volt-ages to avoid the switching-noise coupling. Fig. 18 shows that the modulator can achieve a dynamic range of 86 dB, a peak SNR of 83 dB, and a peak SNDR of 78.5 dB for a 2.5-MHz signal band.
Fig. 19 illustrates the unfiltered power spectral density (PSD) of the measured output signal; the input sinusoid is dB and 268 kHz. The result implies that the spurious-free dynamic range (SFDR) is 93 dB. When the modulator operates at a sampling rate of 100 MHz, as shown in Fig. 20, it still stay functional without significant harmonic distortion and can achieve a SFDR of 90 dB. However, because of the switching activity of the digital output buffers, the in-band noise floor
Fig. 20. Measured output PSD of RMASH2-1-1 operating at 100-MHz sampling rate.
rapidly increases at the pad-pin level, and the dynamic range and peak SNDR become 77.5 and 72 dB, respectively, for a 3.125-MHz bandwidth.
To quantitatively evaluate the efficiency among power dis-sipation, dynamic range, and conversion rate, we use the for-mulas for the effective number of bits (ENOB) of ADC and the figure-of-merit (FOM) as shown below [9], [22]
(17) Fig. 21 shows the FOM distribution of our work and existing wide-band ( MHz) SC modulators. Table III summa-rizes the measured performance and specifications of the pro-posed modulator, and compares with the other wide-band SC
modulators as well.
V. CONCLUSION
This work addresses a low-power low-cost cascaded modulator architecture appropriate for ADSL2+ application. This work uses three approaches to improve performance and reduce power consumption: using the resonator-based topology, applying the tri-level quantization, and using two
Fig. 21. FOM distribution of wide-band SC61 modulators with respect to conversion rate.
pairs of reference voltages. The proposed modulator has been fabricated in TSMC 0.25- m 1P5M CMOS technology. In-cluding reference voltage buffers and bandgap circuitry, the chip dissipates 62.5 mW from a 2.5-V supply. As shown in the experimental result, the designed RMASH - - can achieve a peak SFDR of 93 dB, a dynamic range of 86 dB, a peak SNDR of 78.5 dB, and a low FOM of 0.77 pJ/conv with the OSR of 16 and bandwidth of 2.5 MHz.
ACKNOWLEDGMENT
The authors would like to acknowledge Trendchip Technolo-gies Corporation for fabrication of the prototype chip. They also thank Dr. L.-C. Yin, Dr. J.-Y. Guo, and Dr. H.-H. Li for their valuable discussion during layout and design phases, as well as C.-Y. Hsu and H.-C. Tseng for testing support.
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Teng-Hung Chang (S04) was born in 1976. He
received the M.S. degree from the Department of Electrical Engineering, Tamkang University, Taipei, Taiwan, R.O.C., in 2001. He is currently working toward the Ph.D. degree in the Department of Electrical and Control Engineering, National Chiao Tung University, Hsinchu, Taiwan, R.O.C.
His research interests are high-performance analog-to-digital converters and low-voltage analog circuit design.
Mr. Chang received the Best Student Award for his M.S. degree.
Lan-Rong Dung (SM’93–M’97) received the
B.S.E.E. degree from Feng Chia University, Taichung, Taiwan, R.O.C., in 1988, the M.S. degree in electronics engineering from National Chiao Tung University, Hsinchu, Taiwan, R.O.C., in 1990, and the Ph.D. degree in electrical and computer engineering from Georgia Institute of Technology, Atlanta, in 1997.
From 1997 to 1999, he was with Rockwell Sci-ence Center, Thousand Oaks, CA, as a Member of the Technical Staff. He joined the faculty of National Chiao Tung University in 1999 where he is currently an associate professor in the Department of Electrical and Control Engineering. His current research in-terests include VLSI design, digital signal processing, hardware-software code-sign, and system-on-chip architecture.
He received the VHDL International Outstanding Dissertation Award cele-brating in Washington, DC, in October, 1997. He is a member of IEEE Circuits and Systems and Signal Processing societies.
