IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 17, NO. 8, AUGUST 2007 589
A Wideband Bandpass Filter With Wide Upper
Stopband Using Stepped-Impedance
Cascadable 180 Hybrid Rings
Chun-Hsiang Chi, Student Member, IEEE, and Chi-Yang Chang, Member, IEEE
Abstract—This letter proposes a wideband bandpass filter by
cascading two 3-dB stepped-impedance cascadable 180 hybrid rings with a pair of stepped-impedance lines. Thanks to the stepped-impedance lines, a broad upper stopband is achieved. The stepped-impedance vertically installed planar (VIP) coupler is used to implement the ideal 180 phase inverter and crossovers. The experimental results show that this 2-GHz center frequency, fourth order filter has a 10-dB return loss bandwidth of 92.5% and upper stopband rejection levels of better than 20 dB up to 6.8 GHz. This proposed filter achieves wide passband and broad stopband performance simultaneously.
Index Terms—Phase inverter, stepped-impedance, vertically
in-stalled planar (VIP) coupler, wideband bandpass filter (BPF).
I. INTRODUCTION
I
N recent years, the design of a wideband bandpass filter (BPF) is an important issue for the wideband communica-tion systems. This type of filter needs not only wide bandwidth to meet the required large data rate but also broad stopband to suppress harmonics. Based on the insertion loss method [1], a wideband BPF function of Chebyshev type can be easily syn-thesized. However, to realize a wideband BPF with the conven-tional microstrip half-wavelength parallel coupled-line struc-ture, very small gaps between coupled strips are required to achieve a tight coupling. This may cause some difficulties in the fabrication processes. Moreover, due to the unequal even-and odd-mode phase velocity, this filter has a first spurious pass-band at 2 , resulting in poor harmonic suppression. To improve bandwidth performance, several methods [2]–[4] have been pro-posed. In [2], cascading of two cascadable 180 hybrid ring cou-plers could form a wideband filter. In [3], a multiple-mode res-onator (MMR) is used to construct a wideband filter. In [4], a mi-crostrip ring BPF with dual stopbands below 3.1 GHz and above 10.6 GHz was built up. However, all of the above-mentioned fil-ters show narrow upper stopband performance and some of them even show notch-like upper stopband response. Therefore, it is still a challenge to design a single filter having wide passband and broad stopband characteristics simultaneously.In this letter, we modify the cascadable 180 hybrid ring in [5] with a stepped-impedance structure [6] to design a wideband
Manuscript received February 8, 2007; revised March 22, 2007. This work was supported by the National Science Council of Taiwan, R.O.C., under Grant NSC 95-2221-E-009-042-MY3.
The authors are with the Department of Communication Engineering, National Chiao Tung University, Hsinchu 300, Taiwan, R.O.C. (e-mail: [email protected]; [email protected]).
Digital Object Identifier 10.1109/LMWC.2007.901771
Fig. 1. Schematics of the proposed wideband filter: (a) basic structure and (b) modified structure whereY = 1=Z , i = 1, 2 3, are normalized values.
BPF so that the above-described shortcomings can be solved. This filter with the stepped-impedance structure acquires not only size reduction, but also better upper stopband clearance. The experimental results show good agreements with the simu-lated results.
II. ANALYSIS ANDDESIGN OF THEPROPOSEDBPF The basic structure of the proposed BPF is shown in Fig. 1(a) where the electrical length of each transmission line is 90 and the admittances and are normalized values with re-spect to system admittance . Note that the filter configuration shown in Fig. 1(a) is different from the filter in [2] where, in [2], the second rat-race ring is flipped with respect to horizontal central line of Fig. 1(a). From [5], at the center frequency, the
-parameters of the circuit shown in Fig. 1(a) is
(1) When , the signal excited at port 1 is equally divided between 3 and 4 and no power reaches 2 . The signal arriving at 3 and 4 are out of phase. Then, the signal at 3 is equally divided between 3 and 4 and the signal arriving at 3 and 4 are out of phase. The signal at 4 is equally divided between 3 and 4 and the signal arriving at 3 and 4 are in phase. Therefore, all of the power excited at port 1 will be delivered to port 4 and no power reaches port 3 and port 2 (corresponding to a 0-dB coupler). This means that a bandpass response can be obtained by cascading two 3-dB 180 hybrid rings. With the design method described in [5], the bandwidth of this filter cor-responding to specific return losses can be obtained. Here, to
590 IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 17, NO. 8, AUGUST 2007
Fig. 2. Simulated results of the proposed wideband filter: (a) basic structure and (b) modified structure.
increase the order and the bandwidth of the proposed filter, a 90 transmission line of impedance can be added between two 180 hybrids [7], [8] as shown in Fig. 1(b).
The simulated results of the filters in Fig. 1 with 15-dB re-turn loss are depicted in Fig. 2 that they correspond to third
53.31 and fourth ( 58.65 and 41.16 ) order filter respectively. Although wide passband can be easily achieved, the first spurious passband at 3 results in poor upper stopband performance. In [9], the fre-quency of the first spurious response of the 180 hybrid ring can be moved using the stepped-impedance structure by adjusting its structural parameters. Thus, by applying the stepped-impedance structure to Fig. 1(b), a wideband filter with broad upper stop-band characteristics can be obtained.
A quarter-wave transmission line of impedance Z is shown in Fig. 3(a) and it serves as a basic building block of the pro-posed BPF. The quarter-wave transmission line can be replaced by a stepped-impedance structure shown in Fig. 3(b) where and are the characteristic impedances of the cascaded sec-tions and and are the corresponding electrical lengths. By choosing , the overall electrical length of the stepped-impedance section can be shorter than 90 . Equating the ABCD matrix of the quarter-wave transmission line and the stepped-impedance structure shown in Fig. 3, we have
Z Z cos cos 2 0(Z +Z ) cos sin sin Z Z
=0 (2) Z cos sin +Z (0Z sin sin +Z cos sin 2 )
Z
0Z = 0 (3) Z cos sin +Z (0Z sin sin +Z cos sin 2 )
Z Z
01 Z=0: (4)
The first step to implement the proposed wideband stepped-impedance BPF is to specify the required return loss and to ob-tain the corresponding circuit parameters in Fig. 1(b). This can be easily done using optimization process provided in most cir-cuit simulators because only two circir-cuit parameters (
Fig. 3. (a) Quarter-wave transmission line and (b) stepped-impedance circuit equivalent to a quarter-wave transmission line.
TABLE I
PARAMETERS OF THESTEPPED-IMPEDANCEBPF
Fig. 4. Cross-sectional view of the VIP coupler.
TABLE II
COMPUTEDRESULTS BYHFSS(f = 2 GHz)
58.65 and 41.16 in this example) need to be deter-mined. The next step is to employ the stepped-impedance struc-ture to each line section in Fig. 1(b). Followings show the pro-cedures. First, arbitrarily choose the and and then use (2)–(4) to find and . Second, check the obtained and to be realizable or not, if not, redo the first step again. Finally, fine-tuning the obtained circuit parameters might be required to approach the equal ripple response in the passband. The opti-mized circuit parameters are shown in Table I. It should be em-phasized that previously described procedures are based on an ideal 180 phase inverter.
III. REALIZATION ANDMEASUREMENT OF THEPROPOSEDBPF Following the above-described design steps, the circuit pa-rameters of the proposed BPF can be obtained. However, the realization of an ideal 180 phase inverter on microstrip cir-cuits would be a problem. Fortunately, this can be solved by using an opposing ends short-circuited 4 coupled line (we simply call it “short-ended coupled line“) [10] with the verti-cally installed planar (VIP) structure [11]. Therefore, the ideal 180 phase inverter and crossovers, shown in Fig. 1, can be implemented by a 4 short-ended coupled line with the VIP structure [5]. Then, we employ stepped-impedance structure to the 4 short-ended VIP coupler. To simplify this design, the short-ended stepped-impedance VIP coupler is approximately
CHI AND CHANG: WIDEBAND BPF WITH WIDE UPPER STOPBAND 591
Fig. 5. Geometry of the fabricated filter (a) main circuit sectionW = 101 mil, W = 12 mil, W = 135 mil, W = 17 mil, S = 82 mil, L = 158 mil, L = 283 mil, L = 200 mil, L = 195 mil, L = 201 mil and (b) VIP stepped-impedance coupler.W = 133 mil, W = 29 mil, L = 184 mil, L = 296 mil.
Fig. 6. 3-D structure of the proposed wideband filter with the stepped-impedance structure.
Fig. 7. Measured and simulated results of the proposed wideband filter with the stepped-impedance structure.
equivalent to a balanced stepped-impedance transmission line with 180 of twist (physically 180 of twist equivalent to an ideal 180 phase inverter). This equivalence is valid as long as the even- to odd-mode impedance ratio is large. The charac-teristic impedance of each balanced line section approximately equals to 2 . Its physical length can be de-termined by the effective odd-mode dielectric constant because the signal going through it is mainly odd-mode. Fine tuning the length of each VIP line section to fit the insertion phase of other three step impedance lines at the center frequency may be required. Fig. 4 shows the cross-sectional view of the VIP coupler. Both the main circuit and the VIP couplers are imple-mented on a RO4003 substrate with thickness ( and ) of 20-mil and dielectric constant ( and ) of 3.38. The filter
is designed to operate at the center frequency of 2 GHz. Based on the above-described method, the characteristic impedances and effective dielectric constants for even- and odd-modes of the VIP coupler can be obtained by the 3-D EM simulator (An-soft HFSS) as given in Table II. Fig. 5 indicates the geometry of the fabricated filter. The 3-D structure of the proposed wide-band filter is shown in Fig. 6 where port 2 and port 3 are the input and output port, respectively. Both port 1 and port 4 are isolated ports and are terminated with a 50 load.
Fig. 7 shows the simulated and measured results of the pro-posed filter and the measured insertion loss is approximately 0.4 dB. Note that the simulated results of the proposed filter are sim-ulated with the computed parameters in Table II by the circuit simulator (AWR Microwave Office). The measured 10-dB re-turn loss bandwidth is from 0.95 to 2.8 GHz (92.5%). The stop-band rejection is better than 20 dB from 3.3 to 6.8 GHz. The measured bandwidth is a little narrower than the simulated re-sponses and the experimental stopband rejection is a little worse than the simulated results. This may be caused by the junction effect (especially junctions at microstrip and VIP coupler) and circuit fabricating imperfections.
IV. CONCLUSION
In this letter, by cascading two 3-dB 180 hybrid rings with a pair of 90 transmission lines, a stepped-impedance wide-band BPF has been designed and fabricated to exhibit wide passband and broad stopband performance simultaneously. The short-ended step impedance VIP coupler has provided a good phase inverter as well as a crossover in this realized filter. In addition, the bandwidth can be further enhanced by cascading more sections of 90 transmission lines in this proposed struc-ture. The experimental results are in good agreement with the simulated results.
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