A novel phase-noise-suppressed and delay-time-tunable
mode-locked erbium-doped fiber laser
Gong-Ru Lin
*, Yinchieh Lai
Institute of Electro-Optical Engineering, National Chiao Tung University, 1001 Ta Hsueh Rd., Hsinchu 300, Taiwan, ROC Received 07 November 2001; received in revised form 06 August 2002; accepted 26 August 2002
Abstract
A phase-noise-suppressed and delay-time-tunable actively mode-locked erbium-doped fiber ring laser is demon-strated by using voltage-controlled and frequency-discriminated phase shifter as an inherent timing module. By changing the voltage from)3.35 to 3.6 V, the maximum tuning range of phase and delay-time of 500 MHz optical pulse-train from the laser can be up to 340° (1.9p) and 1.9 ns (nearly 1 period), respectively. Optimized responsivity and tuning deviation are 0.27 ps/mV and 6 4%. The single-sided-band phase the laser operated at feedback controlled mode is significantly reduced to)100 dBc/Hz (measured at frequency of 10 kHz offset from center), which is at least 10 dB smaller than that of the laser operated at free-running mode.
Ó 2002 Published by Elsevier Science B.V.
Keywords: Erbium-doped; Fiber laser; Mode-locking; Phase noise; Phase-shift; Delay-time; Frequency-discriminated
Harmonic mode-locking has been considered as one of the promising techniques to construct high-repetitive optical pulse laser sources with relatively short pulsewidth [1–3]. It has been demonstrated that the mode-locked erbium-doped fiber lasers (EDFLs) based on this technique can provide stable and nearly transform-limited picosecond optical pulses with very low timing jitter [2]. These active harmonic mode-locked EDFLs have found versatile applications in ultra-high bit-rate optical transmission, high-speed optical time division
multiplexed (OTDM) [4], as well as electro-optical sampling (EOS) [5] at the wavelength range around 1:55 lm [6–11]. The typical optical pulse-width of these lasers operated at 10 GHz is around ps [6] and subpicosecond pulses can be obtained by using a dual-modulator scheme [7] or by external pulse compression techniques [8,9]. On the other hand, for the applications in phased-array an-tenna, wireless/fiber OTDM communication and optoelectronic sampling systems, the capability of continuously tuning the phase of microwave sig-nals or optical clock via optical or optoelectronic techniques has been the main goal of many ex-tensive research studies. The photonic microwave techniques can greatly improve the performance of the mechanical- or electronic-type microwave
Optics Communications 212 (2002) 169–175
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*
Corresponding author. Tel.: +886-3-5712121 ext. 56376; fax: +886-3-5716631 or +886-2-28281873.
E-mail address:[email protected](G.-R. Lin).
0030-4018/02/$ - see front matterÓ 2002 Published by Elsevier Science B.V. PII: S 0 0 3 0 - 4 0 1 8 ( 0 2 ) 0 1 9 6 4 - 8
phase shifters developed in the early stage, which may be bulky, incapable of phase-tuning, and with narrow bandwidth or frequency-dependent oper-ation. Recently, several types of true-time-delay optoelectronic phase-shifters (OEPSs) have been demonstrated by using different techniques like a set of optical fibers with different lengths or mod-ified waveguide structures [12–14], a directly modulated laser-diode [15], and a modified inte-grated optical modulator [16,17]. Some of the OEPSs have already supported broadband (fre-quency-independent) operation and wide phase-tuning range. To facilitate the compatibility of mode-locked EDFLs for the aforementioned applications, we demonstrate an inherently delay-time-tunable picosecond EDFL source by incor-porating a voltage-controlled phase shifter that is modified from a frequency-discriminator. The performance of this delay-time-tunable, mode-locked EDFL including its properties of pulse-width, tuning range, responsivity, and linearity will also be discussed.
The key element of this delay-time-tunable EDFL is a voltage-controlled and frequency-discriminated phase shifter, which is constructed by using a voltage-controlled oscillator (VCO), an integral feedback (loop filter) circuit, a radio-wave frequency synthesizer (RFS), two frequency di-viders (or so-called prescalars), a phase frequency detector (PFD), a temperature-controlled voltage sourceðVREFÞ, and a second-order active loop filter
(LF). Several frequency prescaler ICs (MC120XX series) with divisors of 2, 10, and 256 are used to build up the frequency dividing modules with total devisor of 5120 for the VCO and RFS. The fre-quency-divided signals from the VCO and the RFS are phase-compared each other to generate an er-ror voltage corresponding to the frequency/phase deviation between two signals. In general, there are two different types of PFDs that can be em-ployed: one is the analog multiplier based PFD ICs (such as EXAR XR2208 or Harris ICL8013), and the other is the digital PFD ICs (such as Motorola MC4044 or 12040 series). The analog multiplier ICs although exhibit advantage of high sensitivity, but suffers from more drawbacks such as low phase-detection range (<180°) and input-level dependent gain constant. In contrary, the
maximum phase-detecting range of a digital PFD can be up to 4p, which is particularly suitable for our proposed PLL-PS circuitry due to its linear transfer function within such a large phase-tuning range. In addition, the gain coefficient of the dig-ital PFD is constant. The temperature-controlled voltage source is constructed by use of a temper-ature-stabilized and high precision voltage regu-later (National Semiconductor, LM399) in connection with a resistor-based variable voltage tuning circuit. The second-order active LF is im-plemented by use of operational amplifier (Analog Device, OP27), RC, and discharging circuits. Ex-perimentally, a microwave oscillator (VCO, HP8640B) operated in DC/FM mode is employed as the VCO, which is frequency-discriminated to a frequency synthesizer (RFS, HP8648A) operated at a harmonic frequency of the linear repetition rate. The frequency of the VCO is synchronized to the RFS and the error voltage feedback to the VCO exhibits only a DC component. The timing jitter of the EDFL can thus be suppressed under the frequency discriminating condition. However, the error voltage can be further added with a controlling voltage ðVREFÞ to achieve the
adjust-ment of the relative phase shift as well as the delay-time between VCO and RFS [18,19]. This leads to a voltage-controlled phase difference between the signals from RFS and locked VCO. Moreover, there are two different operating modes designed for the proposed laser. In free-running mode, the reference clock is coming from the RFS as afore-mentioned. However, the reference clock may also come from the signal optoelectronic converted from output of the EDFL for the laser operated at a feedback controlling mode (as shown by a dotted line in Fig. 1). In this case, mode-locking of EDFL can directly be built up by synchronizing the phase of modes involved in the amplified spontaneous emission (ASE) spectrum.
In more details, the basic principle of the fre-quency-discriminated phase shifter can be ex-plained by deducing the phase change of the VCO from the equivalent noise model of the the cir-cuitry schematically shown in Fig. 2. By using the nodal analysis, we can express the effect of the additional controlling voltageðVREFÞ on the phase
he¼ hIF1 hIF2 hIF1¼ URFS N þ U1=N ; hIF2¼ UVCO N þ U1=N þ hVCO
hVCO¼ K½ dheþ NSðUPFDÞ VREFF ðsÞK0=sþ UVCO
ð1Þ
where he is the phase difference between the input
signal and the reference at the phase detector, hIF1
is the phase of the intermediate frequency (IF) reference signal which is frequency-divided from the RF synthesizer through the divider, hIF2 is the
phase of the intermediate frequency (IF) reference signal derived from the VCO through another frequency divider, and hVCO is the phase of the
VCO under closed-loop condition. The single-siband (SSB) phase noise of each component is de-noted by U with corresponding sub-indices. For example, URFS, U1=N, UPFD, and UVCO denote the
corresponding SSB phase noises of the reference frequency synthesizer, frequency divider, phase/ frequency detector, and voltage controlled coscil-lator, respectively. In addition, FðsÞ is the Laplace-transformed transfer function of the second-order active loop filter (LF), the Kd and K0 are the gain
constants of the phase detector and the VCO re-spectively. In Eq. (1), we have assumed that an offset voltage ðVREFÞ is added prior to the LF. By
substituting hIF1 and hIF2 into he, we can rewrite
the phase change of the output signal from the VCO which is phase-locked to the clock signal: hVCO Kd URFUN VCO þ NsðUPDÞ VREF FðsÞK0=s 1þ KdFðsÞK0=s þ UVCO 1þ KdFðsÞK0=s UPLL VREF FðsÞK0=s 1þ KdFðsÞKo=s UPLLþ DhVCO: ð2Þ Here DhVCO is the phase shift of the VCO
caused by adding VREF. Eq. (2) therefore represents
the effect of the offset voltage VREF on the phase
change of the microwave signal from the VCO. Owing to a relatively large loop gain in the pass-band, we can make the following approximation:
FðsÞK0=s
1þ KdFðsÞK0=s
1=Kd: ð3Þ
As a result, the phase change DhVCO of the
mi-crowave signal from the frequency-discriminated VCO is approximately in linear proportion to the offset voltage ðVREFÞ as follows:
Fig. 1. The setup of a delay-time-tunable, mode-locked erbium-doped fiber ring laser with (a) free-running mode (reference clock coming from RFS) and (b) feedback controlling mode (reference clock coming from PD). WDM: wavelength-division multiplexing coupler; EDF: erbium-doped fiber; PC: polariza-tion controller; MZM: Mach–Zehnder intensity modulator; PD: high-speed photodetector; PLL-PS: phase-locked-loop phase shifter; DSO: digital sampling oscillator; Comb: comb generator (optional); BPF: band-pass filter.
Fig. 2. The configuration and equivalent noise model of the frequency-discriminated phase shifter. RFS: radio-wave fre-quency synthesizer; LF: loop filter; PFD: phase frefre-quency de-tector; VCO: voltage controlled oscillator; 1=N : the frequency divider.
DhVCO¼ VREF
FðsÞK0=s
1þ KdFðsÞK0=s
VREF=Kd
RdVREF: ð4Þ
That is, the phase change of the microwave signal as well as the delay-time of the EDFL controlled by the VCO based servo-loop can be linearly changed by tuning the DC voltage ðVREFÞ
without the use of high- frequency circuit compo-nents. The parameter Rd 1=Kd (with a unit of
degree/volt) represents the phase-tuning sensitiv-ity. Choosing a phase comparator with a larger phase-gain constant Kd is mandatory, however,
will also lead to smaller phase sensitivity and a reduced phase-shifting range. Therefore, an opti-mal design of the phase-comparing circuit for achieving larger phase sensitivity without degrad-ing the phase-compardegrad-ing ability is needed for practical applications.
Fig. 2 shows the block diagram of actively mode-locked EDFL operated in either free-run-ning mode or delay-time controlled mode. The EDFL consists of a unidirectional ring cavity, which is constructed by using a EDF with length of 3 m (as the gain medium), a laser diode at 978 nm (as the pumping source), a 1 2 coupler with 90/10 output coupling ratio, three fiber-based po-larization-independent optical isolators (although one should be enough), two polarization control-lers, an optical pass filter with 3 dB band-width of 0.7 nm (wavelength tunable), and a Mach–Zehnder intensity modulator with a 3 dB bandwidth greater than 10 GHz (acts as a active mode-locker). The total length of the EDF ring laser cavity is 16 m, which corresponds to a fun-damental cavity mode spacing of 12.5101 MHz. The mode-locked condition is achieved by tuning the frequency of the RFS ðfhÞ to match a higher
order harmonic component (with an integral multiple N) of the cavity mode (fh¼ 500:404
MHz, N ¼ 40). The VCO signal is further ampli-fied by a microwave power amplifier with a gain of 27 dB, and then sent into a comb generator to output an electrical pulse train for driving the ac-tive mode-locker. This facilitates a more stable mode-locking performance with reduced timing jitter. The optical pulse-train generated from the actively mode-locked EDFL is monitored by using
a high-speed photodetector (f3dB¼ 45 GHz) and a
sampling oscilloscope (f3dB¼ 50 GHz).
The temporal trace of the optical pulse-train generated from the mode-locked EDFL under the control of VREF is simultaneously monitored by
DSO. The leading and delayed optical pulse-trains with respect to the original one are detected and shown in Figs. 5(a) and (b). The upper trace in each figure illustrates the original 500.404 MHz optical pulse-train from the mode-locked fiber ring laser without tuning the VREF of
frequency-dis-criminated phase shifter. As VREF tuned from 0 to
)3.35 V, the pulse train is found to be delayed by 0.9 ns (see in Fig. 3(a)). Alternately, a leading time of 1 ns for the pulse train by setting VREF¼ 3:6 V
can be seen in Fig. 3(b). By changing the VREFfrom
)3.35 to 3.6 V, the laser pulse-train is found to shift from )0.9 to 1 ns with respect to the RFS signal as well as the originally un-shifted optical pulse-train. This tuning range of the delay-time thus corresponds to nearly one period of the laser
Fig. 3. The time-delayed optical pulse-train repeated at 500.404 MHz controlled at different VREF with respect to the original one (upper trace). The lower traces in (a) and (b) show the optical pulse-train delayed by setting VREFas)3.35 and 3.6 V, respectively (x-axis: temporal span in 500 ps/div).
pulse train, i.e., a maximum phase-tuning range close to 2p. Note that the optical pulsewidth of the mode-locked EDF ring laser maintains constant (sFWHM 59 ps with the adding band-pass filter)
during the delay-time tuning procedure. The rela-tive phase shift as well as the delay-time of the optical pulse-train as a function of VREF is shown
in Fig. 4. The maximum phase-shifting range of the laser pulse train is up to 1.9p, which corre-sponds to a maximum temporal duration of about 0.95 period (also corresponding to a delay-time of 1.9 ns at a repetition rate of 500.404 MHz). It is worthy mentioning that neither the pulse repeti-tion rate nor the optical pulsewidth is changed during the delay-time tuning process since the frequency-discriminated phase shifter is working at synchronous condition. The tuning linearity of the frequency-discriminated phase shifter as a function of controlling voltage ðVREFÞ is nearly
sinusoidal with a period of of 3.2 V and the esti-mated maximum deviation ðDh=hÞ is < 5% (see Fig. 5). The tuning responsivity of the EDFL pulse-train is calculated to be 0.27 ps/mV, which corresponds to a tuning resolution of 0.5 ps, lim-ited by the voltage fluctuations of about 2 mV in our current system. It is estimated that a relative delay-time of 10 ps is already enough for estab-lishing an OTDM system with the bit-rate up to 100 Gb/s by using an array of our proposed delay-time-tunable mode-locked EDFLs. On the other
hand, enhancing the tuning resolution of the cur-rent system is still mandatory for high-speed op-tical sampling measurements. In comparison with commercially available analog microwave phase shifters, the demonstrated frequency-discriminated phase shifter has been shown to exhibit better performances on pulse-tuning linearity and maxi-mum shifting range. Nonetheless, further im-provement on the tuning linearity and the shifting range of the system to reach the theoretical limits are straightforward and should be achievable via the use of a PFD with higher operating bandwidth and shorter rising/falling time.
Finally, the single-sided-band (SSB) phase noise spectra of the delay-time-tunable mode-locked EDFL operated at free running (dotted line) and feedback controlling (solid line) modes are mea-sured, respectively. The single-side-band (SSB) phase noise spectrum is measured by using a high-speed photodetector (New Focus 1014, f3dB>45
GHz) in connection with a DC-signal blocker and a microwave spectrum analyzer (Rhode and Sch-wartz FSEK30, f3dB¼ 40 GHz) with resolution
bandwidth of 1 Hz. The SSB phase noise density spectrum is taken from the double-side-band spec-trum associated with its power level of the central peak normalizing to 0 dBm. The Rhode and Sch-wartz offers a standard programmable for extrac-tion of the SSB phase noise density spectrum from the general spectrum. The carrier frequency (for
Fig. 4. The repetition rate (solid square with dashed line) and the relative delay-time (solid line) of the optical pulse-train from mode-locked EDFL plotted as a function of controlling voltage ðVREFÞ.
Fig. 5. The tuning linearity as a function of controlling voltage ðVREFÞ is nearly sinusoidal with period and maximum estimated deviationðDh=hÞ.
example, fcarrier¼ 500:404 MHz) is offset to zero
and positioned at left margin of the figure when transferring the SSB phase noise spectrum. The scanning time is set as 60 s. The resolution band-width is 1 Hz at offset frequency of lower than 200 Hz, which is subsequently increasing up to 5 and 10 Hz as the offset frequency increases up to 1 and 5 kHz, respectively. Note that the system limitation of about)120 dBc/Hz. In Fig. 6, it is observed that the instantaneous SSB phase noise of the EDFL operated at free-running mode is higher than)50 dBc/Hz (measured at offset frequency of less than 50 Hz), which further decreases down to)70 dBc/Hz at offset frequency of 100 Hz or larger. When op-erating at feedback controlling mode, the SSB phase noise of the EDFL is significantly reduced by 10 dBc/Hz or larger at offset frequencies of >100 Hz as compared to that of the laser operated at free-running mode. In general, the SSB phase noise of the optical pulse generated from the feedback con-trolled EDFL can remain as <)105 dBc/Hz at offset frequency of 1 kHz or larger, which is only 10 dB larger than that of the RFS used in our experiment. It is observed that the noise density spectrum at offset frequencies below 100 Hz is greatly fluctuated due to the frequency independence between the RFS and EDFL longitudinal mode at free-running mode. This indicates there are still a residual fre-quency drift (or modulation) of the longitudinal mode of the EDFL cavity with a free-running mode-locker due to thermal and environmental
fluctuations, which significantly changes the noise spectrum in low offset frequency region (foffset<100
Hz) at different scans (with scanning period of 5–60 s under a frequency span of 10 kHz). This inevitably causes the unexpected spectrum dip at lower offset frequency region during scans. Indeed, the excellent tracking performance of the PLL-PS facilitates the noise suppression of the mode-locked EDFL at offset frequencies within the pass-band of the loop filter in the PLL-PS. The further reduction in phase noise density at low offset frequencies (foffset<100
Hz) depends strictly on both the noise figure of the electrical frequency reference (under the frequency-tracking mode) the stability control of the EDFL cavity itself (under the regeneratively mode-locking mode). Nonetheless, the feedback controlling and fast tracking capability of the PLL-PS has effi-ciently reduced the noise densities. These results interpret that the feedback controlled operation would be promising for certain applications re-quiring optical pulses with low phase noises and jitters.
In conclusion, we have demonstrated a jitter-suppressed and delay-time-tunable mode-locked EDFL by using a modified frequency-discrimi-nated phase shifting circuit as the pulse delay-time controller. The optical pulse-train can be syn-chronized and phase-tuned with respect to the external microwave clock. By changing the voltage from)3.35 to 3.6 V, the maximum tuning range of the phase and the delay-time of the 500.404 MHz optical pulse-train from the laser can be up to 340° (1.9p) and 1.9 ns (nearly 1 period) respectively under a tuning responsivity of 0.27 ps/mV. The tuning linearity as a function of the controlling voltage is slightly sinusoidal with maximum esti-mated deviation is < 5%. The SSB phase noise of the laser operated at feedback-controlled mode is significantly reduced to the system limitation of about)120 dBc/Hz (measured at offset frequency of 100 kH), which is at least 7 dB smaller than that of the laser operated at free-running mode. High-speed delay-time keying is possible via program-mable control of the VREF in the servo-loop.
Extension of this technique to a higher repetition rate should be possible and it is expected that such lasers should find many important applications in high bit-rate OTDM and EOS systems.
Fig. 6. The single-sided-band phase noise spectra of the delay-time-tunable mode-locked EDFL operated at free running (dotted line) and feedback controlling (solid line) modes.
Acknowledgements
This work was supported in part by the Na-tional Science Council (NSC) of the Republic of China under Grants NSC89-2215-E-027-007 and NSC90-2215-E-027-008. The authors appreciate Mr. Y.-L. Cheng for his support in taking some of the experimental data.
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