前瞻智慧型天線系統之關鍵元組件研製

行政院國家科學委員會專題研究計畫 成果報告

前瞻智慧型天線系統之關鍵元組件研製

研究成果報告(精簡版)

計 畫 類 別 ： 個別型

計 畫 編 號 ： NSC 95-2221-E-002-085-

執 行 期 間 ： 95 年 08 月 01 日至 96 年 07 月 31 日

執 行 單 位 ： 國立臺灣大學電信工程學研究所

計 畫 主 持 人 ： 江簡富

計畫參與人員： 博士班研究生-兼任助理：張子軒

碩士班研究生-兼任助理：李宜音、陳純熙、鄧平援、李偉

暘、杜博仁、楊善詠、麥肇倫

報 告 附 件 ： 出席國際會議研究心得報告及發表論文

處 理 方 式 ： 本計畫可公開查詢

中 華 民 國 96 年 12 月 11 日

行政院國家科學委員會補助專題研究計畫成果報告

前瞻智慧型天線系統之關鍵元組件研製

計畫類別：■ 個別型計畫 □ 整合型計畫

計畫編號：NSC 952221E002 085

-執行期間：九十五年八月一日至九十六年七月三十一日

計畫主持人：江簡富 教授

共同主持人：

計畫參與人員：張子軒、李宜音、陳純熙、鄧平援、李偉暘、杜博仁、

楊善詠、麥肇倫

成果報告類型(依經費核定清單規定繳交)：□精簡報告 ■完整報告

本成果報告包括以下應繳交之附件：

□赴國外出差或研習心得報告一份

□赴大陸地區出差或研習心得報告一份

■出席國際學術會議心得報告及發表之論文各一份

□國際合作研究計畫國外研究報告書一份

處理方式：除產學合作研究計畫、提升產業技術及人才培育研究計畫、

列管計畫及下列情形者外，得立即公開查詢

■涉及專利或其他智慧財產權，□一年□二年後可公開查詢

執行單位：國立台灣大學電信研究所

中 華 民 國 九 十 六 年 十 月 三 十 日

中文摘要

關鍵詞：介電質共振天線、單極天線、低雜訊放大器、超寬頻、切換電感振盪器、充電時

間、液晶螢幕顯示器、驅動電路、場序式。

本計畫研究成果分為六項子題，分別呈現在六篇期刊論文及六件專利申請中，該六項

子題分別為︰

子題一、寬頻介電質共振-單極天線

子題二、多頻之分離式介電質共振器天線

子題三、低電壓低功率24 GHz CMOS低雜訊放大器

子題四、適用於超寬頻的雙頻切換電感振盪器

子題五、延長充電時間並可精確充電之液晶螢幕驅動電路

子題六、場序式顯示器之設計

英文摘要

Key words: dielectric resonator antenna, monopole antenna, low-noise amplifier, ultra-wideband,

switched-inductor oscillator, charging time, liquid crystal display, driver, field sequential color.

The outcome of this project is divided into six subjects, respectively presented in six journal

papers and six patents. These six subjects are

Subject 1: Broadband dielectric resonator antenna with metal coating

Subject 2: Dualband split dielectric resonator antenna

Subject 3: A K-band CMOS low-noise amplifier with low dc power consumption

Subject 4: Dual-band VCO with switched inductors for UWB applications

Subject 5: Active and adaptive charging method on data lines for delay compensation

Subject 6: Design constraints on FSC LCD

可供推廣之研發成果資料表

■ 可申請專利 ■ 可技術移轉

日期：96 年 10 月 30 日

國科會補助計畫

計畫名稱：前瞻智慧型天線系統之關鍵元組件研製

計畫主持人：

江簡富 教授

計畫編號：NSC 95-2221-E-002 -085 –

學門領域：電信學門

技術/創作名稱

寬頻介電質共振-單極天線

發明人/創作人

張子軒、江簡富

中文：

本發明結合介電質共振天線(DRA)與單極天線(monopole)達到 49%

寬頻效果。不僅體積小、構造簡單、製作容易，且利用共面波導

(CPW)饋入，易與其他平面電路整合。在頻帶內的輻射場型皆為全

方位性。

技術說明

英文：

In this invention, DR antenna is integrated with monopole antenna to

achieve a broadband of 49% bandwidth. This proposed DR-monopole

antenna has many advantages such as small volume, simple structure

and ease of fabrication. The antenna is fed by coplanar waveguide,

which can be easily integrated with other planar circuits. The radiation

pattern is omnidirectional over the bandwidth.

可利用之產業

及

可開發之產品

WLAN 802.11a 網路相關產品。

技術特點

新穎性：本發明結合介電質共振天線與單極天線，具有良好的線性

極化、場型為全方向性、體積小且可達到 49%的頻寬。結構簡單容

易實作、利用共面波導結構饋入，易於和其他平面元件整合。

進步性：在 WLAN 802.11a 的應用中，接入點(access point)和個人

電腦收發機(transceiver)的相對位置會隨著使用者移動而改變，為了

使用上的方便，天線場型必須為全方向性。本發明不僅天線體積

小，製作簡單、成本低廉，具有全方向性場型的優點，符合 WLAN

802.11a 無線網路應用。

推廣及運用的價值

產業上利用性：電腦無線網路日益普及，傳輸速度快，廣為大眾所

接受。本發明將單極天線與介電質共振天線的頻帶相連結，並設計

共面波導饋入系統整合兩天線以達到 49%的寬頻。

可供推廣之研發成果資料表

■ 可申請專利 ■ 可技術移轉

日期：96 年 10 月 30 日

國科會補助計畫

計畫名稱：前瞻智慧型天線系統之關鍵元組件研製

計畫主持人：

江簡富 教授

計畫編號：NSC 95-2221-E-002 -085 –

學門領域：電信學門

技術/創作名稱

多頻之分離式介電質共振器天線

發明人/創作人

張子軒、江簡富

中文：本發明在一矩形介電質共振器中嵌入一狹縫與四個凹口，獲

得雙頻的特性，並增加頻寬。該狹縫將本體從中分成兩個相同之部

份，每一部份各嵌入兩個凹口，利用微帶線當作訊號線，透過耦合

槽縫饋入天線。本發明易與其他平面電路整合，並減少其他元件對

天線的干擾。

技術說明

英文：In this invention, the resonant frequencies of a DR antenna is

tuned by embedded a split and four notches, and the bandwidth is

increase. The split separates the dielectric resonator into two identical

parts, with two notches in each part. A microstrip line is used to feed

this antenna via a coupling aperture. This invention is easy to integrate

with other planar circuits, and can minimize interference from other

components.

可利用之產業

及

可開發之產品

WiMAX、WLAN 802.11a 網路相關產品。

技術特點

新穎性：本發明係在介質共振器天線中嵌入凹口與狹縫，調整

、

、

之共振頻率，以移動頻帶至所需的應用。在狹

縫與凹口處的電場會被增強，使能量能更有效率地輻射，降低天線

品質因子，增加頻寬。利用微帶線透過槽縫耦合至天線，易於和其

他平面元件整合。

y 111

TE

y 112

TE

y 113

TE

進步性：本發明具有寬波束的垂直極化輻射場型，可用於 WiMAX

與 WLAN 無線網路應用。

推廣及運用的價值

產業上利用性：無線網路日益普及，傳輸速度快，廣為大眾所接受。

本發明係一多頻介電質共振器天線，該天線的頻寬涵蓋 3.375-3.93

GHz 與 5.08-5.415 GHz，滿足 WiMAX 與 WLAN 之規格，在水平

面上，具由寬波束、垂直極化的輻射場型。

可供推廣之研發成果資料表

■ 可申請專利 ■ 可技術移轉

日期：96 年 10 月 30 日

國科會補助計畫

計畫名稱：前瞻智慧型天線系統之關鍵元組件研製

計畫主持人：

江簡富 教授

計畫編號：NSC 95-2221-E-002 -085 –

學門領域：電信學門

技術/創作名稱

低電壓低功率 24 GHz CMOS 低雜訊放大器

發明人/創作人

鄧平援、江簡富

中文：本發明利用 TSMC 0.18 µm CMOS 製程設計一個應用於 24

GHz 短距離雷達感測器系統的低雜訊放大器。主要架構為三級的

共源放大器串接，並包含前端輸入級匹配網路、中間級匹配電路以

及輸出匹配電路。本發明的功率消耗僅 8.3 mW (1 V supply)，power

gain 為 13.5 dB，noise figure 為 4.7 dB，input/output return loss > 10

dB，晶片面積為 0.64 mm x 0.48 mm。

技術說明

英文：This invention presents a 24 GHz LNA for short-range radar

system using TSMC 0.18 um CMOS process. The architecture is a

cascade of three stages of common-source amplifiers, accompanied by

input, inter-stage, and output matching networks. This invention

consumes 8.3 mW under 1 V supply, its power gain is 13.5 dB, noise

figure is 4.7 dB, return loss is less then – 10 dB, the chip size is 0.64

mm x 0.48 mm.

可利用之產業

及

可開發之產品

毫米波低雜訊放大器、短距離雷達感測系統。

技術特點

本發明的特點為：用適當的剪裁技巧來選取電晶體尺寸，使用較小

的尺寸來達到阻抗匹配，因此降低功率消耗。本發明的被動元件(如

電感)均為 on-chip 的繞線平面式電感，電容為 TSMC 所提供的 MIM

電容。

推廣及運用的價值

產業上利用性：毫米波雷達系統將是汽車防撞系統的重要技術，如

何接收並放大訊號，且不額外增加雜訊是一個重要的研究主題。

一般熟知的射頻電路多需要特殊製程，如 GaAs HEMT、SiGe

HBT，不僅價格昂貴，且無法與數位電路結合，製作 SOC 並不實

際。本發明採用低價格的 CMOS 製程，符合產業發展趨勢。

可供推廣之研發成果資料表

■ 可申請專利 ■ 可技術移轉

日期：96 年 10 月 30 日

國科會補助計畫

計畫名稱：前瞻智慧型天線系統之關鍵元組件研製

計畫主持人：

江簡富 教授

計畫編號：NSC 95-2221-E-002 -085 –

學門領域：電信學門

技術/創作名稱

適用於超寬頻的雙頻切換電感振盪器

發明人/創作人

李偉暘、江簡富

中文：本發明使用切換電感當開關切換兩個不同的頻帶，與一般用

切換電容當開關不同。不僅構造簡單、相位雜訊低、消耗功率與現

有設計互有高低，且能切換兩個頻帶來取代使用兩個 VCO 架構，

降低了晶片的面積與成本。

技術說明

英文：In this invention, a VCO is implemented with switched inductors

to achieve dual-band operation. Compared with VCO using switched

capacitors, this proposed VCO has advantages such as simple structure,

low power consumption, low phase noise, and is capable of switching

two frequency bands to replace two separate VCOs. Hence, its chip

size is small and the cost is reduced.

可利用之產業

及

可開發之產品

UWB 無線網路相關產品。

技術特點

新穎性：本發明使用切換電感開關切換兩個不同的頻帶，與一般用

切換電容當開關不同，若以電容開關要達到能切換較寬的兩個頻

帶，必須加裝較多的切換電容開關，而開關越多所產生的雜訊就越

多，而導致相位雜訊過高。

進步性：在 UWB 的應用中，為了使用 OFDM 的傳輸方式，需要

產生不同的 LO 訊號，而本發明的 VCO 相位雜訊較低，且能產生

雙頻的 LO 訊號，符合 UWB 無線網路應用。

推廣及運用的價值

產業上利用性：本發明利用切換電感來切換兩個頻帶，取代使用兩

個 VCO 架構，降低了晶片的面積與生產成本。

可供推廣之研發成果資料表

■ 可申請專利 ■ 可技術移轉

日期：96 年 10 月 30 日

國科會補助計畫

計畫名稱：前瞻智慧型天線系統之關鍵元組件研製

計畫主持人：

江簡富 教授

計畫編號：NSC 95-2221-E-002 -085 –

學門領域：電信學門

技術/創作名稱

延長充電時間並可精確充電之液晶螢幕驅動電路

發明人/創作人

陳純熙、江簡富

中文：大尺寸及高解析度的液晶顯示器是顯示器的發展趨勢，其最

需解決的是充電時間不足的問題。本發明利用兩排畫素之間的差

值、資料線上的時間常數計算出快速充電所需的電壓。此外，因本

發明將三條資料線同時開啟，可減少兩條資料線的延遲時間，大幅

提高資料驅動電路可用的充電時間。與傳統的資料驅動電路相比能

夠達到快速充電的目的。另外，本發明亦較其他預充電的方法精確。

技術說明

英文：A fast charging method for large-size or high-resolution liquid

crystal display is proposed by comparing data of adjacent rows. The

proposed method bundles three rows in one set, and the charging

period allocated for these three rows are rearranged to charge all three

rows more precisely than conventional methods.

可利用之產業

及

可開發之產品

液晶顯示器驅動電路相關產品。

技術特點

新穎性：以相鄰畫素的資料以及資料線上的時間常數產生微幅調整

所需的電壓，以達到減少充電時間的目的。

進步性：本發明利用重新分配充電時間的方法，在三條掃瞄線為一

組之內重新分配各掃瞄線所需的時間延遲，可以大幅提高充電的準

確性。

推廣及運用的價值

產業上利用性：適用於大尺寸的液晶螢幕之資料及掃瞄驅動電路。

可供推廣之研發成果資料表

■ 可申請專利 ■ 可技術移轉

日期：96 年 10 月 30 日

國科會補助計畫

計畫名稱：前瞻智慧型天線系統之關鍵元組件研製

計畫主持人：

江簡富 教授

計畫編號：NSC 95-2221-E-002 -085 –

學門領域：電信學門

技術/創作名稱

場序式顯示器之設計

發明人/創作人

李宜音、江簡富

中文：本發明提出場序式顯示器之晝素電路設計法則，滿足充電、

電位保持、電容耦合及信號延遲四個限制。利用此方法，可將場序

式面板晝素耗電控制為濾光片式面板晝素耗電之 1/6，場序式面板

儲存電容所需面積僅為濾光片式面板儲存電容之 30%，場序式面板

之開孔率為濾光片式之 1.7-2 倍。

技術說明

英文：This invention proposes a methodology to design pixels for field

sequential color LCDs, satisfying four constraints on charging, holding,

asymmetric kickback and delay. Compared with color filter LCDs, the

power consumption can be reduced to 1/6, the storage capacitor can be

reduced to 30 %, the aperture ratio can be increased to 1.7-2 folds.

可利用之產業

及

可開發之產品

液晶顯示器相關產品。

技術特點

本發明以簡易之方法進行場序式顯示器之設計，較傳統的設計能節

省約 67%的功率，適合應用在省電式低功率液晶面板。

本發明提出場序式顯示器在充電、電位保持、電容耦合及信號延遲

四個限制。

推廣及運用的價值

產業上利用性：本發明可利用同樣解析度同樣尺寸濾光片式顯示器

之設計參數，快速換算出場序式顯示器之設計參數。

1254 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 55, NO. 5, MAY 2007

Broadband Dielectric Resonator Antenna With Metal

Coating

Tze-Hsuan Chang, Student Member, IEEE, and Jean-Fu Kiang, Member, IEEE

Abstract—A broadband dielectric resonator (DR) antenna is

proposed, which consists of a rectangular DR coated with metal on three sides and placed on a ground plane. The structure is ana-lyzed by modelling the dielectric-air interface as perfect magnetic conductor (PMC). A coplanar waveguide (CPW) with terminating slots is used to feed the antenna. Measurement results exhibit a wide bandwidth of about 47% over which the pattern on the horizontal plane is nearly omnidirectional. The 10-dB bandwidth of this broadband DR monopole covers 4.2–6.8 GHz. Hence, it can be used for WLAN 802.11a applications.

Index Terms—Dielectric resonator (DR) antenna, monopole

an-tenna.

I. INTRODUCTION

T

HE prevalence of wireless communication demands broadband antennas which can be embedded within a handset to provide versatile applications. Since it is difficult to obtain wide impedance bandwidth with single resonant antenna, multiple antennas with different operating frequencies have been integrated to satisfy the bandwidth requirement [1], [2].

High-permittivity dielectric material has been used in mi-crowave circuits such as filters or oscillators [3]. In order to fa-cilitate the design of dielectric resonator, a heuristic approach that models the dielectric-air interface as a perfect magnetic wall was proposed to predict the resonant frequencies of cylindrical resonators in 1965 [4]. Since 1983, dielectric resonators have been designed as antenna elements by exciting different modes of DR using conventional feeding mechanisms [5].

A DR antenna exhibits a broader bandwidth if its factor is lower. In [6], a notched rectangular DR antenna with a low factor is proposed. By lifting a DR above the ground plane, its

factor can be effectively reduced [7]. The bandwidth of DR can also be increased by modifying its geometry. For example, a truncated tetrahedral DR with its narrow base attached to the ground reaches an impedance bandwidth of 40% [8]. A split conical DR with split side attached to the ground can reach a bandwidth of 50% [9].

High-permittivity material can be used to reduce the size of DR at the expense of bandwidth reduction. However, DRs with Manuscript received October 24, 2006. This work was sponsored by the Na-tional Science Council, Taiwan, ROC, under contract NSC 93-2213-E-002-034. The authors are with the Department of Electrical Engineering and the Grad-uate Institute of Communication Engineering, National Taiwan University, Taipei, Taiwan (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TAP.2007.895582

larger aspect ratio has been used to reduce the factor and hence obtain a wider impedance bandwidth [10], [11].

Multiple DRs with close resonant frequencies can exhibit broadband characteristics by coupling their resonant modes. For example, two cylindrical DRs can be stacked to couple their modes [1]. In [2], two rectangular DRs with different sizes are placed at proximity, leaving a slot to couple their modes.

The bandwidth of DR antenna can also be extended by at-taching additional parasitic elements to incur another resonance. In [12], two metal strips are attached to the top of a DR to incur additional resonance close to that of the DR. The inductance of the metal strip and the capacitance between the strip and the ground plane form an LC tank which can be coupled to the DR resonant mode to exhibit a wider bandwidth.

The impedance bandwidth of DR antennas can be further in-creased by modifying their feeding structures. In [13], a cou-pling slot is proposed to excite the DR. The resonant modes of slot and DR are coupled to increase the antenna bandwidth.

In [14], a patch resonant mode and a dielectric resonant mode are coupled to increase the DR antenna bandwidth. The signal is fed from the microstrip feed line, through the slot on the ground plane and the slot on the patch, to the DR. In [15], a DR is attached to a circular slot and an eccentric ring slot to achieve a broad bandwidth. A grounded metal plate placed in a plane of symmetry of the electric field distribution can reduce the DR size by half without perturbing the original field distribution. In [16], a rectangular DR integrated with an inverted L-plate antenna is proposed. The DR not only serves as a radiator but also serves as a feeding element for the L-plate.

Typical bandwidth of a rectangular DR antenna is about 6–10%, which can be increased to more than 10% by using lower-permittivity dielectric at the cost of increasing the DR size. Stacking DRs of different sizes or using parasitic DRs can further increase the impedance bandwidth to more than 20% [1], [2]. The former incurs a higher antenna profile, while the latter occupies larger space. Stacking DRs of different permittivities can achieve well coupling to microstrip line and a wider bandwidth of 40% simultaneously. However, the antenna complexity increases. Conical or truncated conical DR can provide more than 50% of impedance bandwidth, but the radiation pattern varies over the band due to the presence of higher-order modes [9].

In this work, a broadband dielectric resonator antenna with a nearly-omnidirectional radiation pattern is proposed. The di-electric resonator is partially coated with metal on its surface, which can be modeled as a cavity having perfect electric con-ductor (PEC) and perfect magnetic concon-ductor (PMC) walls on 0018-926X/$25.00 © 2007 IEEE

CHANG AND KIANG: BROADBAND DR ANTENNA WITH METAL COATING 1255

Fig. 1. DR with metal coating on the bottom and on the other: (a) five sides and (b) three sides, gap of heighth is open near the bottom, metal coating is marked by gray shade.

Fig. 2. Current distribution on metal coating (gray surface), and electric field distribution on DR surface (white surface), corresponding to the DR in Figs. 1(a) and (b), respectively.

different portions of the surface. The resonant modes of the DR are investigated. The electric field and the current distributions of these modes are carefully studied to understand their rela-tion to the radiarela-tion patterns. The metal coating is also fed as a monopole. The input impedance can be matched by adjusting the DR position and the slot length, and broad bandwidth is achieved by coupling the resonant modes of the metal-coated DR and the monopole.

II. RESONANTMODES OFCOATEDDR

Fig. 1(a) shows a rectangular dielectric resonator partially coated with metal, and a small gap of height is open near the bottom of the dielectric resonator. Since the permittivity of the dielectric is much higher than that of the air, the dielectric-air in-terface can be approximated as a PMC boundary, and the metal coating is treated as a PEC boundary. Hence, the structure is a cavity with PEC and PMC on different portions of the surface, filled with high-permittivity dielectric.

The current and the electric field distributions of the funda-mental mode are plotted in Fig. 2(a). The fields and the currents concentrate near the bottom of the dielectric. The electric field across the gap is mainly parallel to the PMC surface. The cur-rent flows vertically from the bottom, changes direction on the metal coating, and ends on the opposite side.

The effects of varying parameters are summarized in Table I. It is observed that the resonant frequency is significantly af-fected by the DR dimensions and , and is less affected by the metal height , since the fields concentrate near the bottom. The effect of decreasing the DR height while keeping constant

TABLE I

EFFECT OFSTRUCTUREDIMENSIONS ON THERESONANTFREQUENCY(f )

has also been considered. The height has significant effect on the resonant frequency only when is comparable to .

Fig. 2(a) shows that the current of the fundamental mode on the back coating has strong horizontal component, which generates electric field with horizontal polarization on the -plane. Hence, the back coating is removed to reduce the horizontal current, as shown in Fig. 1(b). Fig. 2(b) shows the current and the electric field distributions on the metal coating and the DR surface, respectively. The current distribution is similar to that in Fig. 2(a). The electric field starts from the bottom vertically, gradually decreases and bends to terminate at the metal coating. The effects of varying parameters are also summarized in Table I. Compared to that in Fig. 2(a), the width

now has little effect on the resonant frequency. III. MONOPOLEMODE OFMETALCOATING

Place the DR with metal coating as shown in Fig. 1(b) on a ground plane as shown in Fig. 4(a). The metal coating is connected to coplanar waveguide (CPW) signal line to form a monopole antenna, and its resonant frequency is increased as its height is decreased. Fig. 3 shows that the current flows mainly vertically, having a maximum near the ground plane. The current gradually decreases and vanishes at the top. The electric field starts from the ground plane, flows vertically inside the DR, bends and terminates at the coating. Both the current and the electric field have dominant vertical component, which generates strong vertical polarization on the -plane. Since the coating width is comparable to its height, the current has a small amount of horizontal component which generates the horizontal polarization on the plane.

IV. ANTENNAPROPERTIES

Fig. 4(a) shows the configuration of the proposed DR an-tenna. The DR with three-side metal coating shown in Fig. 1(b)

1256 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 55, NO. 5, MAY 2007

Fig. 3. Current and electric field distributions of the monopole mode on metal coating (gray surface) and DR surface (white surface), respectively.

Fig. 4. (a) Configuration of DR-monopole antenna with feeding structure, (b) layout of feeding structure, and (c) photograph.

is placed on the ground plane, the metal coating is connected to the signal line of the CPW to excite the monopole mode, and a pair of open-circuited slots at the end of the CPW are used to excite the DR mode.

Fig. 4(b) shows the layout of the feeding structure, and Fig. 4(c) shows the photograph of the DR-monopole antenna. The DR is placed over the terminating slots of the CPW, and the length of the terminating slots is . The size of ground plane is , and the thickness of the substrate is . The width and the gap of the CPW are adjusted to obtain the characteristic impedance of 50 .

By tuning the monopole height and the dimensions of DR, the resonant frequencies of the monopole and the fundamental mode of DR can be moved close to each other. By changing the lengths of the terminating slots and the offset between the DR and the terminating slots , good impedance matching can be

Fig. 5. Return loss of DR-monopole,a = 3:3 mm, b = 5:6 mm, h = 12 mm, h = 0:5 mm, g = 0:5 mm, w = 10 mm, s = 0 mm, t = 0:6 mm, l = 5:25 mm, W = L = 70 mm, : measurement, : simulation.

achieved. The broad impedance bandwidth is achieved by cou-pling the resonant bands of the monopole and the fundamental mode of DR with metal coating. Fig. 5 shows the return loss of the DR-monopole antenna, the measurement and the simula-tion results match reasonably well at the band edges. The 10-dB bandwidth is about 47.3% (4.2–6.8 GHz), which is wide enough to cover the IEEE 802.11a applications. Two nulls occurs at 4.56 GHz and 6.32 GHz, which are close to the resonant frequencies of the monopole and the fundamental mode of DR, respectively. Fig. 6 shows the radiation patterns generated by the DR-monopole with coating on three sides and five sides, respectively. For frequency associated with the monopole mode, the DR-monopole with three-side coating has a more omnidirectional pattern on the -plane than that with five-side coating. The component with three-side coating is lower than that with five-side coating at . For frequency associated with the fundamental mode of DR, the DR-monopole with three-side coating also has a more omnidi-rectional pattern on the -plane that with five-side coating. Hence, the DR with three-side coating is preferred.

Figs. 7 and 8 show the measurement and simulation radiation

patterns at and , respectively.

At , the pattern on the -plane is nearly omnidirectional, with the gain of about 1.2 dBi. The horizontal current exists on the side metal coating as shown in Fig. 3 and incurs component with multiple lobes. Hence, the com-ponent is only a few dB lower than the component in some directions. The pattern on the -plane is symmetric, and the maximum gain of 3.2 dBi occurs at . The pattern on the -plane is asymmetric, and the maximum gain of 5.2 dBi occurs at .

At , the horizontal current on the side coating as shown in Fig. 2(b) incurs components on the -plane with multiple lobes. The component on the -plane is

CHANG AND KIANG: BROADBAND DR ANTENNA WITH METAL COATING 1257

Fig. 6. Comparison of radiation patterns onxy-plane between DR with coating on three sides and five sides: (a) monople mode and (b) fundamental mode of DR, :E of DR with coating on three sides, :E of DR with coating on five sides, 1 : E of DR with coating on three sides, 2 : E of DR with coating on five sides, 10-dB per division on radials.

nearly omnidirectional with gain of 1.9 dBi. The pattern on the -plane is symmetric and has a maximum gain of 3.0 dBi at , while that on the -plane is asymmetric with the maximum gain of 5.7 dBi at .

The maximum of component on the - and the -planes is tilted from due to the finite size of ground plane. Hence, the antenna gain of vertical polarization is less than 2 dBi on the -plane. For WLAN applications, for example, this DR antenna can be placed on a desk with the -axis pointing to zenith, providing a nearly omnidirectional radiation pattern with vertical polarization on the horizontal plane ( -plane).

Fig. 9(a) shows the effect of antenna height on the reso-nant frequency. When , the resonant modes of the monopole and the DR are strongly coupled. As is increased to 11 mm, the coupled band is split into two resonant bands. When

Fig. 7. Radiation patterns atf = 4:56 GHz, (a) xy-plane, (b) yz-plane, (c) xz-plane, : measuredE , : measuredE , 1: simulated E , 1 2 1: simulatedE , 10-dB per division on radials, all parameters are the same as in Fig. 5.

is further increased, the first resonant frequency gradually de-creases, and a third null appears between the two nulls

associ-1258 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 55, NO. 5, MAY 2007

Fig. 8. Radiation patterns atf = 6:32 GHz, (a) xy-plane, (b) yz-plane, (c) xz-plane, : measuredE , : measuredE , 1: simulated E , 1 2 1: simulatedE , 10-dB per division on radials, all parameters are the same as in Fig. 5.

ated with the monopole mode and the DR mode, respectively. Fig. 10 shows the current distribution and the electric field

dis-Fig. 9. Effects of antenna heighth and dielectric constant on resonant fre-quency,a = 3 mm, b = 6 mm, h = 1 mm, l = 5:35 mm, s = 0:2 mm, g = 0:5 mm, w = 10 mm, W = L = 70 mm, t = 0:6 mm, (a)  = 20, :h = 10 mm, :h = 11 mm, 1 : h = 12 mm,  : h = 13 mm, (b)h = 12 mm, : = 16, : = 18, 1 :  = 20,  :  = 22.

tribution at , associated with the curve shown in Fig. 9(a) with . The current distribution is similar to that of the DR mode near the bottom and similar to that of the monopole mode around the upper portion.

Fig. 9(b) shows the effect of the DR permittivity on the reso-nant frequency. As the dielectric constant is increased, the wave-length in the cavity is reduced, rendering a lower resonant fre-quency. Note that the first resonant frequency is hardly affected by the dielectric constant.

V. CONCLUSION

In this paper, a broadband CPW-fed DR-monopole is pro-posed. The resonant bands of monopole and dielectric resonator are coupled to render a wide bandwidth of 47%. The bandwidth can be adjusted by tuning the resonant frequencies of the DR

CHANG AND KIANG: BROADBAND DR ANTENNA WITH METAL COATING 1259

Fig. 10. Current distribution on metal coating (gray surface), and electric field distribution on DR surface (white surface) atf = 5:5 GHz, parameters are the same as in Fig. 9(a) withh = 13 mm.

and the monopole separately. The component is nearly om-nidirectional on the horizontal plane over the band. The size of

the DR-monopole is , and the

band-width is wide enough to cover the IEEE 802.11a applications. REFERENCES

[1] A. A. Kishk, B. Ahn, and D. Kajfez, “Broadband stacked dielectric res-onator antenna,” Electron. Lett., vol. 25, no. 18, pp. 1232–1233, Aug. 1989.

[2] Z. Fan and Y. M. M. Antar, “Slot-coupled DR antenna for dual-fre-quency operation,” IEEE Trans. Antennas Propag., vol. 45, no. 2, pp. 306–308, Feb. 1997.

[3] R. D. Richtmyer, “Dielectric resonators,” J. Appl. Phys., vol. 10, pp. 391–398, Jun. 1939.

[4] Y. Y. Hung, “Natural resonant frequencies of microwave dielectric res-onators,” IEEE Trans. Microw. Theory Tech., vol. 13, pp. 256–256, Mar. 1965.

[5] S. A. Long, M. W. McAllister, and L. C. Shen, “The resonant cylin-drical dielectric cavity antenna,” IEEE Trans. Antennas Propag., vol. 31, no. 3, pp. 406–412, May 1983.

[6] A. Ittipiboon, A. Petosa, D. Roscoe, and M. Cuhaci, “An investigation of a novel broadband dielectric resonator antenna,” in Proc. IEEE APS Int. Symp., Jul. 1996, vol. 3, pp. 2038–2041.

[7] Y.-D. Kim, M.-S. Kim, and H.-M. Lee, “Internal rectangular dielec-tric resonator antenna with broadband characteristic for IMT-2000 handset,” in Proc. IEEE APS Int. Symp., Jun. 2002, vol. 3, pp. 22–25. [8] A. A. Kishk, “Wide-band truncated tetrahedron dielectric resonator

an-tenna excited by a coaxial probe,” IEEE Trans. Anan-tennas Propag., vol. 51, no. 10, pp. 2913–2917, Oct. 2003.

[9] A. A. Kishk, Y. Yan, and A. W. Glisson, “Conical dielectric resonator antennas for wide-band applications,” IEEE Trans. Antennas Propag., vol. 50, no. 5, pp. 469–474, Apr. 2002.

[10] R. K. Mongia, A. Ittibipoon, and M. Cuhaci, “Low profile dielectric resonator antennas using a very high permittivity material,” Electron. Lett., vol. 30, no. 17, pp. 1362–1363, Aug. 1994.

[11] K. W. Leung, K. M. Chow, and K. M. Luk, “Low-profile high-per-mittivity dielectric resonator antenna excited by a disk-loaded coaxial aperture,” IEEE Antennas Wireless Propag. Lett., vol. 2, pp. 212–214, 2003.

[12] F. R. Hsiao, C. Wang, K. L. Wong, and T. W. Chiou, “Broadband very-high-permittivity dielectric resonator antenna for WLAN application,” in Proc. IEEE APS Int. Symp., Jun. 2002, vol. 4, pp. 490–493. [13] A. Buerkle, K. Sarabandi, and H. Mosallaei, “Compact slot and

dielec-tric resonator antenna with dual-resonance, broadband characteristics,” IEEE Trans. Antennas Propag., vol. 53, no. 3, pp. 1020–1027, Mar. 2005.

[14] K. P. Esselle and T. S. Bird, “A hybrid-resonator antenna: Experi-mental results,” IEEE Trans. Antennas Propag., vol. 53, no. 2, pp. 870–871, Feb. 2005.

[15] T. A. Denidni and Q. Rao, “Hybrid dielectric resonator antennas with radiating slot for dual-frequency operation,” IEEE Antennas Wireless Propag. Lett., vol. 3, pp. 321–323, 2004.

[16] K. Lan, S. K. Chaudhuri, and S. S. Naeini, “Design and analysis of a combination antenna with rectangular dielectric resonator and inverted L-plate,” IEEE Trans. Antennas Propag., vol. 53, no. 1, pp. 495–501, Jan. 2005.

Tze-Hsuan Chang (S’00) was born in Hsin-Chu, Taiwan, R.O.C., on February 1, 1978. He received the B.S. degree in electrical engineering from the National Chung Hsing University, Taiwan, R.O.C., in July 2000. Currently, he is working towards the Ph.D. degree in the Graduate Institute of Commu-nication Engineering, National Taiwan University, Taipei, Taiwan, R.O.C.

Jean-Fu Kiang (M’89) was born in Taipei, Taiwan, R.O.C., on February 2, 1957. He received the B.S. and M.S. degrees from National Taiwan University, Taiwan, R.O.C., and the Ph.D. degree from the Massachusetts Institute of Technology, Cambridge, in 1979, 1981, and 1989, respectively, all in electrical engineering.

From 1985 to 1986, he was with Schlum-berger-Doll Research, Ridgefield, CT; from 1989 to 1990, IBM Watson Research Center, Yorktown Heights, NY; from 1990 to 1992, Bellcore, Red Bank, NJ; from 1992 to 1994, Siemens Electromedical Systems, Danvers, MA; and from 1994 to 1999, National Chung-Hsing University, Taichung, Taiwan, R.O.C. Since 1999, he has been a Professor in the Department of Electrical Engineering and the Graduate Institute of Communication Engineering, National Taiwan University. His research interests include the applications and system issues on electromagnetics, wireless communications, antennas, electromagnetic compatibility, microwave components, etc.

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 55, NO. 11, NOVEMBER 2007 3155

Dualband Split Dielectric Resonator Antenna

Tze-Hsuan Chang, Student Member, IEEE, and Jean-Fu Kiang, Member, IEEE

Abstract—A dualband dielectric resonator antenna (DRA) is

designed by splitting a rectilinear dielectric resonator (DR) and carving notches off the DR. It is observed that notches engraved at different positions affect different modes. Removal of dielectric material from where the electric field is strong incurs a significant increase in resonant frequency. The abrupt change of normal electric field across the discontinuities reduces the -factor and increases the impedance bandwidth. Both theTE₁₁₁andTE₁₁₃ modes incur broadside radiation patterns on the -plane. The proposed DRA can cover both the worldwide interoperability for microwave access (WiMAX, 3.4–3.7-GHz) and the wireless local area network (WLAN, 5.15–5.35-GHz) bands.

Index Terms—Dielectric resonator (DR).

I. INTRODUCTION

D

IELECTRIC resonators made of low-loss and high-per-mittivity material have been used to implement antennas [1]. They have higher radiation efficiency than printed antennas at higher frequency due to the absence of ohmic loss and surface wave, in addition to compact size, light weight, and low cost.

Many efforts have been devoted to developing multiband or wideband dielectric resonator antennas (DRAs) [2]–[15]. For example, make the feeding aperture radiate like a slot antenna to incur another band [2]–[4] and induce parasitic effects with attached metal strips [5]–[7].

In [8], specific higher order modes with the electric field dis-tribution on the top surface of the dielectric resonator (DR) sim-ilar to that of the fundamental mode are intentionally excited. In [9] and [10], higher order modes of truncated conical or tetra-hedral DR are excited to obtain wide impedance bandwidth.

DRs of different sizes have been placed vertically to form a stacked DRA, or at close proximity, to form a multielement DRA to attain wideband or dualband features [12]–[15].

In this paper, a dualband DR antenna is proposed by split-ting a rectilinear DR evenly. The electric field over the gap in between is significantly enhanced, hence reducing the -factor. Two notches are also engraved in each piece to tune the reso-nant frequencies and increase the impedance bandwidth as well. The effect of the gap and notches on the resonant frequencies are carefully studied and the resonant bands associated with the and modes can be adjusted to cover the worldwide interoperability for microwave access (WiMAX, 3.4–3.7-GHz) and the wireless local area network (WLAN, 5.15–5.35-GHz) bands.

Manuscript received January 20, 2007; revised May 1, 2007. This work was supported by the National Science Council, Taiwan, under Contract NSC 93-2213-E-002-034.

The authors are with the Department of Electrical Engineering and the Grad-uate Institute of Communication Engineering, National Taiwan University, Taipei 106, Taiwan (e-mail: [email protected]).

Digital Object Identifier 10.1109/TAP.2007.908830

Fig. 1. Configuration of split DRA. (a) Panoramic view. (b) Top view. (c) Photograph.

II. ANTENNACONFIGURATION

Fig. 1 shows the configuration of the DRA, which is com-posed of two identical rectangular DRs of dimension , separated by a gap . Each DR is engraved with two notches at its bottom and side edge, with dimensions and , respectively. The DRs are placed on a ground plane of size on an FR4 substrate of thickness and permit-tivity 4.4. A microstrip line is used to feed the DRs through an aperture of size . The microstrip line is extended over the aperture by . The offset between the aperture and the DR is .

The resonant frequency is mainly determined by the DR dimensions and permittivity . The carved notches change the electric field distribution in the original DRs, hence the resonant frequencies. Since the gap is perpendicular to the electric field of the mode of the otherwise intact DR, the electric field is enhanced within the gap. Thus, the resonant frequency of the mode and the input impedance are significantly affected. The input impedance can be fine tuned by adjusting the DR offset , the length of the extended microstrip line, and the aperture length .

3156 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 55, NO. 11, NOVEMBER 2007

Fig. 2. Single block of dielectric resonator.

III. PREDICTION OFRESONANTFREQUENCYSHIFT The electric field and the magnetic field in a dielectric resonator taking the space satisfy the Maxwell’s equations

(1) (2) where is the resonant frequency. When the shape of dielec-tric resonator is modified by engraving gap, tunnel, or notch, the dielectric constant in the space becomes a function of location and the field distributions and the resonant frequency be-come and , respectively, satisfying the Maxwell’s equa-tions as well. Applying the reaction operation between the orig-inal field and the perturbed field [16], the resonant frequency of the modified DR can be expressed as

(3) where

which indicates that the resonant frequency is affected by the reaction between the field distributions of the original and the modified DR structures. It also implies that the resonant fre-quency can be more accurately predicted if the perturbed field can be approximated with reasonable accuracy. For example, if a small gap is carved off a DR, the electric field normal to the air-dielectric interface will be significantly enhanced, which can be observed by simulation.

IV. RECTANGULARDIELECTRIC RESONATORWITH SHAPEMODIFICATIONS

A DR of dimension on an infinite ground plane can be viewed as a single block of rectangular dielectric with height in free space, as shown in Fig. 2. Since the permittivity of DR is much higher than that of the air, the air-dielectric interface can

be approximated as a perfect magnetic conductor (PMC) wall in a first-order analysis [17], and the modes can be categorized into transverse electric (TE) and transverse magnetic (TM) modes [18]. It is shown that the PMC approximation gives more accu-rate results with the TM modes than with the TE modes [17]. The dielectric waveguide model (DWM) is proposed to render more accurate prediction, in which the DR is treated as a por-tion of a dielectric waveguide truncated in the propagapor-tion di-rection [19]–[21]. The PMC approximation is imposed on the guide surfaces and total reflection is assumed in the propaga-tion direcpropaga-tion. By this way, the fields of the modes with odd can be derived as

(4)

where is an arbitrary constant, , and

is determined from [22]

(5) The resonant frequency can thus be calculated as

(6) The field expressions of the modes with even can be derived as

(7)

where is an arbitrary constant, ,

and the resonant frequency can be determined from (5) and (6), respectively.

Fig. 3 illustrates the electric field distributions of the first three modes indexed by the third suffix, which indicates the number of variations of the electric field in the DR. The component along the -axis has an odd number of variations for the odd modes and has an even number of variations for the even modes. The component is antisymmetric with respect to the -axis for the odd modes and is symmetric for the even modes.

CHANG AND KIANG: DUALBAND SPLIT DIELECTRIC RESONATOR ANTENNA 3157

Fig. 3. Electric field distribution of (a)TE mode, (b)TE mode, and (c)TE mode of a solid DR.

Fig. 4. (a) DRs on a ground plane with gap in between. (b) Panoramic view of the equivalent problem.

A. Field Enhancement by a Gap

Fig. 4(a) shows two rectangular DRs placed on a ground plane, separated by a gap at . At component of the and modes reaches the maximum while that of the mode vanishes. The gap is much smaller than and the resonant modes associated with the single DR formed by filling the gap between the aforementioned two DRs are excited. The air-dielectric interface of the gap is normal to , hence the component is significantly enhanced to satisfy the continuity condition on .

Fig. 5 shows the effect of gap width on the return loss. It is observed that the resonant frequency of the mode increases significantly, while those of the and modes are slightly affected. Note that the band associated with the mode merges with that of the mode.

By image theory, the structure in Fig. 4(a) is equivalent to that in Fig. 4(b) if the ground plane is of infinite extent. The two DRs with a separating gap can be regarded as an inhomogeneous DR with permittivity . The gap width is assumed much smaller than , hence the field distribution inside the single in-homogeneous DR is almost the same as that without the gap, except the normal electric field inside the gap is enhanced to satisfy the air-dielectric continuity condition. Thus, the fields of

Fig. 5. Effect of gap widthp on return loss, a = 28 mm, b = 9 mm, d = 10 mm, = 20; w = 2 mm, L = 10 mm, L = 8 mm, d = 7 mm, W = L = 70 mm, t = 0.6 mm, w = 1.15 mm. ( —) p = 0 mm. (- - - ) p = 0.2 mm. (0 1 0)p = 0.4 mm. (0 2 0) p = 0.5 mm.

TABLE I

COMPARISON OFRESONANTFREQUENCYSHIFTDUE TOGAPWIDTH; FREQUENCYUNIT: GIGAHERTZ, LENGTHUNIT: MILLIMETER

the and modes in the air gap can be approximated as

(8) Note that the component is enhanced by a factor . For the mode, approaches as the gap width is very small. For the mode, it is observed that the component is only slightly enhanced, incurring a small of about 2 to 3. Hence, the resonant frequency of the mode is slightly increased. In contrast, the fields of the modes in the air gap are approximately

(9) Substituting (4) and (8) with and , respec-tively, into (3), the resonant frequencies of the and modes can be estimated. Substituting (7) and (9) with

into (3), the resonant frequency of the mode can be esti-mated. The theoretical prediction and simulated results are sum-marized in Table I. The resonant frequency of the DR is also af-fected by the feeding position, resulting in a deviation between

3158 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 55, NO. 11, NOVEMBER 2007

Fig. 6. (a) DR on ground plane with tunnel engraved at its bottom. (b) Equiv-alent problem of DR in free space with a tunnel.

simulation and prediction. Note that the increments of resonant frequency listed in the parentheses match reasonably well.

The radiation patterns can be determined from the tangential electric fields on the DR surfaces. Since the electric field

distri-bution of the mode has opposite

direc-tions on different pordirec-tions of the DR top surface, a null in the pattern occurs in the -direction. The resonant frequencies of the

and modes move closer as is increased and the two bands are merged at 0.5 mm. However, due to the differ-ence of radiation pattern, it is preferred to separate the band asso-ciated with the mode from that with the mode. B. Effect of an Air Tunnel

Based on (3), the resonant frequency of the mode can be shifted away from that of the mode if an air tunnel is engraved at where the electric field of the mode is strong while that of the mode is negligible. As shown in Fig. 6(a), an air tunnel is engraved at the center bottom of the DR with the dimensions of . The effect of the tunnel half-width is shown in Fig. 7. The resonant frequency of the mode is increased as and increase, while those of the and modes are almost unaffected since their electric field at the tunnel is weak.

Fig. 6(b) shows an equivalent problem in free space by dou-bling the heights of the DR and the tunnel using the image theory. Since the electric field of the and the modes rotates about the -axis, the field is tangential to the air-dielectric interface of the tunnel. Hence, it is reasonable to

assume that and .

As for the mode, the tunnel is located at where the electric field reaches the maximum. The component is en-hanced in the tunnel and can be approximated as

(10) By observing the simulated field distributions and fitting the

data, we record and at 0.5 mm,

and at 4 mm. Substituting (7) and

Fig. 7. Effect ofs on return loss, a = 28 mm, b = 9 mm, d = 10 mm, p = 0 mm,d = 4 mm,  = 20; w = 2 mm, L = 10 mm, L = 8 mm, d = 7 mm,W = L = 70 mm, t = 0.6 mm, w = 1.15 mm. ( —) s = 0.5 mm. (- - - )s = 1 mm. (0 1 0) s = 1.5 mm. (0 2 0) s = 2 mm.

TABLE II

COMPARISON OFRESONANTFREQUENCYSHIFTDUE TOTUNNELWIDTH; LENGTHUNIT: MILLIMETER, FREQUENCYUNIT: GIGAHERTZ

(10) with into (3), the resonant frequency shift of the mode is predicted. The simulated and predicted re-sults are summarized in Table II. The tunnel has stronger effect on the resonant frequency of the mode than that of the and modes. Hence, the effect of tunnel height of 0.5 mm and 4 mm, respectively, is investigated and listed in Table II. It is observed that the is strongly enhanced by fold as the tunnel is thin. The resonant frequency of the mode is 3.646 GHz. The increments of resonant fre-quency listed in the parentheses match reasonably well. C. Modification by Engraving Notches

Since the component of the , and

modes reaches maximum at , their resonant frequen-cies should be affected by notches near . Fig. 8(a)

CHANG AND KIANG: DUALBAND SPLIT DIELECTRIC RESONATOR ANTENNA 3159

Fig. 8. (a) Grounded dielectric resonator with two notches on its edges. (b) Panoramic view of an isolated DR with one notch.

Fig. 9. Effect ofs on return loss, a = 28 mm, b = 9 mm, d = 10 mm,  = 20; w = 2 mm, L = 10 mm, L = 8 mm, d = 7 mm, W = L = 70 mm,t = 0.6 mm, w = 1.15 mm. ( —) s = 0.5 mm. (- - - ) s = 1 mm. (0 1 0) s = 1.5 mm. (0 2 0) s = 2 mm.

TABLE III

COMPARISON OFRESONANTFREQUENCYSHIFTDUE TONOTCHDEPTH m = 1:5; LENGTHUNIT: MILLIMETER, FREQUENCYUNIT: GIGAHERTZ

shows a grounded DRA with two notches engraved around its edge. The notches will distort the electric field distribution and the -factor of the DR will decrease, incurring a wider impedance bandwidth. Fig. 9 shows that the resonant frequencies of the three modes are increased by increasing the notch depth . By image theory, the grounded DR with two notches is equiv-alent to an isolated DR with four notches on its edges. First, con-sider only one notch of dimensions engraved off a DR in free space, as shown in Fig. 8(b). The electric field within the notch is more complicated since both and components exist. The simulation show that the component is stronger

Fig. 10. Return loss,a = 28 mm, b = 9 mm, d = 10 mm, p = 1 mm, d = 4 mm,s = 2 mm, d = 4 mm, s = 2 mm,  = 20; h = 4 mm, w = 2 mm,L = 10 mm, L = 2.5 mm, d = 4 mm, W = L = 70 mm, t = 0.6 mm,w = 1.15 mm. ( —) Measurement. (- - - ) Simulation.

Fig. 11. Electric field distribution at (a) 3.45 and (b) 5.26 GHz.

than the component. The component normal to the air-di-electric interface of the notch is enhanced to satisfy the conti-nuity condition and can be approximated as

for and modes (11)

for modes (12)

With 4 mm, is about 1.5. Substituting (4) and (11) into (3), the resonant frequencies of the DR with notches are ob-tained. The predicted and the simulated results are summarized in Table III. The prediction for the mode is less accurate, but the increasing trend is consistent.

3160 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 55, NO. 11, NOVEMBER 2007

Fig. 12. Radiation patterns atf = 3.45 GHz: (a) xy-plane and (b) xz-plane. ( —) MeasuredE . (- - - ) Measured E (0  0) Simulated E . (0 2 0) SimulatedE , the gain at  = 90 and  = 0 is 5.6 dBi, 10-dB per division on radials; all parameters are the same as in Fig. 10.

V. DESIGNWITHCOMBINATION

The design begins with a rectangular DR of dimension 10 mm

9 mm 29 mm, 7 mm, 8 mm, 2 mm, and

10 mm. The resonant frequencies of the , and modes are 2.92, 3.58, and 4.62 GHz, respectively. In order to tune the resonant frequencies of the and modes to cover the WiMax (3.4–3.7-GHz) and the WLAN (5.15–5.35-GHz) bands, the DR is modified to the shape as shown in Fig. 1(a), with 1 mm, 4 mm, and 2 mm. The resonant frequencies of the three modes are shifted to 3.58, 4.3, and 5 GHz, respectively. By adjusting

Fig. 13. Radiation patterns atf = 3.6 GHz: (a) xy-plane and (b) xz-plane. ( —) MeasuredE . (- - - ) Measured E . (0  0) Simulated E . (0 2 0) SimulatedE , the gain at  = 90 and  = 0 is 3 dBi, 10-dB per division on radials; all parameters are the same as in Fig. 10.

the offset , the extended length of microstrip line , and the length of the aperture , the DR can be matched to 50 microstrip line feed, with the resonant frequencies slightly af-fected by the feeding structure. Fig. 10 shows the measured and simulated return loss. There are three bands over 3.375–3.93 GHz (15%), 4.6–4.79 GHz (4%), and 5.08–5.415 GHz (6%), associated with the , and modes, respec-tively. The first band covers the WiMax (3.4–3.7 GHz) and the third band covers the WLAN (5.15–5.35 GHz).

Fig. 11 shows the electric field distributions over the first and the third bands, respectively. The third resonant band around

CHANG AND KIANG: DUALBAND SPLIT DIELECTRIC RESONATOR ANTENNA 3161

Fig. 14. Radiation pattern at 5.265 GHz: (a)xy-plane and (b) xz-plane. ( —) MeasuredE . (- - - ) Measured E . (00) Simulated E . (020) Simulated E , the gain at  = 90 and  = 0 is 7.2 dBi, 10-dB per division on radials; all parameters are the same as in Fig. 10.

5.265 GHz is associated with the mode. The split DRs can be viewed as two radiators placed closely along the

-direction.

Figs. 12 and 13 show the measured and simulated radiation patterns at 3.45 GHz and 3.6 GHz, respectively. On the -plane, the component is stronger than the component by about 10 dB over , the maximum gain is 5.6 dBi at 3.45 and 3 dBi at 3.6 GHz. The gain at 3.6 GHz is lower because the main beam of the pattern is slightly tilted on the -plane.

On the -plane, the component is stronger than the component by 10 dB over and the maximum gain is 6.5 dBi at 3.45 GHz and 6 dBi at 3.6 GHz. The front-to-back ratio is about 10 dB.

Fig. 14 shows the measured and simulated radiation patterns at 5.265 GHz. On the -plane, the component is stronger than the component by about 10 dB over

, and the front-to-back ratio is about 12 dB. The an-tenna gain is 7.22 dBi, which is higher than that at 3.45 and 3.6 GHz because the beam of the pattern on the -plane is narrower and is slightly tilted to . The gain at the beam direction is 8.4 dBi.

The efficiency is about 94% for the and modes. The insertion loss of the feeding microstrip is about 0.5 dB due to the substrate loss. For WiMax or WLAN applications, this DRA can be mounted on a vertical wall with the -axis pointing to zenith, providing a broadside, vertically polarized radiation pattern in front of the wall ( -direction).

VI. CONCLUSION

A dualband DRA is proposed, which is composed of two notched DRs separated by a narrow air gap. The effects of gap and notches on the resonant frequency shift are carefully studied. Two bands are attained in 3.375–3.93 GHz (15%) and 5.08–5.415 GHz (6%) with broadside pattern on the -plane; the third band in 4.6–4.79 GHz (4%) is not practical. The proposed DRA can be used in the WiMAX (3.4–3.7 GHz) and WLAN (5.15–5.35 GHz) bands.

REFERENCES

[1] S. A. Long, M. W. McAllister, and L. C. Shen, “The resonant cylin-drical dielectric cavity antenna,” IEEE Trans. Antennas Propag., vol. AP-31, no. 3, pp. 406–412, May 1983.

[2] T. A. Denidni, Q. Rao, and A. R. Sebak, “Multi-eccentric ring slot-fed dielectric resonator antennas for multi-frequency operations,” in IEEE Int. Symp. Antennas Propag., Jun. 2004, vol. 2, pp. 1379–1382. [3] T. A. Denidni and Q. Rao, “Hybrid dielectric resonator antennas with

radiating slot for dual-frequency operation,” IEEE Antennas Wireless Propag. Lett., vol. 3, no. 1, pp. 321–323, Dec. 2004.

[4] A. Buerkle, K. Sarabandi, and H. Mosallaei, “Compact slot and dielec-tric resonator antenna with dual-resonance, broadband characteristics,” IEEE Trans. Antennas Propag., vol. 53, no. 3, pp. 1020–1027, Mar. 2005.

[5] T. W. Li and J. S. Sun, “Dual-frequency dielectric resonator antenna with inverse T-shape parasitic strip,” in IEEE Appl. Comput. Electro-magn. Soc. Int. Conf., Apr. 2005, pp. 384–387.

[6] L. T. Wei and S. J. Shiun, “Wideband dielectric resonator antenna with parasitic strip,” in IEEE Appl. Comput. Electromagn. Soc. Int. Conf., Apr. 2005, pp. 376–379.

[7] B. Li and K. W. Leung, “Strip-fed rectangular dielectric resonator an-tennas with/without a parasitic patch,” IEEE Trans. Anan-tennas Propag., vol. 53, no. 7, pp. 2200–2207, Jul. 2005.

[8] C. S. D. Young and S. A. Long, “Investigation of dual mode wide-band rectangular and cylindrical dielectric resonator antennas,” in Proc. IEEE Int. Symp. Antennas Propag., Jul. 2005, vol. 4, pp. 210–213. [9] A. A. Kishk, “Wide-band truncated tetrahedron dielectric resonator

an-tenna excited by a coaxial probe,” IEEE Trans. Anan-tennas Propag., vol. 51, no. 10, pp. 2913–2917, Oct. 2003.

[10] A. A. Kishk, Y. Yin, and A. W. Glisson, “Conical dielectric resonator antennas for wide-band applications,” IEEE Trans. Antennas Propag., vol. 50, no. 4, pp. 469–474, Apr. 2002.

[11] J. I. Moon and S. O. Park, “Dielectric resonator antenna for dual-band PCS/IMT-2000,” Electron. Lett., vol. 36, pp. 1002–1003, Jun. 2000. [12] Z. Fan and Y. M. M. Antar, “Experimental investigation of

multi-el-ement dielectric resonator antennas,” in IEEE Int. Symp. Antennas Propag., Jul. 1996, vol. 3, pp. 2034–2037.

3162 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 55, NO. 11, NOVEMBER 2007

[13] Z. Fan and Y. M. M. Antar, “Slot-coupled DR antenna for dual-fre-quency operation,” IEEE Trans. Antenna Propag., vol. 45, no. 2, pp. 306–308, Feb. 1997.

[14] A. A. Kishk, B. Ahn, and D. Kajfez, “Broadband stacked dielectric res-onator antenna,” Electron. Lett., vol. 25, no. 18, pp. 1232–1233, Aug. 1989.

[15] K. Pliakostathis and D. Mirshekar-Syahkal, “Stepped dielectric res-onator antennas for wideband applications,” in IEEE Int. Symp. An-tennas Propag., Jun. 2004, vol. 2, pp. 1367–1370.

[16] R. F. Harrington, Time-Harmonic Electromagnetic Fields. New York: McGraw-Hill, 1961.

[17] H. Y. Yee, “Natural resonant frequencies of microwave dielectric res-onators,” IEEE Trans. Microw. Theory Tech., vol. MMT-13, no. 2, pp. 256–256, Mar. 1965.

[18] A. K. Okaya and L. F. Barash, “The dielectric microwave resonator,” Proc. IRE, vol. 50, no. 10, pp. 2081–2092, Oct. 1962.

[19] T. Itoh and C. Chang, “Resonant characteristics of dielectric resonators for millimeter-wave integrated circuits,” in Proc. IEEE Microw. Theory Tech. Soc. Int. Symp., Jun. 1978, vol. 78, pp. 121–122.

[20] R. K. Mongia, “Theoretical and experimental resonant frequencies of rectangular dielectric resonators,” in Proc. Inst. Electr. Eng. Microw. Antennas Propag., Feb. 1992, vol. 139, no. 1, pp. 98–104.

[21] R. K. Mongia and A. Ittipiboon, “Theoretical and experimental investi-gations on rectangular dielectric resonator antennas,” IEEE Trans. An-tennas Propag., vol. 45, no. 9, pp. 1348–1356, Sep. 1997.

[22] Y. M. M. Antar, D. Cheng, G. Seguin, B. Henry, and M. G. Keller, “Modified waveguide model (MWGM) for rectangular resonator an-tenna (DRA),” Microw. Opt. Tech. Lett., vol. 19, no. 2, pp. 158–160, Oct. 1998.

Tze-Hsuan Chang (S’01) was born in Hsin-Chu, Taiwan, on February 1, 1978. He received the B.S. degree in electrical engineering from the National Chung Hsing University, Taichung, Taiwan, in July 2000 and the Ph.D. degree in electrical engineering from in the Graduate Institute of Communication Engineering, National Taiwan University, Taipei, Taiwan, in July 2007.

Jean-Fu Kiang (M’89) was born in Taipei, Taiwan, on February 2, 1957. He received the B.S. and M.S. degrees from National Taiwan University, Taipei, Taiwan, and the Ph.D. degree from Massachusetts Institute of Technology, Cambridge, in 1979, 1981, and 1989, respectively, all in electrical engineering.

He was with Schlumberger-Doll Research, Ridge-field, CT, in 1985–1986, IBM Watson Research Center, Yorktown Heights, NY, in 1989–1990, Bell-core, Red Bank, NJ, in 1990–1992, Siemens Elec-tromedical Systems, Danvers, MA, in 1992–1994, and National Chung-Hsing University, Taichung, Taiwan, in 1994–1999. He has been the Professor at the Department of Electrical Engineering and the Graduate Institute of Communication Engineering, National Taiwan University since 1999. His research interests are the applications and system issues on electromagnetics, including wireless communications, antennas, electromag-netic compatibility, microwave components, etc.

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES 1

A K-Band CMOS Low-Noise Amplifier with Low

DC Power Consumption

Ping-Yuan Deng, Student Member, IEEE and Jean-Fu Kiang, Member, IEEE

Abstract— A K-band low-noise amplifier (LNA) is designed and

fabricated in a standard 0.18 µm CMOS technology. A design method of CMOS LNA is used to render the optimum source resistance (Ropt) close to 50 Ω and Zin = Zopt∗ by using small

devices and small bias currents. This LNA chip achieves a peak gain of 13.5 dB and a noise figure of 4.7 dB at 24 GHz. The supply voltage and supply current are 1 V and 8.3 mA, respectively. The input and output return loss are lower than −10 dB. The input referred 1-dB compression ( P1dB) is −7 dBm. The chip size is

0.64 mm × 0.48 mm.

Index Terms— CMOS, K-band, low-noise amplifier (LNA),

microwave monolithic integrated circuit (MMIC).

I. INTRODUCTION

A

T frequencies above 20 GHz, GaAs-based HEMT and HBT processes dominate most of the applications in the past. With the rapid advance of CMOS technologies, it is becoming plausible to implement RF systems operating at 20 GHz and higher using CMOS technologies. RF and baseband circuits realized in CMOS are expected to reduce the cost of systems such as FMCW (24 GHz ISM band) [2], ultra-wideband (UWB, 22-29 GHz) short-range radars [3], and local multipoint distribution systems (LMDS). Low-noise amplifier is a key module for any RF system. CMOS LNAs have been designed for frequencies above 20 GHz [4]-[6]. LNAs with low noise figures (NFs) are usually achieved at the expense of high dc power consumption.

In this paper, we present a design method of CMOS low-noise amplifiers to achieve simultaneous low-noise matching and power matching by tailoring the device size. It is found that Ropt close to 50 Ω and Zin = Zopt∗ can be obtained

by using small devices and small currents. This design also achieves acceptable power gain, low noise figure and low power consumption. Circuit design considerations of LNA will be presented in Section II. Simulation and measurement results will be discussed in Section III, followed by the conclusions.

II. CIRCUITDESIGN A. General Considerations

Fig. 1 shows the two-port network of a microwave amplifier. The transducer power gain GT and available power gain GA

can be derived as [1] GT = 1 − |ΓS| 2 |1 − S11ΓS| 2|S21| 2 1 − |ΓL| 2 |1 − ΓoutΓL| 2 GA= 1 − |ΓS|2 |1 − S11ΓS| 2|S21| 2 1 1 − |Γout| 2

Fig. 1. Two-port network of a microwave amplifier.

(a) (b)

Fig. 2. LNA topology, (a) common-source amplifier, (b) cascode amplifier.

The transducer gain GT is affected by the input and output

matching networks. In the LNA design, the input matching network transforms Z1 to Zopt (ΓS = Γopt) to achieve

min-imum noise figure NFmin. If the load reflection coefficient ΓL is adjusted to transfer maximum power to the load (ΓL =

Γ∗out), then GT = GA. In practice, input matching in the LNA

design usually involves trade-offs among power gain, noise figure, and VSWRs .

Many LNA topologies have been proposed in the literatures [5], [7]. The most frequently used LNA topology is the common-source (CS) and cascode configurations as shown in Fig. 2. The CS-stage can exhibit minimum noise figure with a proper input matching, and it can work at low supply voltage below 1 V. The cascode amplifier has higher power gain and reduced Miller effect, but it can not work at low supply voltage. Compared to a common-source amplifier, the cascode amplifier has a poorer noise performance especially when the operation frequency is close to fT. Hence, the

CS-stage is chosen in this work.

B. Noise Matching with Power Matching

Fig. 3 (a) shows a common-source amplifier with source inductor Ls1 as a feedback. Fig. 3 (b) shows the

small-IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES 2

(a) (b)

(c)

Fig. 3. (a) Schematic of the source inductive degeneration amplifier, (b) small-signal equivalent circuit of the input matching network of (a), where ω_T1= gm1/Cgs1, (c) gain circle and noise circle on the Smith chart.

signal equivalent circuit of the input impedance in Fig. 3 (a). Neglecting the gate-drain capacitance, source-bulk capacitance and parasitic resistance of Ls1, the input impedance can be

calculated as Zin' gm1Ls1 Cgs1 + sLs1+ 1 sCgs1

Thus, proper choice of gm1, Ls1, and Cgs1will yield an input

impedance Zin= Rin− jX, where Rin= gm1Ls1/Cgs1= 50 Ω and X = 1/ωCgs1− ωLs1. In practice, the last two terms

may not resonate at the frequency of interest. By selecting appropriate Ls1, the input impedance will be moved close to

the constant 50 Ω circle in the Smith chart, thereby simplifying the input matching network [7]. Another advantage of this technique is that the gain circles around the maximum gain and the noise circles around the minimum noise figure become closer as shown in Fig. 3 (c) [8], [9].

The equivalent transconductance can be expressed as

g0m1=

gm1

1 + jωLs1(gm1+ jωCgs1)

where Cgs1 is the gate-to-source capacitance, and gm1 is the

transconductance without inductive source degeneration. The inductive source degeneration will make the transistor more resistive and increase the linearity of M1, at the cost of

decreasing its equivalent transconductance.

In the LNA design, the source impedance is chosen as ZS=

Zoptfor noise matching to reach NFmin. The input impedance

is chosen as Zin= ZS∗to achieve power matching. By tailoring

the device size of M1and the value of Ls1, we can make Ropt

close to 50 Ω and Zin= Zopt∗ with a small bias current, thus

(a) (b)

Fig. 4. (a) Schematic of source inductor feedback amplifier with series inductor, (b) effect of adding Lg1.

(a) (b)

Fig. 5. (a) Schematic of source inductor feedback amplifier with shunt inductor and series capacitor, (b) effect of adding Lg1and C1.

the dc power consumption can be reduced. In other words, if the conditions

Re (ZS) = Re (Zopt) = 50 Ω Zin= ZS∗ = Z

∗ opt

are met, then the goals of high gain, low noise figure, reason-able VSWRs, and low power consumption can be achieved simultaneously. Because Zin is frequency independent, thus

simultaneous noise and power matching can be achieved in a specific frequency band.

C. Input Matching Network

Fig. 4 (a) shows the schematic of the source inductor feed-back amplifier with the gate inductor Lg1for input impedance

matching [9]. To achieve noise matching, ZS = Zopt, and

power matching, Zin = ZS∗ = Z

∗

opt = Ropt − jX, an

additional gate inductor Lg1 is used to resonate with −jX

at the intended frequency, as shown in Fig. 4 (b). However, the inductance Lg1 is about 0.5∼1 nH at 24 GHz, and the

noise performance of Lg1is sensitive to its parasitic resistance.

Additional bias network such as a large shunt resistor (or RFC) and large dc-block capacitors are required. Consequently, an alternative input matching network is required at this fre-quency range.

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES 3

Fig. 6. Schematic of K-band LNA.

Fig. 5 (a) shows the proposed schematic of source inductor feedback amplifier with shunt inductor and series capacitor for input impedance matching. Proper device size of M1is chosen

to tune the source impedance into ZS = Zopt= Ropt+ jX = 50+jX. At conjugate matching (Zin = Zopt∗ ) as shown in Fig.

5 (b), a shunt inductor Lg1 will change the negative reactance

to a positive reactance jX on the same constant-conductance circle on the Smith chart, the series capacitance C1 is then

used to resonate with jX. The combination of Lg1 and C1

transfers RS to Zopt, Lg1 and C1 also act as part of the bias

network and dc block, respectively.

By using this approach, the required value of Lg1 is about

0.2∼0.3 nH, which can be practically implemented in Si-based technology at frequency above 20 GHz. A bypass capacitor

Cb is added to stabilize the supply voltage Vg1and to isolate

the noise to it. Thus, no additional bias networks such as large shunt resistors and large dc-block capacitors are required, which leads to a smaller chip size and lower noise figure.

D. Design of 24 GHz LNA

Since a single transistor does not generate enough gain at high frequencies, the three-stage cascaded common-source structure is proposed as shown in Fig. 6. In the first stage, proper choice of gm1and Cgs1of M1and Ls1are used to shift

the input impedance for noise matching. In order to reduce the power consumption, the device size of M1should be as samll

as possible to reduce the bias current. In this work, M1 is

designed to have 11 fingers with the total gate width of 33

µm, and is biased at 1 V with drain current of 3 mA. Under

these conditions, the input impedance Zinis conjugate to Zopt.

Thus, minimum noise figure, high power gain, low bias current and good input impedance matching (VSWR'1) are achieved simultaneously.

The shunt inductor Lg1and series capacitor C1are used to

conjugate match the input impedance. Higher gain of the first stage will suppress the noise contribution of the subsequent stage, leading to a better noise performance. In this work, we choose Lg1= 0.28 nH and C1= 112.3 fF.

Common-source amplifiers with inductive degeneration are used as the second and the third stages to increase the overall

(a) (b)

Fig. 7. Stability concern: (a) Feedback through supply loop or nonideal ground, (b) large bypass capacitors are used to quench the oscillation at low frequency.

Fig. 8. Die micrograph of the 24 GHz LNA, the chip size is 0.64 mm × 0.48 mm.