IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 20, NO. 2, FEBRUARY 2010 67
New Miniaturized Dual-Mode Dual-Band Ring
Resonator Bandpass Filter With
Microwave
C-Sections
Yi-Chyun Chiou, Member, IEEE, Cho-Yu Wu, and Jen-Tsai Kuo, Senior Member, IEEE
Abstract—A new miniaturized dual-mode dual-band ring
res-onator bandpass filter is implemented by a cascade of several mi-crowave -sections. Each -section is used to substitute a trans-mission line section of designated electrical length. Through proper design of input/output coupling configuration, two transmission zeros can be created on both sides of each passband. Two circuits with four and six microwave -sections are fabricated for confir-mation. They occupy less than 30% of the area of a traditional ring resonator filter. Measured responses agree very well with the sim-ulation.
Index Terms—Bandpass filter (BPF), dual-band, dual-mode,
mi-crowave -section, miniaturized.
I. INTRODUCTION
R
ING resonator filters have been widely used in front-ends of microwave systems due to their compact size and easy design [1]–[3]. The circuit in [1] could be the first ring resonator bandpass filter (BPF) with two degenerate modes that are of essential importance for constituting the passband. The quasi-elliptic BPF in [2] is designed to have tunable transmission zeros with a constant bandwidth. In [3], a periodic stepped-impedance ring resonator is devised to develop a miniaturized dual-mode filter with a wide upper stopband. Notice that all these circuits involve BPFs with a single passband.Recently, rapid development of modern wireless systems, such as GSM and WLAN, has created a need for dual-band RF devices. Several dual-mode dual-band BPFs have been published [4]–[8]. In [4], the stacked-loop structure consists of two dual-mode rings on different layers and each ring resonator controls one passband. Based on a similar idea, an alternative dual-band filter with a coplanar waveguide (CPW) feed line is carried out in [5]. In [6], a loop resonator is proposed for a planar dual-band filter. The feed lines are placed between two resonators to offer sufficient coupling for both passbands. The dual-band filter in [7] is designed in a multilayer structure con-sisting of dual-mode resonators in a reflector cavity. It is noted
Manuscript received August 06, 2009; revised November 17, 2009. First pub-lished January 19, 2010; current version pubpub-lished February 10, 2010. This work was supported in part by the MoE ATU program and by the National Sci-ence Council of Taiwan under Grants NSC 98-2221-E-009-032-MY2 and NSC 98-2218-E-009-011.
The authors are with the Institute of Communication Engineering, National Chiao Tung University, Hsinchu 300, Taiwan (e-mail: [email protected]). Color versions of one or more of the figures in this letter are available online at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/LMWC.2009.2038432
that each selected passband in [4]–[7] is mainly controlled by a resonator. The dual-mode dual-band filter in [8] is contrived with a single stepped-impedance ring resonator.
In this letter, a dual-mode dual-band BPF is implemented with microwave- sections, which have nonlinear phase shift property in frequency [9] and are suitable for development of dual-band devices [10]. Here, each microwave -section is used to substitute a - or -section of a traditional ring. Analysis will be conducted and design curves plotted for facilitating the circuit realization. Emphasis is also put on the input/output con-figuration for creating transmission zeros on both sides of each passband. Two circuits are simulated, fabricated and measured for confirmation.
II. FORMULATION
Fig. 1 shows the elementary two-port for constructing the dual-mode dual-band filter. It consists of two transmission line sections of length and characteristic impedances with a microwave -section of electric length in between. In our approach, a uniform ring is treated as a cascade of identical sections and each of them will be implemented by the network in Fig. 1. Then, let the two designated frequencies be and , and the characteristic impedance of the ring periph-eral be . For simplicity, is chosen, where and are the even- and odd-mode characteristic impedances of the coupled-line. By enforcing the ma-trix of the two-port to be equal to those of a uniform section of and at and , respectively, the following four equations can be readily obtained:
(1a) (1b) (1c) (1d) where (2a) (2b) Based on (1) and (2), Fig. 2 plots , and as functions of for with and with . Here, is chosen and and are, respectively, and normalized with respect to , the reference port impedance. Such a high is chosen since narrow lines are capable of providing sufficient coupling to our dual-band design. In Fig. 2, and gradually decrease and increase,
68 IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 20, NO. 2, FEBRUARY 2010
Fig. 1. Microwave-C section for substituting a transmission line section.
Fig. 2. z , z and as functions of n.
respectively, when is increased from 1.5 to 2. It is noted that when , the -section has . This means that the two-port becomes a folded uncoupled section at , which in turns implies that our structure is limited to .
III. IMPLEMENTATION, SIMULATION ANDMEASUREMENT
Fig. 3(a) depicts the layout of the filter with four -sections on a substrate with and . The perturbation patches and are used to split off the degen-erate modes at the two designated frequencies (
and ). Fig. 3(b) plots the simulated results [12] with for testing the electrical length (eval-uated at ), where the circuit is weakly coupled by the feed lines. One can see that all curves possess a dual-resonance response at and with two transmission zeros on both the upper and lower sides. Note that only one resonant peak exists and no zero around when , since one of the two de-generate modes has null voltage at the excitation position and it is not activated. Herein, rather than 45 is used since enough space should be saved for .
Table I lists the resonant frequencies when the patches are changed, showing the control of the two bandwidths. The patches are placed at the two symmetric planes of the resonant modes. At , can change the even mode frequency but has no influence on the odd mode, where the voltage is zero. This is exactly the same as the function of the perturbation patch in design of the traditional ring resonator filter [2]. The impact of
Fig. 3. Filter with fourC-sections. (a) Circuit layout. (b) jS j response with ` = ` = 0. (c) Simulated and measured results. Circuit dimensions in mm:S = 0:164, W = 0:25, W = W = S = 0:15, S = 0:188, ` = 3:81, ` = 3:23, ` = 1:57, ` = 1:5. A = 1:7 2 0:418 mm , A = 0:715 2 0:4 mm .
the changes of on the resonance modes at and can be explained in a similar way.
The interdigital structure is used to provide cou-pling from the feed line to the circuit. It is found that the first band needs more coupling to establish the passband. Increasing , however, will decrease the bandwidth at . We had such ex-perience in developing the multi-mode filter in [11]. Then, are increased to compensate this effect. This will decrease the bandwidth at ; this in turns contributes a positive factor for establishing a passband with satisfactory return loss. Fig. 3(c) plots the simulated and measured results of the experimental circuit. The fractional bandwidths and at and are 3.03% and 6.97%. The measurement shows that the insertion
CHIOU et al.: NEW MINIATURIZED DUAL-MODE DUAL-BAND RING RESONATOR BPF 69
TABLE I
CONTROL OFTWOBANDWIDTHSWITHA ANDA
losses are 2.56 dB and 1.45 dB, and return losses 23.5 and 26.4 dB in the first and the second passbands, respectively. The circuit occupies only 27.3% of the area of a traditional ring res-onator filter designed at .
Fig. 4(a) plots the layout of the second dual-mode dual-band BPF with six microwave sections, and Fig. 4(b) shows the sim-ulated and measured results. The circuit is designed to have and with and , respectively. At and , the measured are 2.39 dB and 1.56 dB, respectively, and the in-band return losses are better than 15 dB. The transmission zeros are at 1.93, 2.57, 4.35, and 5.49 GHz. The circuit has 29.1% of the area of a traditional ring filter. Good agreement between simulated and measured results can be observed.
IV. CONCLUSION
New miniaturized dual-mode dual-band ring resonator BPFs are developed by cascading the microwave -sections. The cir-cuits use less than one-third of the area of a conventional ring resonator BPF designed at the first frequency. The two exper-imental circuits demonstrate good inband insertion losses and return losses. In addition, two transmission zeros are created on both sides of each passband. The shortcoming of this circuit structure is the limited tuning range of the second frequency.
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Fig. 4. Dual-mode dual-band BPF with six microwaveC-sections. (a) Circuit layout. (b) Simulated and measurement results. Circuit dimensions in mm:W = 0:148, S = 0:157, W = 0:25, W = S = S = 0:15, ` = 2:51, ` = 2:01, ` = 1:94, ` = 1:44. A = 1:1 2 0:7 mm , A = 0:9 2 0:47 mm . = 60 .
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