Published in IET Electric Power Applications Received on 4th January 2013 Revised on 17th February 2013 Accepted on 4th March 2013 doi: 10.1049/iet-epa.2013.0002 ISSN 1751-8660
Design of buck-type current source inverter fed
brushless DC motor drive and its application to
position sensorless control with square-wave current
Hung-Chi Chen, Hung-He Huang
Department of Electrical and Computer Engineering, National Chiao Tung University, HsinChu, Taiwan E-mail: hcchen@cn.nctu.edu.tw
Abstract: Owing to the widely used brushless DC motors (BDCMs) in high-efficiency applications, many position sensorless control methods based on voltage source inverters had been developed in the literature. Recently, current source inverters (CSIs) are receiving more and more attention because of their inherent short-circuit protection characteristics. But no position sensorless control for buck-type CSI with square-wave current had been found in the literature. In this study, the buck-type CSI-fed BDCM drive is designed and its application to the square-current position sensorless control is first proposed. The provided simulation and experimental results verify the effectiveness of the proposed CSI-based position sensorless control.
1 Introduction
Owing to the permanent-magnetic rotor field, brushless DC motors (BDCMs) possess higher efficiency than the popular induction motors. Therefore more and more BDCMs are used in various high-efficiency variable-speed applications, such as fan motors [1, 2], compressor motors [3, 4], vehicle motor [5–7] and home applications [8].
In the normal square-wave current operations of BDCMs, the discrete rotor position should be monitored by position sensors to yield adequate current commutations [9]. The six-switch voltage source inverter (VSI) fed BDCM is plotted in Fig. 1 where the feedback position signals are used to synthesise the switching signals. The DC voltage amplitude can be equivalently varied via the pulse width modulation (PWM) ratio by the speed controller. In order to avoid short-circuit condition, both switches in the same leg could not conduct in the same time. A 120° conduction (six-step) method [3–9] and 150° conduction (12-step) method [1,2] are two commutation schemes for six-switch VSI. Additionally, various VSI topologies can also be found in [10,11].
Although VSIs are widely used in motor drives, reliability concerns have been raised on the motor because of the high dv/dt that comes from the PWM output voltage. Voltage surges resulting from these rapid voltage transitions can cause motor insulation degradation, bearing failure because of erosion caused by the resulting shaft leakage current, and unacceptable electromagnetic interference effects on the control circuits, as well as acoustic noise in the motor.
In addition, the concern about the reliability of the electrolytic capacitor has forced user to use costly and bulky film capacitors [12]. Moreover, the possible
shoot-through problem in VSI has always been a concern associated with system reliability.
The other topology is the current source inverter (CSI). CSI uses an inductor as the energy storage component, and thus avoids many drawbacks of VSI. It also has an inherent advantage of the shoot-through short-circuit protection capability, and no PWM voltage in the motor terminals. The bulky inductor has longer lifetime than the capacitors.
In fact, CSI topology had been widely used in high-power application. The common thyristor-based load commutated inverter (LCI) topology has been reported for BDCM drive [10]. Four quadrant operation and current sensorless control over most of the operating speed range are good features of this topology. However, thyristor-based CSI is suitable for the high-power utility-connected industry applications, but it is not suitable for kilowatt (kW)-level residential applications, such as electric vehicles and variable-speed air conditioners.
Recently, more and more researches are focused on insulated gate bipolar transistor (IGBT)-based CSI drive for the automotive applications [5–7]. The common IGBT-based CSI for BDCM is plotted in Fig. 2 where the current source is controlled by the speed controller. In order to provide current conducting path, at least one of the three upper switches and at least one of the three lower switches need to turn on at the same time. The series-connected diodes need to withstand the negative voltages. Three AC capacitors are connected between the CSI and BDCM, and they provide the current flowing path during the current commutations [5–7].
For square-wave current commutations in BDCM drive, the hall position sensors are required to provide the position information. However, the hall sensors may be faulted and may not be used in high-temperature environment, such as refrigerant compressors.
It is difficult to make clear comparison between CSI and VSI. For potential applications, such as electric vehicles, system reliability is an important issue. It is clear that the life time of the bulky electrolytic capacitor is shorter than the bulky inductor. To further improve the reliability of the CSI-based drive, position sensorless control is required to be able to ride through sensor-fault conditions.
Owing to the concern of system reliability for BDCM operations, many position sensorless control methods for VSI had been developed in the literature and most of them are based on the back-electromotive force (EMF) detecting methods [1–5,11,13–16]. The method in [11] was developed for a four-switch VSI topology and the others are developed for the common six-switch VSI topology. The current-mode Fig. 1 Conventional VSI-fed BDCMs
sensorless control in [13] should be seen as a VSI-based method because that the terminal voltages had significant PWM voltage transitions.
In VSI-based sensorless control methods, only zero-crossing points of the phase back-EMFs can be detected from the terminal voltages. Therefore delaying these zero-crossing points by 30° is required to obtain the correct commutation instants [11,14–17], but it may introduce some commutation errors.
In CSI, the line-to-line back-EMFs can be easily sensed from the terminal voltages, and thus the commutation instants can be obtained directly without the processing of the additional 30° delay. However, no position sensorless control methods for IGBT-based CSI with square-wave current commutations had been found in literature.
In this paper, the current source in Fig.2is implemented by the buck converter and the buck-type CSI-fed BDCM drive is designed. It is noted that the position sensorless control for buck-type CSI-fed BDCM is first proposed. The provided simulation and experimental results support the validity of the proposed CSI-based position sensorless control.
2 Current source inverter fed BDCM
For servo applications, the torque performance is most important. However, in the promotion of residential variable-speed products, such as air conditioners, the cost issue is more critical to the performance issue. Thus, the manufactures take a simple strategy – producing near-sinusoid-EMF BDCMs to reduce the cost of motor.
For the high-performance series of the air conditioners, the high-cost sinusoid-current drive is applied with the other high-efficiency system techniques. However for the popular series, the low-cost square-current drive is utilised even when the yielded torque is not constant. Thus, the sinusoidal rotor magnets are initially assumed.
2.1 Ideal current source inverter
The variable-speed drive BDCM fed by ideal current source is plotted in Fig. 2, where the current source is variable according to the speed controller output Id∗. According to the winding distribution in Fig.1, theflux linkages of three phase windings can be assumed as φas = ΦMAXsin θe, φbs= ΦMAXsin(θe− 2π/3) and φcs = ΦMAXsin(θe + 2π/3), where ΦMAX is the maximum flux linkage. Thus, the induced voltages for each phase winding can be expressed as
ea =dwas dt = FMAXvecosue= P 2FMAXvrcos P 2ur eb =dwbs dt = P 2FMAXvrcos P 2ur− 2p 3 ec= dwcs dt = P 2FMAXvrcos P 2ur+ 2p 3 (1)
where P is the pole number,ωeis the synchronous frequency in‘rads/s’, and θr = 2θe/P is the rotor position.
When the positionθeis between 0 andπ/3 (0 ≤ θe < π/3), both switches T1 and T2 are conducting and the current Id flows through the a-phase and c-phase windings Id = ia = −ic. The winding currents ia, iband icare in phase with the back-EMF voltages ea, eband ec, respectively.
The voltage esis denoted by (see (2))
where k is an integer and KE is the voltage gain KE = PΦMAX/2. It is clear that the voltage es is a periodic waveform and its average value is es= 3√3KEvr/p.
Since ia+ ib + ic = 0 in Y-connected windings, the neutral voltage vn can be represented in terms of the terminal voltages
vn=va+ vb+ vc
3 (3)
When the switch T3is conducting, the terminal voltage vbis equal to the bus voltage vb = Vbus, and when the switch T6is conducting, the terminal voltage vbmust be zero. When both switches T3and T6 are not conducting, b-phase winding is floating and the terminal voltage vb is equal to the sum of the neutral voltage plus the induced b-phase voltage vb = vn + eb.
Therefore when the positionθeis located between 0 andπ/3 (0 ≤ θe< π/3), both switches T1and T2are conducting. The terminal voltage vacan be expressed as va= Vbus= ea− ec+
2IdRs≃ es and the terminal voltage vcis zero. From (3), the neutral voltage terminal is vn= (Vbus + vb)/3 and thus, the floating b-phase voltage vbis closed to
vb=Vbus 2 + 3 2eb≃ ea− ec 2 + 3 2eb≃ eb− ec≃ 3 √ KEvrsinue when 0≤ue,p/3 (4) The illustrated waveforms are plotted in Fig.3. Three position signals Ha, Hb and Hc are used as the commutation signals to obtain the BDCM speed ωr and generate six switching signals by the common six-step conduction scheme.
2.2 Buck-type current source inverter
In addition, there is always one upper switch and one lower switch at the same time in common six-step commutation and thus, the BDCM can be modelled as a series-connected circuit with an inductance 2Ls, a resistance 2Rs and the voltage es as shown in Fig. 4, where Ls and Rs are the winding inductance and the winding resistance, respectively. LB is the output inductance of the buck converter.
In practice, the current source can be implemented by a buck converter as plotted in Fig. 4 because of the output inductor LB in the buck converter. The bus current Id is
es= ea− ec=√3KEvrcos (ue−p/6), when 2kp ≤ ue, 2kp + p/3 eb− ec=√3KEvrcos (ue−p/2), when 2kp + p/3 ≤ ue, 2kp + 2p/3 eb− ea=√3KEvrcos (ue− 5p/6), when 2kp + 2p/3 ≤ ue, (2k + 1)p ec− ea=√3KEvrcos (ue− 7p/6), when (2k+ 1)p ≤ ue, 2kp + 4p/3 ec− eb=√3KEvrcos (ue− 3p/2), when 2kp + 4p/3 ≤ ue, 2kp + 5p/3 ea− eb=√3KEvrcos (ue− 11p/6), when 2kp + 5p/3 ≤ ue, (2k + 2)p ⎧ ⎪ ⎪ ⎪ ⎪ ⎪ ⎪ ⎨ ⎪ ⎪ ⎪ ⎪ ⎪ ⎪ ⎩ (2)
sensed and a current loop is designed for regulating the bus current Id to the current reference Id∗. The gate signal GTB of the controllable switch TB is obtained from the comparison of the controller output vcont and the fixed-frequency sawtooth signal vtri with a fixed amplitude
ˆVtri.
The duty ratio dB of the controllable switch TB can be expressed as dB= vcont/ ˆVtri and thus, the average diode voltagekVo
ilTs within the switching period Tsbecomes
kVoilTs =
vcont
ˆVtri Vin (5)
From Fig. 4, the voltage drop Vdrop across the inductance (LB+ 2Ls) and the resistance 2Rs is the difference between the diode voltage Vo
i and the equivalent voltage es. Thus,
the transfer function between the voltage drop Vdrop and the yielded bus current Idcan be expressed as
Id(s) Vdrop(s)=
1
(LB+ 2Ls)s+ 2Rs (6) From (6), the equivalent block diagram of Fig. 4 can be plotted in Fig. 5. The current controller Gc(s) is the proportional and integral controller and it can be expressed as
Gc(s)=skcP+ kcI
s (7)
To design the controller parameters, the ratio of the proportional gain kcPand the integral gain kcIis set to be
kcP kcI = ˆVtri Vin LB+ 2Ls 2Rs (8)
Then, the transfer function between the yielded bus current Id and the bus current command Id∗ can be obtained by
Id(s) Id∗(s)=
kcI/2Rs
s+ k cI/2Rs (9)
Equation (9) behaves like a low-pass filter with the cut-off frequency fcc = kcI/(4πRs) in hertz (Hz). Since the frequency of the equivalent voltage es is six times the electrical frequency in Hz corresponding to the BDCM speed ωr in revolutions per minutes (rpms). The cut-off frequency fcc in Hz should be carefully selected between one tenth the PWM frequency ( ftri/10) and six times the electrical frequency of the maximum BDCM speed
6×P 2× vr, MAX 60 = Pvr, MAX 20 ≤ fcc≤ ftri 10 (10)
where ωr,MAX is the maximum operating speed of BDCM in rpm.
2.3 Terminal voltages of CSI-fed BDCM
When both T1and T2are conducting (i.e. 0≤ θe < π/3), the bus current Idflows through T1and T2. It shows that both a-and c-phase windings can be seen as the excited phases a-and the b-phase winding is seen as the floating phase. The equivalent circuit with turning on and turning off the switch TB are plotted in Figs. 6a and b, respectively. With
Fig. 4 Buck-type CSI with equivalent circuit of BDCM Fig. 3 Illustrated waveforms for ideal CSI-fed BDCM
consideration of the conducting state of the switch TB, the terminal voltages va and vb may be expressed as the following two equations, respectively
va = GTBVin− ea+ ec− 2IdRs 2Ls LB+ 2Ls+ ea− ec + 2IdRs 0≤ue≤ p 3 (11) vb = G TBVin− ea+ ec− 2IdRs Ls LB+ 2Ls+ eb− ec + IdRs 0≤ue≤ p 3 (12)
where the switching signal GTBis GTB= 1,
0,
when vcont. vtriand the switch TBturns on when vcont, vtriand the switch TBturns off
(13)
By neglecting the voltage drop across the equivalent resistance Rs, the terminal voltages in (11) and (12) can be simplified to be va≃ GTBVin 2Ls LB+ 2Ls+ ea LB LB+ 2Ls− ec LB LB+ 2Ls 0≤ue≤p 3 (14) vb≃ GTBVin Ls LB+ 2Ls − ea Ls LB+ 2Ls + eb− ec LB+ Ls LB+ 2Ls 0≤ue≤p 3 (15)
From (14) and (15), it is clear that the switching signal GTB contributes to the voltage ripples in the terminal voltages. From (14), the voltage ripple 2LsVin/(LB + 2Ls) in the terminal voltage va is double the ripple LsVin/(LB + 2Ls) in the terminal voltage vb. The illustrated waveforms for
buck-type CSI-fed BDCM are plotted in Fig. 7. The ripple in the bus voltage Vbus is fixed and equal to 2LsVin/(LB+ 2Ls).
Fig. 6 Circuit and the currentflowing path
a With turning on TB b With turning off TB
3 Position sensorless control for buck-type CSI-fed BDCM
The proposed position sensorless control for buck-type CSI-fed BDCM is plotted in Fig.8, where a buck converter is connected in front of the common CSI inverter. There are seven switches T1∼ T6 and TB in the buck-type CSI. The gate signal of switch TB is generated from the comparison of the current controller output vcontand the sawtooth signal vtri. The other gate signals are obtained from three commutation signals Sab, Sbcand Sca. Their generating rules are as follows GT 1 = Sab· Sca (16) GT 2 = Sbc· Sca (17) GT 3= Sbc· Sab (18) GT 4 = Sca· Sab (19) GT 5 = Sca· Sbc (20) GT 6= Sab· Sbc (21)
The low-passfilter circuits composed of the capacitance C2and the resistances R1are used to sense the three terminal voltages and to generate threefiltered signals va′, vb′ and vc′. Then, the three signals Sab, Sbcand Scaare obtained by comparing two of the threefiltered signals va′, vb′ and vc′.
In order to attenuate the PWM voltage ripple across the terminal voltages, the cut-frequency fLP = (R1+ R2)/ (CR1R2) of the low-pass filter is selected between the
switching frequency ftri in Hz and ten times the electrical frequency corresponding to the maximum BDCM speed ωr, MAX. ftri. fLP≥ 10 ×P 2× vr, MAX 60 = Pvr, MAX 12 (22)
Assume that the PWM voltage ripples arefiltered out without introducing phase delay by the low-passfilter. From (14) and (15), thefiltered signals va′ and vb′ can be simplified to be
v′a = R2 R1+ R2 LB LB+ 2Ls ea− ec 0≤ue≤p 3 (23) v′b= R2 R1+ R2 LB LB+ 2Ls eb− ec + R2 R1+ R2 Ls LB+ 2Ls 2eb− ea− ec 0≤ue≤p 3 (24)
By selecting the buck inductance much larger than the winding inductance LB≫ Ls, the second term in (24) can be neglected and the filtered signal v′b of the floating phase can be expressed as v′b≃ R2 R1+ R2 LB LB+ 2Ls eb− ec = Av eb− ec 0≤ue≤p 3 (25)
where Av = R2/(R1 + R2) is the gain factor.
According to the various conducting states, the expressions of the filtered signals are tabulated in Table1, and they are also illustrated in Fig. 7. It is clear that the commutating instants occur at the crossing points of the three filtered
signals. Therefore the signals Sab, Sbcand Scacan be used to generate the switching signals in (16)–(21).
By using the exclusive (XOR) operator, the signal S is generated from three commutation signals Sab, Sbcand Scaby
S= Sab⊗ Sbc⊗ Sca (26)
where⊗ is the common XOR operator. Then, the period tS(in s) between the rising/falling edges of the combined signal S is counted and the BDCM speedωrin rpm is calculated by
vr = 1 ts× 6× 2 P× 60 = 20 P× ts (27)
Three AC capacitors C1 are connected across the BDCM terminals to provide flowing path for the commutation currents. However, the commutation currents flow through the capacitors and may make the terminal voltages either smaller than zero or larger than the bus voltage Vbus. So, each switch is connected with a diode in series to withstand the negative voltage because of the commutation currents.
When both T4and T5are conducting (i.e.π ≤ θe < 4π/3), the bus current Idflowing through T4and T5, and the terminal voltage is zero va = 0 as shown in Fig.9a. Since the bulky inductance is selected LB≫ Ls, the bus current can be seen as a current source in Fig.9. After turning on the switch T6
and turning off the switch T4, the commutation current iCC mayflow through the capacitor and lift the terminal voltage va until the commutation current iCC decays to zero. The steady-state equivalent circuit during turning on both T5and T6(i.e. 4π/3 ≤ θe < 5π/3) is plotted in Fig.9b.
From Fig.3, the induced voltages of phases a and b is closed to each other ea≃ ebat the commutation instantθe = 4π/3, and thus, the Kirchhoff voltage law (KVL) equation for the commutation current path can be expressed as
2LsdiCC(t) dt + 2RsiCC(t)+ 1 C1 iCC(t)dt≃ 0 (28) The commutation current iCC has initial value Id at the beginning of the current commutation, and the initial energy E stored in the inductance Lscan be expressed as
E≃1 2LsI
2
d (29)
Assume quarter energy E is transferred to the capacitance C1, and then dissipated by the resistances. The peak terminal voltage during the commutation instant θe = 4π/3 would be vpk≃ Id Ls 4C1 (30)
Therefore the capacitance C1should be selected as C1. LsI
2 d
4 v limit2 (31) where vlimitis the maximum voltage with consideration of the diode blocking voltage.
When permanent magnet synchronous motor (PMSM) is standstill, the back-EMFs are zero and the proposed sensorless control needs starting strategy [18]. The proposed starting strategy plotted in Fig. 10 is divided into constant current mode (CCM) and constant speed mode (CSM). In CCM, the bus current command is given by Id∗= I1 and the currents commutate with an increasing frequency until the rotor runs at the constant speed ω1 in rpm. After a given time t1, the operating mode changes from CCM to CSM.
Table 1 Filtered signals in various conducting states
v′a v′b v′c
T1, T2 Av(ea− ec) Av(eb− ec) 0 exciting phase floating phase exciting phase T2, T3 Av(ea− ec) Av(eb− ec) 0
floating phase exciting phase exciting phase T3, T4 0 Av(eb− ea) Av(ec− ea)
exciting phase exciting phase floating phase T4, T5 0 Av(eb− ea) Av(ec− ea)
exciting phase floating phase exciting phase T5, T6 Av(ea− eb) 0 Av(ec− eb)
floating phase exciting phase exciting phase T6, T1 Av(ea− eb) 0 Av(ec− eb)
exciting phase exciting phase floating phase
Fig. 9 Steady-state currentflowing path
a When both T4and T5are conducting b When both T5and T6are conducting
In CSM, the bus current command Id∗ decreases linearly, but the currents commutate with fixed frequency. At the same time, the rotor speed ωris calculated and checked if the difference between the speeds ω1 and ωr is near zero. Once the speed difference |ω1− ωr| is smaller than 30 rpm, the operation mode will change from CSM to the sensorless run mode (SRM).
In SRM, the currents commutate according to the commutation signals Sab, Sbc and Sca, and the bus current command Id∗ is obtained from the speed controller.
4 Simulation results
In this section, some simulation results are provided and the simulated parameters are shown in Table 2. A eight-pole BDCM with the resistance Rs = 0.3Ω and the inductance Ls = 1.7 mH is used, and its maximum operating speed is ωr,MAX= 2000 rpm (i.e. 133.33 Hz in electrical frequency). According to (22) and the parameters in Table2, the cut-off Fig. 10 Operating modes during start-up
Table 2 Simulated parameters of buck-type CSI and BDCM BDCM stator resistance/phase Rs= 0.3Ω BDCM stator inductance/phase Ls= 1.7 mH
BDCM pole number P = 8
BDCM maximum speed ωr,MAX= 2000 rpm
BDCM voltage gain KE= 75 V/krpm
buck input voltage Vin= 300 V
buck inductor LB= 20 mH
buck switching frequency ftri= 10 kHz CSI output capacitor C1= 0.033μF
frequency fLPmust be within the range
1.333 kHz ≤ fLP =R1+ R2
R1R2C2 ≤ 10 kHz (32)
After choosing the gain factor Av= 0.07 in (25) and the cut-off frequency fLP= 3 kHz from (32), the parameters for the terminal voltage sensing circuit are obtained by R1 = 130 kΩ, R2= 7.5 kΩ and C2 = 0.047μF.
Since the blocking voltage of the diode is 600 V and the maximum amplitude of the square-wave current is 5 A, the CSI AC capacitor is selected with C1 = 0.033μF from (31). As shown in Fig.8, two identical BDCMs are used and their shafts are coupled together to become a motor-generator set. One BDCM is connected to CSI, and the other is connected to the Y-connected resistors Rg. Therefore both BDCMs have the same motor speed ωr= 2000 rpm and the same BDCM induced voltages ea, eband ec.
The waveforms during the starting process are plotted in Fig. 11 and the used parameters are I1= 1 A and ω1= 400 rpm. BDCM is successfully changed from CCM and CSM to SRM. The simulated waveforms of the speed command v∗r = 2000 rpm with load resistors Rg= 100Ω and Rg= 33.3Ω are plotted in Figs.12a and b, respectively. The buck current Idis well regulated and the yielded motor current iais in phase with the induced voltage ea. Owing to CSI, the rising time and the falling time of the square-wave currents are relatively small. But the peak of the bus voltage Vbusincreases with the yielded bus current Id.
From Fig. 12, the terminal voltage vamay change rapidly because of the commutating current iCCillustrated in Fig.9. In Fig.12b, the peak of the terminal voltage vamay be 400
Fig. 13 Waveforms during the starting process
Fig. 14 Experimental results for the proposed position sensorless control
aωr= 500 rpm and Rg= 100Ω bωr= 500 rpm and Rg= 33.3Ω Fig. 12 Simulated results for the proposed position sensorless control
aωr= 2000 rpm and Rg= 100Ω bωr= 2000 rpm and Rg= 33.3Ω.
and −100 V. It means that the switch voltage may be negative. Since the controllable semiconductor devices are not able to withstand the negative voltage, the diodes are connected to the switches in series to provide the bipolar withstanding ability.
5 Experimental results
The proposed position sensorless control is implemented in a FPGA-based system. The nominal parameters are the same as those in Table2. Owing to no A/D and no D/A function in commercial FPGA XC3S200 chip, an external A/D converter is used to sense the current input and some D/A converters are used to show the control variables in the scope.
The waveforms during the starting process are plotted in Fig. 13. After 1.5 s, BDCM successfully enters into SRM. The experimental waveforms of the speed command v∗
r = 500 rpm with load resistors Rg = 100Ω and 33.3 Ω are plotted in Figs. 14a and b, respectively. BDCM runs stably at 500 rpm with various loads.
From (14) and (15), the ripple of the terminal voltage during the exciting and the floating phases can be calculated to be 43.6 and 21.8 V, respectively, which meets the observations from the experimental waveforms. From Fig.14b, the peak voltage appearing in the terminal voltage va are near 200 and −100 V, which are smaller than the diode blocking voltage 600 V. Thus, the voltage pulse would not damage to the switch.
The experimental waveforms of the speed command v∗
r = 2000 rpm with load resistors Rg= 100Ω and 33.3 Ω are plotted in Figs. 15a and b, respectively, which shows that the proposed position sensorless control for BDCM runs stably at 2000 rpm with various loads.
From Fig. 15b, the peak voltage appears in the terminal voltage va are near 600 and −400 V, which are smaller than the diode blocking voltage 600 V. Thus, the voltage pulse would not damage to the CSI circuit.
From Figs.14 and15, the voltage ripples in the terminal voltage are fixed, which also demonstrates the derived (14) and (15). In (14) and (15), the profile of the voltage ripples because of the switching signal GTB is fixed regardless of the BDCM speed ωr.
6 Conclusions
In this paper, the behaviour of the buck-type CSI has been studied and the design rules for the buck inductor and the output AC capacitors have been provided. Additionally, the position sensorless control method with square-wave currents for buck-type CSI-fed BDCM has been proposed. The provided simulation and experimental results show that the proposed position sensorless control works stably.
7 References
1 Lelkes, A., Krotsch, J., De Doncker, R.W.:‘Low-noise external rotor BLDC motor for fan applications’. Proc. IAS, 2002, pp. 2036–2042 2 Wang, C.M., Wang, S.J., Lin, S.K., Lin, H.Y.:‘A novel twelve-step
sensorless drive scheme for a brushless DC motor’, IEEE Trans. Magn., 2007,43, (6), pp. 2555–2557
3 Chen, H.C., Chang, Y.C., Huang, C.K.:‘Practical sensorless control for inverter-fed BDCM compressors’, IET Electr. Power Appl., 2007, 1, (1), pp. 127–132
4 Lee, K.W., Kim, D.K., Kim, B.T., Kwon, B.I.:‘A novel starting method of the surface permanent-magnet BLDC motors without position sensor for reciprocating compressor’, IEEE Trans. Ind. Appl., 2008, 44, (1), pp. 85–92
5 Shao, J.:‘An improved microcontroller-based sensorless brushless DC (BLDC) motor drive for automotive applications’, IEEE Trans. Ind. Appl., 2006,42, (5), pp. 1216–1221
6 Wu, Z., Su, G.-J.:‘High-performance permanent magnet machine drive for electric vehicle applications using a current source inverter’. Annual Conf. IEEE Industrial Electronics Society (IECON), November 2008, p. 2812
7 Tang, L., Su, G.J.: ‘Boost mode test of a current-source-inverter-fed permanent magnet synchronous motor drive for automotive applications’, Control Model. Power Electron. (COMPEL), 2010, pp. 1–8
8 Park, J.W., Hwang, S.H., Kim, J.M.:‘Sensorless control of brushless DC motors with torque constant estimation for home applications’, IEEE Trans. Ind. Appl., 2012,48, (2), pp. 677–684
9 Han, Q., Samoylenko, N., Jatskevich, J.:‘Average-value modeling of brushless DC motors with 120° voltage source inverter’, IEEE Trans. Energy Convers., 2008,23, (2), pp. 423–432
Fig. 15 Experimental results for the proposed position sensorless control
aωr= 2000 rpm and Rg= 100Ω bωr= 2000 rpm and Rg= 33.3Ω
10 Singh, B., Singh, S.:‘Single-phase power factor controller topologies for permanent magnet brushless DC motor drives’, IET Power Electron., 2010,3, (2), pp. 147–175
11 Lin, C.T., Huang, C.W., Liu, C.W.:‘Position sensorless control for four-switch three-phase brushless DC motor drives’, IEEE Trans. Power Electron., 2008,23, (1), pp. 438–444
12 Gu, B., Nam, K.:‘A DC-link capacitor minimization method through direct capacitor current control’, IEEE Trans. Ind. Appl., 2006, 42, (2), pp. 573–581 13 Chen, H.C., Liaw, C.M.:‘Current-mode control of sensorless BDCM drive with intelligent commutation tuning’, IEEE Trans. Power Electron., 2002,17, (5), pp. 747–756
14 Su, G.J., Mckeever, W.:‘Low-cost sensorless control of brushless DC motors with improved speed range’, IEEE Trans. Power Electron., 2004,19, (2), pp. 296–302
15 Kim, D.K., Lee, K.W., Kwon, B.I.: ‘Commutation torque ripple reduction in a position sensorless brushless DC motor drive’, IEEE Trans. Power Electron., 2006,21, (6), pp. 1762–1768
16 Lai, Y.S., Lin, Y.K.:‘Novel back-EMF detection technique of brushless DC motor drives for wide range control without using current and position sensors’, IEEE Trans. Power Electron., 2008, 23, (2), pp. 934–940
17 Damodharan, P., Vasudevan, K.:‘Sensorless brushless DC motor drive based on the zero-crossing detection of back electromotive force (EMF) from the line voltage difference’, IEEE Trans. Energy Convers., 2010, 25, (3), pp. 661–668
18 Wang, Z., Lu, K., Blaabjerg, F.:‘A simple start-up strategy based on current regulation for back-EMF-based sensorless control of PMSM’, IEEE Trans. Power Electron., 2012,27, (8), pp. 3817–3825
