Chapter 5 Summary and Future Work
5.2 Future Work
The flip-chip technology makes it possible to realize the system building on the package. We can integrate more RF modules to complete a single IC with high integration.
Bulk-driven mixer uses only one transistor to down-convert the RF signal. As shown in Fig. 5.1, it is possible to down-convert the RF signal twice in one stage by stacking the bulk-driven mixer with an additional switching pair to increase the flexibility in the receiver design.
LO1+
LO1-RF R
LV
DDR
LIF+
IF-M
2M
1RF
LO2+
LO2-Fig. 5.1. Mixer topology for twice downconversion.
References
[1] Bert Gyselinckx, Chris Van Hoof, Julien Ryckaert, Refet Firat Yazicioglu, Paolo Fiorini, Vladimir Leonov, “Human++: Autonomous Wireless Sensors for Body Area Networks,” IEEE Custom Integrated Circuits Conference, pp. 13-19, Sept.
2005.
[2] S. Chakraborty, K. Lim, A Sutono, E. Chen, S. Yoon, A. Obatoyinbo, S. -W.
Yoon, M. Maeng, M. F. Davis, S. Pinel, and J. Laskar, “A 2.4-GHz Radio Front End in RF System-on-Package Technology”, IEEE Microwave Magazine, pp.
94-104, June 2002.
[3] HongMo Wang, “A 1-V Multigigahertz RF Mixer Core in 0.5-um CMOS,” IEEE
J. Solid-State Circuits, vol.33, no.12, Dec. 1998.
[4] G. Kathiresan and C. Toumazou, “A low voltage bulk driven downconversion mixer core,” IEEE International Symposium on Circuits and Systems, vol. 2, pp.
598-601, June 1999.
[5] C. Kienmaer, M. Tiebout, W. Simburger, A. L. Scholtz, “A low-power low-voltage NMOS bulk-mixer with 20 GHz bandwidth in 90 nm CMOS,” IEEE
International Symposium on Circuits and Systems, vol. 4, pp. 385-388, May 2004.
[6] M. J. Deen, R. Murji, A. Fakhr, N. Jafferali and W. L. Ngan, “Low-power CMOS integrated circuits for radio frequency applications,” IEE Proc.-Circuits
Devices and System, Vol. 152, No. 5, pp. 509-522, Oct. 2005.
[7] W. Heinrich, “The Flip-Chip Approach for Millimeter-Wave Packaging,” IEEE
Microwave Magazine, pp. 36-45, Sep. 2005.
[8] K. Onodera, T. Ishii, S. Aoyama, S. Sugitani, and M. Tokumitsu, “Novel Flip-Chip Bonding Technology for W-Band Interconnections Using Alternate Lead-Free Solder Bumps,” IEEE Microw. Wireless Comp. Lett., vol. 12, no. 10, pp. 372-374, Oct. 2002.
[9] A. Tessmann, M. Riessle, S. Kudszus, and H. Massler, “A Flip-Chip Packaged Coplanar 94 GHz Amplifier Module With Efficient Suppression of Parasitic Substrate Effects,” IEEE Microw. Wireless Comp. Lett., vol. 14, no. 4, pp.
145-147, April 2004.
[10] A. Jentzsch and W. Heinrich, “Theory and measurement of flip-chip interconnects for frequencies up to 100GHz,” IEEE Trans. Microw. Theory Tech., vol. 49, no. 5, pp. 871-878, May 2001.
[11] C.-L. Wang and R.-B. Wu, “A Locally Matching Technique for Broadband Flip-chip Transition Design,” 2002 IEEE MTT-S Int. Microwave Symp. Dig., vol.
3, pp. 1397-1400, June 2002.
[12] David M. Pozar,” Microwave Engineering,” 3rd Edition, Wiley.
[13] Angilent E8364B PNA Manual.
[14] Alain M. Mangan, Sorin P. Voinigescu, Ming-Ta Yang, and Mihai Tazlaunau,“ De-Embedding Transmission Line Measurements for Accurate Modeling of IC Designs,” IEEE Transactions on Electron Devices, vol. 53, no.
12, Feb. 2006.
[15] Y. C. Chen, C.-H. Li, J. K. Huang, C.-N. Kuo, and Y. T. Cheng, "Low Power 3~8 GHz UWB Tunable LNA Desin Using SiP Technology," IEEE Int.
Conference on Electronics, Circuits and Systems Proc., Nice, FR, Dec. 2006.
[16] Albert Jerng, and Charles G. Sodini, “ The Impact of Device Type and Sizing on Phase Noise Mechanisms,” IEEE J.Solid-State Circuits, vol. 40, no. 2, p.
360-369, Feb. 2005.
[17] Piet Wambach and Willy Sansen, “Distortion Analysis of Analog Integrated Circuits,” 1998. Name Stand. Abbrev., in press.
[18] Thomas H. Lee, “The Design of CMOS Radio-Frequency Integrated Circuits,”
2nd Edition, Cambridge.
Appendix A Fundamental of Phase Noise
Phase noise is one of the most important parameters in the VCO. The output of an ideal oscillator in the frequency domain consists of an impulse at the oscillation frequency wo. However, in practical oscillator, the output spectrum has power around harmonics of wo and, as a consequence of the fluctuations represented by
φ
(t) andA(t) in equation (A-1), the spectrum is not a impulse, but a shape of skirt as shown in Fig. A.1(b). These sidebands are generally referred to as phase noise sidebands.
)]
( [
)]
( 1 [ )
(t V A t f w t t
Vout = o⋅ + ⋅ o +φ (A-1) For a radio receiver, suppose there is a strong interference in an adjacent channel, as shown in Fig. A.1(a). If the LO has large phase noise, some down-conversion of
(a) (b)
(c)
Fig. A.1. Influence of strong interference on dynamic range of receiver, (a) a strong interference adjacent to desired signal (b) practical spectrum in VCO output (c) impact of interference on the desired signal.
the interfering signal into the same intermediate frequency (IF) as that of the desired signal will occur as shown in Fig. A.1(c). The resulting interference will significantly degrade the dynamic range of the receiver. Therefore, improving the phase noise clearly improve the signal-to-noise ratio (SNR) of the desired signal.
The spectrum of phase noise is shown in Fig. A.2. The phase noise in the 1/f3 region and the 1/f2 region are dominated by the flicker noise and the thermal noise, respectively. The most widely used phase noise model is Leeson’s model based on a linear time-invariant (LTI) approach. It predicts the following behavior for the phase noise, L(Δw) [18]:
| )]
1 | ( ] 2 )
( 1 2 [
log[
10 ) (
/ 3 2 1
w w w
Q w P
w FkT
L f
L o
s ⋅ + ∆
+ ∆
⋅
⋅
=
∆ (A-2)
where F is an fitting parameter, often called a device excess noise number, and QL is the loaded Q of the tank. The drawback of Leeson’s model is that it cannot predict the
3
1 f
2
1 f )
( w L ∆
) log( w∆
/ 3
w1 f
Fig. A.2. The spectrum of the phase noise. The 1/f3 region is mainly contributed by the flicker noise and the 1/f2 region is dominated by thermal noise.
measurement.
Hajimiri and Thomas Lee propose the time-variant model to predict the response of the phase noise [18]. Since each input noise source generally affects both amplitude and phase, a pair of equivalent systems, one each for amplitude and phase, can be defined. The input of each system in Fig. A.3 is a perturbation current (or voltage) and the outputs are the excess phase
φ
(t), and amplitude, A(t), as defined by (A-1). In response to a current impulse, like a noise, the excess amplitude undergoes some transient behavior but finally converges to zero. However, fluctuations in the excess phase are not quenched by any restoring mechanism and therefore persist indefinitely so that the unit phase impulse response can be easily written as) , ) ( ) (
, (
max
τ τ
φ τ u t
q t w
h Γ o
= (A-3) where
max
max C V
q = ⋅ , u(t) is a unit step function, and Γ(woτ) is an impulse sensitivity function (ISF). Because the injected charge is linear to excess phase if the amount of injected chare is small, the output excess phase
φ
(t) can be calculated) , ( τ
φ t h )
(t i
) (t
i φ(t)
) φ(t
t
τ
t0 0
) , (t τ hA )
(t i
) (t
i A(t)
) (t A
t
τ
t0 0
Fig. A.3. An equivalent systems for excess amplitude and phase response in phase noise analysis.
using the superposition integral:
Since ISF is periodic, it can be expanded in a Fourier series
∑ + Consider a white input noise current with power spectral density i f
n2/∆ .
φ
(t) canbe known from (A-4) and note by (A-6) that
φ
(t) makes a convolution with fundamental tone in the frequency domain to produce the phase noise sideband adjacent to the signal at wo. Hence total single-sideband phase noise spectral density due to one noise source at an offset frequency Δw is given by (A-7).))
Equation (A-7) implys both upward and down ward frequency translations of noise into the noise near the carrier, as illustrated in Fig. A.4. This figure summarizes what the (A-7) tell us: Components of noise near integer multiples of the carrier frequency all fold into noise near the carrier itself. Noise near DC is upconverted, with relative weight given by coefficient co, so 1/f noise ultimately becomes 1/f3 noise near the carrier; noise near the carrier stays there, weighted by c1.; and white noise near higher
harmonic of carrier undergoes downconversion into the 1/f2 region.
And according to Parseval’s theorem,
2 2 re-written as:
)
So minimizing theΓrms will minimize the phase noise.
For the 1/f3 region, the flicker noise can be written as
w Hence the phase noise in the 1/f3 region can be obtained:
∆ w ∆ w ∆ w
Fig. A.4. Evolution of circuit noise into phase noise.
)
The equation (A-11) tells us that VCO with a symmetric waveform can feature in low phase noise at low offset frequency due to less flicker noise contribution because of small c0. So for a VCO design, we shall carefully design the circuit and make a symmetric layout to have a symmetric waveform.
In practical oscillators, the statistical properties of some of the random noise sources may change with time in a periodic manner. These sources are referred to as cyclostationary. A white cyclostationary noise current in(t) can always be decompose as where ino(t) is a white stationary process and α(wot) is a deterministic periodic function describing the noise amplitude modulation and therefore is referred to as the
noise modulation function (NMF). So
φ
(t) may be re-written as:τ τ
As can be seen, cyclostationary noise can be treated as a stationary noise applied to a system with a new ISF given by
) where α(x) can be derived easily from device noise characteristic and the noiseless
steady-state waveform.
Vita
李俊興 Chun-Hsing Li Birthday: 1983/09/12
Birthplace: Ilan Country, Taiwan Education:
2001/09 ~ 2005/06
B.S. Degree in Department of Electrophysics, National Chiao Tung University
2005/09 ~ 2007/07
M.S. Degree in Department of Electronics Engineering & Institute of Electronics, National Chiao Tung University
Publication:
1. Y. C. Chen, C.-H. Li, J. K. Huang, C.-N. Kuo, and Y. T. Cheng, "Low Power 3~8 GHz UWB Tunable LNA Desin Using SiP Technology," the 2006 IEEE Int. Conference on Electronics, Circuits and Systems Proc., Nice, FR, Dec.
2006.
2. Chun-Hsing Li, ChangTsung Fu, Tzu-Yuan Chao, Chien-Nan Kuo, Y. T.
Cheng, and D.-C. Chang, “Broadband Flip-Chip Interconnects for
Millimeter-Wave Si-Carrier System-on-Package,” 2007 IEEE International Microwave Symposium, Honolulu, Hawaii, June, 2007, pp. 1645-1648.