Overall System

The proposed complete feedback scheme applied to an isolated switch-mode power

supply is shown in Fig. 3.5, and two examples of a flyback and a forward converters are

presented in Fig. 3.6. In these implementations, a secondary-side integrated circuit is

substituted for the traditional shunt regulator. It pulls down ILED according to the

difference between VOF and the built-in reference voltage. The higher VOUT is, the

smaller ILED will be conducted. As its operation is reversed compared to the traditional

shunt regulator which will draw a larger ILED with a larger VOUT, we call it the

reverse-type shunt regulator (RTSR). Note that the supply current IQ will not flow

through MP and is not contained in ILED. On the primary side, the only difference in the

controller is that an inverting amplifier is presented before the PWM modulator. Other

off-chip components, including RLED, CP, RC, and CC, are added for implementing a

frequency compensator, which will be described later.

Fig. 3.5. The proposed complete low-standby-power feedback network.

The proposed feedback network basically performs the same function as the

conventional one does, but the key point is that the phase of the intermediate error

signal for optical coupling is reversed. With this proposed feedback scheme adopted, a

higher VFB, which gives a lower VRFB, will correspond to a higher VOUT, and therefore

losses due to ILED and IFB will automatically reach minimum values under the no-load

condition. Concerns may be aroused that whether or not the additional power losses

caused by the inverting amplifier and IQ surpass the saved power under the no-load

condition. As previously mentioned, the current consumption of the inverting amplifier

can be designed to be only a few tens of microamperes. Also, the supply current of the

shunt regulator is not contained in ILED, and thus the minimum values of ILED and IFB for

operating are essentially not limited and can be designed to be very small. With these

two features, the power loss of the feedback network under the no-load condition can be

(a)

(b)

Fig. 3.6. Proposed feedback network adopted in (a) flyback and (b) forward topologies.

minimized. In the following sections, we will present the power loss analysis as well as

the control loop compensation analysis.

3.4 Power Loss Analysis

As what we have done for the conventional feedback network in Section 2.2.4, we

also want to formulate the power loss that is associated with the proposed feedback

circuit. First, we can recognize from Fig. 3.5 that there are five current branches. The

first one is the current flowing through the voltage divider. The second one is IQ, which

is consumed by the error amplifier and the voltage reference in the shunt regulator. The

third one is ILED, which is conducted by MP and the optocoupler. The fourth and fifth

ones are respectively IFB and the current dissipation of the inverting amplifier. Since

there is only a slight power consumed by the inverting amplifier, we directly denote it as

PIV for convenience. Thus, if we first make an assumption of ideal energy conversion,

the power loss (PL,pro.) of the entire feedback network can be estimated by

IV

From observing (3.3), we see that the second term is the power loss caused by currents

flowing through the optocoupler on the primary and the secondary sides. For a

well-designed power converter, this part of loss will vary with VFB, which is determined

by the present load condition. Equation (3.3) is a simplified general estimation for any

transformer-isolated converter adopting the proposed feedback network. If we solely

consider a flyback converter, as shown in Fig. 3.6(a), equation (3.3) can be further

When operating under the no-load condition, converters generally adopt the burst

mode control to regulate their outputs [39], [40]. As previously mentioned in Section

2.2.4, for a conventional PWM controller, it will start using the burst mode to control

the system when VFB is lower than a threshold voltage [57]. This mechanism is

inappropriate for the proposed feedback topology in which, as shown in Fig. 3.1, VFB

increases with the decrease of the output power. Under this circumstance, the burst

mode threshold voltage VBU should be set close to VLO, and the burst mode control

should be activated when VFB is larger than VBU. Fig. 3.7 illustrates simulated

waveforms of VFB and the gate-driving signal VG in the burst mode under the no-load

condition. In this case, VBU is set 4.5 V while VLO is 5 V. The driving signal VG is

Fig. 3.7. Simulated burst-mode waveforms with proposed feedback topology adopted.

minimum values of ILED and IFB are not limited in the proposed feedback scheme) and

therefore the loop response is very slow under the no-load condition, VFB can be

designed deliberately to touch and stay at VLO in the period between the bursts. In this

way, the optocoupler actually conducts zero currents on both sides in that duration.

Because the only current dissipation at the output node comes from the resistor divider,

the switching-ceased period is relatively long. Thus, under the no-load condition, (3.3)

and (3.4) can be respectively approximated as

IV current of the inverting amplifier is 25 μA. The second and the third terms in (3.8)

together add up to merely 3.4 mW under the condition that VOUT = 12 V, VCC = 10 V,

and VD1-D2 = 0.5 V, making PL,pro. mainly dominated by the power consumption of the

resistor divider only. Recall that, in Section 2.2.4, the conventional feedback network in

a typical flyback converter having the same conditions consumes a power of 23 mW

excluding the part of the resistor divider. Comparing the power losses of the

conventional and the proposed feedback circuits, we can find that a power of 19.6 mW

can be saved by simply applying the proposed feedback scheme. It should be noted that

here we do not consider switching losses for simplicity. The estimated saved power is

thus underestimated, which will be discussed more in Chapter 4.