Primary Sensing Technique

The primary sensing [51]-[54] means the output information is not feedback

through the explicit signal path. Instead, it tries to extract the output voltage from the

information already existing on the primary side. Hence, the entire feedback network

can be removed. Not only the cost can be largely saved from this technique, but also the

losses due to current branches in the conventional feedback network disappear. Fig. 2.11

shows a simplified primary-side-control flyback converter, where there is no any direct

signal path for feedback except for the flux coupling through the flyback transformer.

To show how the voltage extraction technique takes effect, the operating

waveforms of the converter in Fig. 2.11 are illustrated in Fig. 2.12. To obtain the VOUT

information, we first recognize that the ratio of the transformer winding voltages is

proportional to that of their turn numbers. That is,

A

the ground. The flyback transformer is charged with VPW equal to VIN, making the

primary-side inductor current ILP continuously climb up. Due to the flux coupling, the

auxiliary winding (tertiary winding) then reflects a voltage of −(NA/NP)VIN. Next, in t2

when the power MOS is turned OFF, the transformer starts discharging through the

secondary winding. The secondary winding sees a voltage VSW equal to VOUT plus the

diode voltage VD1, so the secondary-side inductor current ILS declines gradually with a

slope of −V /L . In the meantime, V reflects a voltage of

Fig. 2.11. Primary-side control flyback converter.

Fig. 2.12. Operating waveforms of primary-side-control flyback converter.

(

)

S A

AUX V V

N

V = N + . (2.23)

VPW also reflects a voltage, and VDRAIN can be expressed as

(

)

S P IN

DRAIN V V

N V N

V = + + . (2.24)

The above two equations thus inspire us that they contain the output information in this

period of time. Although theoretically it is possible to obtain VOUT from VDRAIN [51], the

quite high voltage there which would probably cause troubles and inconvenience makes

it a worse choice. Therefore, VAUX is mostly chosen for extracting VOUT for feedback

and the controller IC generally samples the divided VAUX. In t3, the energy in the

transformer is empty. The parasitic capacitance at the drain of the power MOS and the

primary-side magnetizing inductance LP begin to resonate with each other, leading to

decayed sinusoidal voltage waveforms of the transformer windings. Now we focus on

the t2 period. Since ILS is getting smaller and so is VD1, VAUX in this time is actually not

constant. It then becomes a problem that where is exactly the best point to sample VAUX

for extracting VOUT. Some products samples VAUX at the point VP1 where ILS is just

discharged to zero. At that point, since almost zero current flows through the diode, VD1

is nearly zero and VAUX can be approximated as (NA/NS)VOUT which is proportional to

the output voltage. However, the slope of VAUX around there changes so drastically,

making it very difficult to acquire the voltage accurately. Some researches [52] try to

sample VAUX at a point prior to VP1 (e.g., VP2 in Fig. 2.12). If VAUX is captured at a fixed

ILS in each cycle, VD1 is fixed and we can still take out VOUT from VAUX with high

accuracy as well. Compared to VP1, there is no abruptly voltage change around VP2.

However, how to ensure a fixed ILS in each sampling time imposes another difficulty to

the circuit designers. No matter which sampling point is chosen, both of them tell us

that sampling VOUT from VAUX would basically cause a relatively poor regulation in

comparison with directly using the optical-coupling feedback network.

Although this existing primary-side-control power conversion solution which

indirectly senses the output information through the auxiliary (tertiary) winding and

hence obviates the need for isolated feedback network can achieve low cost and low

standby power consumption in nature, it still has many limitations. First, this technique

is only applicable to flyback topology. Other topologies in essence do not have the

property that the output voltage will be reflected in any of the winding voltages on the

primary side. Second, as described just now, the auxiliary winding voltage contains not

only the output voltage but also the voltage of the secondary-side rectifier diode,

making it very hard to precisely extract the output information. Third, primary-side-

control flyback converter is mostly limited to the discontinuous-conduction-mode

operation. Because in the continuous-conduction mode, ILS is not going to drop down to

zero and thus it is even hard to predict ILS at the sampling point, a primary-side-control

converter operating in this mode would encounter an even worse output variation

problem. Fourth, there is generally a dummy load RDUM required at the output, as shown

in Fig. 2.11, leading to an additional power dissipation. When there is no output load

applied, switching may be required only after a long period of time to maintain the

output voltage. However, this fact also makes the controller blind of the output voltage

in that period of time. To ensure a prompt response to a suddenly applied heavy load,

primary-side controllers must possess a maximum OFF-time limit to keep updating the

output condition and a dummy load is thus necessary to prevent the over voltage at the

output node.

From the above, it is therefore that such converters generally suffer from poor

output regulation and also have a very limited application scope. As a result, using an

optocoupler to feedback information is still necessary in most applications and,

consequently, lowering down the power dissipation of isolated feedback network under

the no-load condition is potentially of great interest. In the following sections, two

circuit techniques are introduced to reduce the no-load power loss of the conventional

feedback network.