國 立 交 通 大 學
電子工程學系 電子研究所碩士班
碩
士
論
文
IEEE 802.16m 空間多工模式閉迴路
多輸出入傳輸之預編碼與等化研究
A Study on Precoding and Equalization for the
Spatial Multiplexing Mode of IEEE 802.16m
Closed-Loop MIMO
研 究 生：柯俊言
指導教授：林大衛 博士
IEEE 802.16m 空間多工模式閉迴路
多輸出入傳輸之預編碼與等化研究
A Study on Precoding and Equalization for the
Spatial Multiplexing Mode of IEEE 802.16m
Closed-Loop MIMO
研 究 生: 柯俊言
Student: Chun-Yen Ko
指導教授: 林大衛 博士 Advisor: Dr. David W. Lin
國 立 交 通 大 學
電子工程學系 電子研究所碩士班
碩士論文
A Thesis
Submitted to Department of Electronics Engineering & Institute of Electronics College of Electrical and Computer Engineering
National Chiao Tung University in Partial Fulfillment of the Requirements
for the Degree of Master of Science
in
Electronics Engineering July 2011
IEEE 802.16m 空間多工模式閉迴路
多輸出入傳輸之預編碼與等化研究
研究生:柯俊言 指導教授:林大衛博士
國立交通大學
電子工程學系 電子研究所碩士班
摘要
多輸出入傳輸技術近來在行動環境中廣受注目且已經應用在許
多數位通訊系統中，因為跟單輸出入傳輸相比之下，容量和可靠度有
顯著的提升。
我們聚焦在 IEEE 802.16m 空間多工模式閉迴路多輸出入傳輸之
預編碼與等化研究。本篇論文採用兩種等化方法。第一種為迫零(ZF)
等化器，他在頻域上為反向濾波器，這是一個最簡單的等化器，他可
以消除符號干擾，但是他會造成雜訊放大。第二種為最小均方差
(MMSE)等化器，最小均方差等化器是使設計估測通道信號及實際信
號的均方誤差為最小，雖然沒辦法完全消除符號干擾，但是不會造成
雜訊放大。
我們依據這兩種等化器，再去設計選擇預編碼的方法。要計算預
編碼的話，完整通道資訊必須要傳給傳輸端。這會產生一個問題，因
為我們的回傳通道的頻寬是有限的。所以我們預先設計好預編碼，再
回傳最合適的預編碼編號。我們提出兩種方法去找尋最適合的預編
碼，第一種是基於最小均方差方法，我們把所有預編碼裡代入運算，
找出最小均方差。第二種是找出使兩根天線最小的訊號雜訊比最大的
預編碼，這個方法必須先計算出每個預編碼天線的訊號雜訊比，我們
在找出最適合的回傳。我們會跟以下兩種方法做比較。第一種為基於
奇異值的方法，首先把通道頻率響應做奇異值分解，其中通道頻率響
應的正交輸入基向量是最佳的迫零等化器，我們把所有可能的預編碼
跟他做旋距離計算，找出最小的預編碼回傳。第二種為最佳預編碼計
算，我們會利用注水原理去求得最佳解。我們先在加成性白色高斯通
道下驗證我們的模擬模型，然後在 IEEE802.16m 標準下於單路徑和
多重路徑通道上模擬。
在本篇論文中，我們首先簡介 IEEE802.16m 的標準機制。接著，
我們依照標準分別各傳輸下介紹兩種等化器搭配四種選擇預編碼方
法並探討其效能。
A Study on Precoding and Equalization for the
Spatial Multiplexing Mode of IEEE 802.16m
Closed-Loop MIMO
Student: Chun-Yen Ko Advisor: Dr. David W. Lin
Department of Electronics Engineering
& Institute of Electronics
National Chiao Tung University
Abstract
MIMO channels arising from the use of multiple antennas at both the
transmitter and at the receiver have recently attracted significant interest
because they provide a significant increase in capacity and reliability over
single-input single-output (SISO) channels under some uncorrelation
conditions.
We focus on Precoding and Equalization for the Spatial Multiplexing
Mode of IEEE 802.16m Closed-Loop MIMO. We present two equalizer
methods. One is Zero-forcing equalizer. It is an inverse filter in frequency
domain. This is the easiest equalizer. It can remove the ISI, but it will
increase the noise. The other method is MMSE equalizer. This method is
designed to minimize the mean square error of the receive signal and the
transmit signal. It can not remove all of the ISI, but it will not increase the
noise.
A problem associated with precoding is that the channel state
information must be known at transmitter. This may be difficult since the
bandwidth of the feedback channel is usually limited. Thus, a
codebook-based limited feedback precoding scheme is generally used.
The main idea is to quantize the precoding matrix and feedback the index
of the optimum precoder. We based on these two equalizers to design the
selection method to select the best precoder. We proposed MMSE-Based
and MaxminSNR-Based method. MMSE-Based method finds the
precoder has the minimum mean square error. MaxminSNR-Based
method finds the precoder that maximizes the minimum SNR of the two
antennas. This method has to calculate each precoder’s antenna SNR.
Then, we select the appropriate one to transmit back. We will compare
with the following two methods. First is SVD-Based method. We take the
singular value decomposition (SVD) of the channel matrix. The right
singular vector of the channel matrix is the best ZF equalizer. We
calculate all the chordal distance of the possible precoder and the best ZF
equalizer. Then, we transmit the precoder. Second is the optimum
precoder computation. We use water-filling method to get the solution.
We verify our simulation model on AWGN channel and then do the
simulation on singlepath and multipath channels for IEEE 802.16m.
In this thesis, we first introduce the standard of the IEEE 802.16m.
Then we describe the precoding and equalization methods we use and
discuss the performance in each transmission condition for IEEE
802.16m.
v
誌謝
本篇論文的完成，首先要特別感謝我的指導教授林大衛博士，在我進入交大 電子所開始，不論是課業或是研究上的困難，老師總能細心的給予適時方向去解 決問題，使我學到了分析以及解決問題的能力。此外老師對於學生的認真以及親 切樂觀的態度更是深遠影響了我，使我在研究所的這幾年得到了學術以外更重要 的智慧。 此外，由衷的感謝通訊電子與訊號處理實驗室所有的成員，包含各位師長、 同學、學長姐以及學弟妹們。特別感謝曲建全學長、林鴻志學長、蕭世璞學長、 洪朝雄學長、蔡彰哲學長、邱頌恩學長、吳思賢學長和盧世榮學長對我在學術上 的不吝指導與建議，謝謝你們幫我解決了許多通訊方面的疑問。感謝 98 級曉盈、 威宇、卓翰、強丹、智凱、兆軒、郁婷、琬瑜、怡茹、頌文、書緯、偵源、凱翔、 復凱及 99 級學弟們等實驗室成員，平日和我一起念書、討論、玩耍，讓我的研 究生涯擁有美好的回憶。期待各位夥伴們畢業後都有不錯的發展。 最後，必須感謝我的家人，我父母親、姐姐及惠婷給予我的支持，使我在外 地讀書時能無後顧之憂，感謝他們的支持，也謝謝所有幫助過我、陪我走過這段 歲月的人。 柯俊言 民國一百年七月 於新竹Contents
1 Introduction 1
1.1 Scope of the Work . . . 1
1.2 Organization and Contribution of this Thesis . . . 3
2 Introduction to IEEE 802.16m OFDMA 5 2.1 Basic OFDMA Symbol Structure and Frame Structure [10] . . . 5
2.1.1 OFDMA Basic Terms . . . 6
2.1.2 Frequency Domain Description . . . 6
2.1.3 Primitive Parameters . . . 7
2.1.4 Derived Parameters . . . 7
2.1.5 Frame Structure . . . 8
2.2 Downlink Transmission in IEEE 802.16m OFDMA[10] . . . 11
2.2.1 Subband Partitioning . . . 12
2.2.2 Miniband Permutation . . . 15
2.2.3 Frequency Partitioning . . . 16
2.3.1 CRU/DRU Allocation . . . 21
2.3.2 Subcarrier Permutation . . . 24
2.3.3 Random Sequence Generation . . . 26
2.4 Test Case Generation . . . 27
2.4.1 Subband Partitioning . . . 28 2.4.2 Miniband Partitioning . . . 30 2.4.3 Miniband Permutation . . . 30 2.4.4 Frequency Partitioning . . . 32 2.4.5 Random Sequence . . . 34 2.4.6 CRU/DRU Allocation . . . 35 2.5 MIMO Midamble [10] . . . 35
2.6 Usage of Downlink Pilots [10] . . . 38
2.7 Downlink control structure [10] . . . 40
2.7.1 Advanced Preamble . . . 40
2.7.2 Primary Advanced Preamble (PA-Preamble) . . . 40
2.7.3 Secondary Advanced Preamble (SA-Preamble) . . . 43
2.8 DL Control Channels [10] . . . 44
2.8.1 Superframe Header . . . 44
2.8.2 Primary Superframe Header . . . 45
2.8.3 Secondary Superframe Header . . . 45
2.8.5 Non-user Specific A-MAP . . . 48
2.8.6 HARQ Feedback A-MAP . . . 48
2.8.7 Power Control A-MAP . . . 49
2.8.8 Assignment A-MAP . . . 49
2.9 Resource Mapping of DL Control Channels [10] . . . 50
2.9.1 Superframe Header . . . 51
2.9.2 Downlink Power Control . . . 57
3 Introduction to IEEE 802.16m MIMO 59 3.1 Downlink MIMO Architecture and Data Processing [10] . . . 59
3.2 MIMO Layer to MIMO Stream Mapping [10] . . . 60
3.2.1 SFBC Encoding . . . 61
3.2.2 Vertical Encoding (VE) . . . 61
3.2.3 Multi-layer Encoding (ME) . . . 61
3.2.4 CDR Encoding . . . 62
3.3 MIMO Stream to Antenna Mapping [10] . . . 62
3.3.1 Non-adaptive Precoding . . . 62
3.3.2 Adaptive Precoding . . . 65
3.4 Downlink MIMO Modes [10] . . . 65
3.5 Transmission Schemes for Data Channels [10] . . . 65
3.5.1 Encoding and Precoding of SU-MIMO . . . 66
3.5.3 Mapping of Data and Pilot Subcarriers . . . 69
3.5.4 Usage of MIMO Modes . . . 70
3.6 Feedback Mechanisms and Operation [10] . . . 70
3.6.1 Open-Loop Region . . . 71
3.6.2 MIMO Feedback Mode Selection . . . 72
3.6.3 MIMO Feedback Modes . . . 72
3.6.4 Downlink Signaling Support of DL-MIMO Modes . . . 74
3.6.5 Quantized MIMO Feedback for Closed-loop Transmit Precoding . . . 78
3.7 Uplink MIMO Transmission Schemes [10] . . . 86
3.7.1 Uplink MIMO Architecture and Data Processing . . . 86
3.7.2 MIMO Layer to MIMO Stream Mapping . . . 87
3.7.3 MIMO Stream to Antenna Mapping . . . 87
3.7.4 Uplink MIMO Transmission Modes . . . 89
4 Equalization and Closed-Loop MIMO Technology 94 4.1 System Model . . . 94
4.1.1 Linear Equalizer Matrix . . . 95
4.2 Precoder Selection Methods . . . 96
4.2.1 MaxminSNR-Based Search Method . . . 96
4.2.2 MMSE-Based Exhaustive Search Method . . . 97
4.2.3 SVD-Based Search Method [14] . . . 98
4.2.5 Matrix Computation Complexity . . . 101
5 Downlink MIMO Simulation for IEEE 802.16m 103 5.1 System Parameters and Channel Model . . . 103
5.2 Simulation Results for IEEE 802.16m . . . 105
5.2.1 Simulation Flow . . . 105
5.2.2 Validation with AWGN Channel . . . 106
5.3 Simulation Results for Single-Path Channels . . . 107
5.4 Simulation Results for Multipath Channels . . . 125
6 Conclusion and Future Work 147 6.1 Conclusion . . . 147
6.2 Future Work . . . 147
7 Using Matlab to Generate MIMO Channel 149 7.1 System Parameters . . . 149 Bibliography 151
List of Figures
2.1 OFDMA parameters (from [10, Table 775]). . . 9 2.2 More OFDMA parameters (from [10, Table 775]). . . 10 2.3 Basic frame structure for 5, 10 and 20 MHz channel bandwidths (Fig.466 in
[10]). . . 10 2.4 Example of downlink physical structure (Fig.485 in [10]). . . 12 2.5 Mapping between SAC and KSB for FFT size 2048 (from [10, Table 783]). . 14
2.6 Mapping between SAC and KSB for FFT size 1024 (from [10, Table 784]). . 14
2.7 Mapping between SAC and KSB for FFT size 512 (from [10, Table 785]). . . 15
2.8 PRU to P RUSB and P RUM B mapping for BW = 10 MHz, and KSB=7. . . . 15
2.9 Mapping from PRUs to P RUSB and P P RUM B for BW = 10 MHz and KSB
= 7. . . 17 2.10 Mapping between DFPC and frequency partitioning for FFT size 2048 (from
[10, Table 786]). . . 18 2.11 Mapping between DFPC and frequency partitioning for FFT size 1024 (from
[10, Table 787]). . . 18 2.12 Mapping between DFPC and frequency partitioning for FFT size 512 (from
2.13 Frequency partitioning for BW = 10 MHz, KSB = 7, FPCT = 4, FPS = 12,
and DFPSC = 2. . . 19
2.14 Mapping between DCASM B,0 and number of miniband based CRUs for F P0 for FFT size 2048 (from [10, Table 789]). . . 22
2.15 Mapping between DCASM B,0 and number of miniband based CRUs for F P0 for FFT size 1024 (from [10, Table 790]). . . 23
2.16 Mapping between DCASM B,0 and number of miniband based CRUs for F P0 for FFT size 512 (from [10, Table 791]). . . 23
2.17 PRU to P RUSB mapping for the test case. . . 29
2.18 PRU to P RUM B mapping for the test case. . . 31
2.19 P RUM B to P P RUM B mapping for the test case. . . 32
2.20 P RUSB and P P RUM B to P RUF P 1 mapping for the test case. . . 33
2.21 The mapping of P RUF P 1s to CRU/DRU for the test case. . . 35
2.22 OFDMA parameters (Table 793 in [10]). . . 37
2.23 OFDMA parameters (from [10, Table 794]). . . 37
2.24 Example of MIMO midamble structure for the case of 4 transmit antennas (Figure 502 in [10]). . . 37
2.25 Location of the A-Preamble symbols (Figure 506 in [10]). . . 41
2.26 Symbol structure of PA-Preamble in frequency domain (Figure 507 in [10]). . 41
2.27 Sequences of PA-Preamble in frequency domain (from [10, Table 796]). . . . 42
2.28 Boosting factors for PA-Preamble symbols (Table 797 in [10]). . . 43 2.29 Parameters and values for resource allocation of SFH (from [10, Table 806]). 45
2.30 Illustration of periodic transmission of S-SFH SPs with example transmission periodicity of 40 ms, 80 ms and 160 ms for SP1, SP2 and SP3, respectively
(Figure 515 in [10]). . . 46
2.31 Transmission Periodicity of S-SFH SPs (from [10, Table 807]). . . 47
2.32 Example of locations of A-MAP regions in a TDD system (From [10, Fig 516]). 47 2.33 Structure of an A-MAP region (Figure 517 in [10]). . . 48
2.34 Physical processing block diagram for the P-SFH (Figure 518 in [10]). . . 50
2.35 Physical processing block diagram for the S-SFH (Figure 519 in [10]). . . 51
2.36 Chain of non-user specific A-MAP IE to non-user specific A-MAP symbols (Figure 520 in [10]). . . 52
2.37 Chain of HF-A-MAP IE to HF-A-MAP symbols (Figure 521 in [10]). . . 53
2.38 Chain of PC-A-MAP IE to PC-A-MAP symbols (Figure 522 in [10]). . . 55
2.39 Chain of A-A-MAP IE to A-A-MAP symbols (Figure 523 in [10]). . . 56
3.1 DL MIMO architecture (Fig.537 in [10]). . . 59
3.2 Codebook subsets used for non-adaptive precoding in DL-DLRU and NLRU (from [10, Table 842]). . . 63
3.3 Codebook subsets used for non-adaptive precoding in DL-SLRU (from [10, Table 843]). . . 64
3.4 Downlink MIMO modes (from [10, Table 844]). . . 66
3.5 DL MIMO parameters (from [10, Table 845]). . . 67
3.7 Supported Permutation for each DL MIMO mode outside the OL region (from
[10, Table 846]). . . 70
3.8 Supported Permutation for each DL MIMO mode in the OL region (from [10, Table 847]). . . 70
3.9 Types of open-loop regions (from [10, Table 848]). . . 72
3.10 MIMO feedback modes (from [10, Table 849]). . . 75
3.11 DL MIMO control parameters (from [10, Table 850]). . . 76
3.12 DL MIMO control parameters (from [10, Table 850]). . . 77
3.13 Subset selection of the base codebok for four transmit antennas (from [10, Table 866]). . . 78
3.14 Base codebook c(2,1,3) (from [10, Table 851]). . . 79
3.15 Base codebook c(2,2,3) (from [10, Table 852]). . . 80
3.16 Quantization parameters for diagonal entries of R (from [10, Table 867]). . . 81
3.17 Quantization parameters for non-diagonal entries of R (from [10, Table 868]). 82 3.18 D(2,1,4) codebook (from [10, Table 869]). . . 86
3.19 D(2,2,4) codebook (from [10, Table 870]). . . 86
3.20 Codebook subsets used for non-adaptive precoding in UL DLRU and NLRU (from [10, Table 921]). . . 88
3.21 Codebook subsets used for non-adaptive precoding in UL SLRU (from [10, Table 922]). . . 88
3.22 Uplink MIMO modes (from [10, Table 923]). . . 89
3.24 Supported permutation for each UL MIMO mode (from [10, Table 925]). . . 91
3.25 UL MIMO control parameters (from [10, Table 926]). . . 92
4.1 System model. . . 94
4.2 System model with feedback. . . 97
4.3 Number of flops of different matrix operations [16]. . . 101
4.4 Number of flops per subcarrier of each method. . . 102
5.1 Suburban macrocell channel model [9]. . . 104
5.2 IEEE 802.16m TDD frame structure for 10 MHz [9]. . . 106
5.3 TDD structure for 10 MHz. . . 106
5.4 MSE and SER for QPSK resulting from simulation compared with theory in AWGN. . . 108
5.5 MSE and SER for QPSK using ZF equalizer with no feedback in Suburban channel. . . 109
5.6 MSE and SER for QPSK using ZF equalizer with SVD-Based search feedback in Suburban channel. . . 110
5.7 MSE and SER for QPSK using ZF equalizer with MMSE-Based exhaustive search feedback in Suburban channel. . . 111
5.8 MSE and SER for QPSK using ZF equalizer with MaxminSNR-Based search feedback in singlepath Suburban channel. . . 112
5.9 MSE and SER for QPSK using ZF equalizer with Optimum precoder feedback in Suburban channel. . . 113
5.10 MSE and SER for QPSK using ZF equalizer with different feedback methods at 30 km/h in Suburban channel. . . 114 5.11 MSE and SER for QPSK using ZF equalizer with different feedback methods
at 120 km/h in Suburban channel. . . 115 5.12 MSE and SER for QPSK using MMSE equalizer with no feedback in Suburban
channel. . . 116 5.13 MSE and SER for QPSK using MMSE equalizer with SVD-Based search
feed-back in Suburban channel. . . 117 5.14 MSE and SER for QPSK using MMSE equalizer with MMSE-Based
exhaus-tive search feedback in Suburban channel. . . 118 5.15 MSE and SER for QPSK using MMSE equalizer with MaxminSNR-Based
search feedback in Suburban channel. . . 119 5.16 MSE and SER for QPSK using MMSE equalizer with Optimum precoder
feedback in Suburban channel. . . 120 5.17 MSE and SER for QPSK using MMSE equalizer with different feedback
meth-ods at 30 km/h in Suburban channel. . . 121 5.18 MSE and SER for QPSK using MMSE equalizer with different feedback
meth-ods at 120 km/h in Suburban channel. . . 122 5.19 MSE and SER for QPSK using ZF and MMSE equalizer with different
feed-back methods at 30 km/h in Suburban channel. . . 123 5.20 MSE and SER for QPSK using ZF and MMSE equalizer with different
5.21 MSE and SER for QPSK using ZF equalizer with no feedback in multipath Suburban channel. . . 126 5.22 MSE and SER for QPSK using ZF equalizer with SVD-Based search feedback
in multipath Suburban channel. . . 127 5.23 MSE and SER for QPSK using ZF equalizer with MMSE-Based exhaustive
search feedback in multipath Suburban channel. . . 128 5.24 MSE and SER for QPSK using ZF equalizer with MaxminSNR-Based search
feedback in multipath Suburban channel. . . 129 5.25 MSE and SER for QPSK using ZF equalizer with Optimum precoder feedback
in multipath Suburban channel. . . 130 5.26 MSE and SER for QPSK using ZF equalizer with different feedback methods
at 30 km/h in multipath Suburban channel. . . 131 5.27 MSE and SER for QPSK using ZF equalizer with different feedback methods
at 120 km/h in multipath Suburban channel. . . 132 5.28 MSE and SER for QPSK using MMSE equalizer with no feedback in multipath
Suburban channel. . . 133 5.29 MSE and SER for QPSK using MMSE equalizer with SVD-Based search
feed-back in multipath Suburban channel. . . 134 5.30 MSE and SER for QPSK using MMSE equalizer with MMSE-Based
exhaus-tive search feedback in multipath Suburban channel. . . 135 5.31 MSE and SER for QPSK using MMSE equalizer with MaxminSNR-Based
5.32 MSE and SER for QPSK using MMSE equalizer with Optimum precoder feedback in multipath Suburban channel. . . 137 5.33 MSE and SER for QPSK using MMSE equalizer with different feedback
meth-ods at 30 km/h in multipath Suburban channel. . . 138 5.34 MSE and SER for QPSK using MMSE equalizer with different feedback
meth-ods at 120 km/h in multipath Suburban channel. . . 139 5.35 MSE and SER for QPSK using ZF and MMSE equalizer with different
feed-back methods at 30 km/h in multipath Suburban channel. . . 140 5.36 MSE and SER for QPSK using ZF and MMSE equalizer with different
feed-back methods at 120 km/h in multipath Suburban channel. . . 141 5.37 MSE and SER for QPSK using ZF equalizer with SVD-Based search feedback
in multipath Suburban channel with TX4 RX4 Rank2. . . 142 5.38 MSE and SER for QPSK using ZF equalizer with MMSE-Based exhaustive
search feedback in multipath Suburban channel with TX4 RX4 Rank2. . . . 143 5.39 MSE and SER for QPSK using ZF equalizer with Optimum precoder feedback
in multipath Suburban channel with TX4 RX4 Rank2. . . 144 5.40 MSE and SER for QPSK using ZF equalizer with different feedback methods
at 30 km/h in multipath Suburban channel with TX4 RX4 Rank2. . . 145 5.41 MSE and SER for QPSK using ZF equalizer with different feedback methods
List of Tables
2.1 PRU Structures for Different Types of Subframe . . . 7 2.2 Mapping Between PRU Index and P RUSB Index for the Test Case . . . 28
2.3 Mapping Between P RU Index and P P RUM B Index for the Test Case . . . . 30
2.4 Mapping Between P RUM B Index and P P RUM B Index for the Test Case . . 30
2.5 Mapping Between P RUSB Index, P P RUM B Index, and P RUF P 1 Index for
the Test Case . . . 32 2.6 Mapping Between P RUSB Index, P P RUM B Index, and P RUF P 2 Index for
the Test Case . . . 34 2.7 Mapping Between P RUSB Index, P P RUM B Index, and P RUF P 3 Index for
the Test Case . . . 34 2.8 Mapping Between P RUSB Index, P P RUM B Index, and P RUF P 4 Index for
the Test Case . . . 34 2.9 Random Sequence for the Test Case . . . 34 2.10 Mapping Between P RUF P 1 Index and CRUF P 1/DRUF P 1 Index . . . 35
Chapter 1
Introduction
1.1
Scope of the Work
Orthogonal frequency division multiple access (OFDMA) has emerged as one of the prime multiple access schemes for broadband wireless networks. Some major examples are IEEE 802.16 Mobile WiMAX, IEEE 802.20 and 3GPP LTE. As a special case of multicarrier multiple access schemes, OFDMA exclusively assigns each subchannel to only one user, eliminating intra-cell interference [12]. In frequency selective channels, an intrinsic advantage of OFDMA is its capability to exploit the so-called multiuser diversity provided by multipath channels. Other advantages of OFDMA include finer granularity and better link budget [12]. OFDMA can be easily generated using an inverse fast Fourier transform (IFFT) and received using a fast Fourier transform (FFT).
The IEEE 802.16 standard committee has developed a group of standards for wireless metropolitan area networks (MANs). OFDMA is used in the 2 to 11 GHz systems. The IEEE Standard 802.16-2004 was for broadband wireless access systems that provide a variety of wireless access services to fixed outdoor and indoor users. The 802.16e was designed to support terminal mobility with a speed up to 120 km/h [15]. The last two standards have now been combined in IEEE 802.16-2009.
In response to International Telecommunication Union Radiocommunication Section (ITU-R)’s plan for the fourth-generation mobile communication standard IMT-Advanced, the IEEE 802.16 standards group has set up the 802.16m (i.e., Advanced WiMAX) task group. The new frame structure developed by IEEE 802.16m can be compatible with IEEE 802.16e, reduce communication latency, support relay, and coexist with other radio access techniques (in particular, LTE). In the IEEE 802.16m working group, the high-level system description and evaluation methodology are captured in [9]. MIMO technologies again play an essential role in achieving the ambitious target set, which requires the 802.16m system to deliver twice the performance gain over a baseline 802.16e system in various measures, including sector throughput, average user throughput, and peak data rate, as well as cell-edge performance. Several new MIMO ingredients are proposed. Noticeable ones are transformed codebook for beamforming feedback, differential beamforming feedback, open-loop multiuser MIMO, and collaborative multicell MIMO [13]. We study the MIMO architecture and signal pro-cessing technology for 802.16m. In particular, we consider the zero-forcing (ZF) equalizer and minimum mean-square error (MMSE) equalizer approach wherein we employ the tech-nique proposed in [2]. We follow [2] to introduce the ZF and MMSE equalizer with channel feedback selection problem.
This thesis focuses on the precoding and equalization for the spatial multiplexing mode of IEEE 802.16m closed-loop (CL) multi-input multi-output (MIMO) systems. A problem associated with precoding is that the channel state information must be known at the trans-mitter. This may be difficult since the bandwidth of the feedback channel is usually limited. Thus, a codebook-based limited feedback precoding scheme is generally used. The main idea is to quantize the precoding matrix and feedback the index of the optimum precoder. We propose two methods to select the best precoder from a finite set of precoding matrices and we will compare with two other methods. We simulate the IEEE 802.16m MIMO system
and study the performance of different channel feedback selection methods.
1.2
Organization and Contribution of this Thesis
This thesis is organized as follows.
• Chapter 2 introduces the IEEE 802.16m OFDMA. • Chapter 3 introduces the IEEE 802.16m MIMO.
• Chapter 4 introduces equalization and closed-loop MIMO technology. • Chapter 5 presents some simulation results.
• Chapter 6 gives the conclusion and indicates future work. In this thesis, our work can be summarized as following:
• Study IEEE 802.16m OFDMA. • Study IEEE 802.16m MIMO
• Study equalization and closed-loop MIMO
– Study the equalization and feedback technology in the IEEE 802.16m MIMO systems.
– Analyze the error performance.
– Discuss on the feedback selection method. Main contribution of this thesis are as follows:
2. Propose two feedback selection schemes for the IEEE 802.16m MIMO systems. 3. Develop an error analysis scheme for the IEEE 802.16m MIMO systems.
4. Compare performance of four different feedback selection methods for the IEEE 802.16m MIMO systems.
Chapter 2
Introduction to IEEE 802.16m
OFDMA
We first introduce the basic concepts of the OFDMA in the specification of IEEE 802.16m and then MIMO techniques for multicarrier modulation in the specification of IEEE 802.16m. Much of the material in this chapter is taken from [10].
2.1
Basic OFDMA Symbol Structure and Frame
Struc-ture [10]
The Advanced Air Interface (AAI) defined by IEEE 802.16m is designed for nonline-of-sight (NLOS) operation in the licensed frequency bands below 6 GHz. The AAI supports both time-division duplexing (TDD) and frequency-division duplexing (FDD) operation, allowing also half-FDD (H-FDD) operation at mobile stations (MSs). Unless otherwise specified, the frame structure attributes and baseband processing are common for all duplex modes.
The AAI uses OFDMA as the multiple access scheme in the downlink and the uplink. The material of this section is taken from [10].
2.1.1
OFDMA Basic Terms
We introduce some basic terms in the OFDMA physical layer (PHY) of IEEE 802.16m. They help us understand the concepts of subcarrier allocation and transmission in IEEE 802.16m OFDMA.
• Physical and logical resource units: A physical resource unit (PRU) is the basic physical unit for resource allocation. It comprises Psc(= 18) consecutive subcarriers by Nsym
consecutive OFDMA symbols, Nsym = 6 for type-1 subframes, Nsym = 7 for type-2
subframes, and Nsym = 5 for type-3 subframes. Table 1.1 summarizes the PRUs sizes
for different subframe types. A logical resource unit (LRU) is the basic logical unit for distributed and localized resource allocations. An LRU contains Psc· Nsym subcarriers
for the three types of subframes. The LRU includes the pilots that are used in a PRU. The effective number of subcarriers in an LRU depends on the number of allocated pilots.
• Contiguous resource unit: The localized resource unit, also known as contiguous re-source unit (CRU), contains a group of subcarriers which are contiguous across the localized resource allocations. The size of CRU equals the size of PRU, i.e., Psc
sub-carriers by Nsym OFDMA symbols.
• Distributed resource unit: A distributed resource unit (DRU) contains a group of sub-carriers which are spread across the distributed resource allocations within a frequency partition. The size of DRU also equals the size of PRU.
2.1.2
Frequency Domain Description
An OFDMA symbol is made up of subcarriers, the number of which determines the discrete Fourier transform (DFT) size used. There are several subcarrier types:
Table 2.1: PRU Structures for Different Types of Subframe Subframe Type Number of Subcarriers Number of Symbols
Type-1 18 6 Type-2 18 7 Type-3 18 5 • Data subcarriers: used for data transmission.
• Pilot subcarriers: used for various estimation purposes.
• Null subcarriers: no transmission at all, used for guard bands and DC subcarrier. The purpose of the guard bands is to help enable proper bandlimiting.
2.1.3
Primitive Parameters
Four primitive parameters characterize the OFDMA symbols: • BW : the nominal channel bandwidth.
• Nused: number of used subcarriers (which includes the DC subcarrier).
• n: sampling factor. This parameter, in conjunction with BW and Nused, determines
the subcarrier spacing and the useful symbol time. This value is given in Figs. 2.1 and 2.2 for each nominal bandwidth.
• G: This is the ratio of CP time to “useful” time, i.e., Tcp/Ts. The following values
shall be supported: 1/16, 1/8, and 1/4.
2.1.4
Derived Parameters
• NF F T: smallest power of two greater than Nused.
• Sampling frequency: Fs= ⌊n · BW/8000⌋ × 8000.
• Subcarrier spacing: △f = Fs/NF F T.
• Useful symbol time: Tb = 1/△f.
• CP time: Tg = G × Tb.
• OFDMA symbol time: Ts= Tb+ Tg.
• Sampling time: Tb/NF F T.
2.1.5
Frame Structure
The AAI basic frame structure is illustrated in Fig. 2.3. Each 20 ms superframe is divided into four 5-ms radio frames. When using the same OFDMA parameters as in Figs. 2.1 and 2.2 with channel bandwidth of 5, 10, or 20 MHz, each 5-ms radio frame further consists of eight subframes for G = 1/8 and 1/16. With channel bandwidth of 8.75 or 7 MHz, each 5-ms radio frame further consists of seven and six subframes, respectively, for G = 1/8 and 1/16. In the case of G = 1/4, the number of subframes per frame is one less than that of other CP lengths for each bandwidth case. A subframe shall be assigned for either downlink (DL) or uplink (UL) transmission. There are four types of subframes:
• Type-1 subframe consists of six OFDMA symbols. • Type-2 subframe consists of seven OFDMA symbols. • Type-3 subframe consists of five OFDMA symbols.
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Figure 2.2: More OFDMA parameters (from [10, Table 775]).
• Type-4 subframe consists of nine OFDMA symbols. This type shall be applied only to UL subframe for the 8.75 MHz channel bandwidth when supporting the WirelessMAN-OFDMA frames.
The basic frame structure is applied to FDD and TDD duplexing schemes, including H-FDD MS operation. The number of switching points in each radio frame in TDD systems shall be two, where a switching point is defined as a change of directionality, i.e., from DL to UL or from UL to DL.
A data burst shall occupy either one subframe (i.e., the default transmission time interval [TTI] transmission) or contiguous multiple subframes (i.e., the long TTI transmission). The long TTI in FDD shall be 4 subframes for both DL and UL. The long TTI in TDD shall be the whole DL (UL) subframes for DL (UL) in a frame. Every superframe shall contain a superframe header (SFH). The SFH shall be located in the first DL subframe of the superframe and shall include broadcast channels.
2.2
Downlink Transmission in IEEE 802.16m OFDMA[10]
Again this section is mainly taken from [10]. Each DL subframe is divided into 4 or fewer frequency partitions; each partition consists of a set of PRUs across the total number of OFDMA symbols available in the subframe. Each frequency partition can include contiguous (localized) and/or non-contiguous (distributed) PRUs. Each frequency partition can be used for different purposes such as fractional frequency reuse (FFR) or multicast and broadcast services (MBS). Fig. 2.4 illustrates the downlink physical structure in an example of two frequency partitions with frequency partition 2 including both CRUs and DRUs.
Figure 2.4: Example of downlink physical structure (Fig.485 in [10]).
2.2.1
Subband Partitioning
The PRUs are first subdivided into subbands and minibands where a subband comprises N1 adjacent PRUs and a miniband comprises N2 adjacent PRUs, where N1 = 4 and N2 =
1. Subbands are suitable for frequency selective allocations as they provide a contiguous allocation of PRUs in frequency. Minibands are suitable for frequency diverse allocation and are permuted in frequency.
The number of subbands reserved is denoted by KSB. The number of PRUs allocated to
subbands is denoted by LSB, where LSB = N1· KSB, depending on system bandwidth. A 5,
4 or 3 bit field called Downlink Subband Allocation Count (DSAC) determines the value of KSB depending on FFT size. The DSAC is transmitted in the SFH. The remaining PRUs
are allocated to minibands. The number of minibands in an allocation is denoted by KM B.
The total number of PRUs is denoted as NP RU where NP RU = LSB + LM B. The maximum
number of subbands that can be formed is denoted as Nsub where Nsub= ⌊NP RU/N1⌋.
Figs. 2.5 and 2.7 show the mapping between SAC and KSB for FFT sizes 2048, 1024, and
512, respectively. For system bandwidths in the range of (10, 20]MHz, the mapping between DSAC and KSB is based on Fig 2.5, and the maximum valid value of KSB is NP RU/4 − 3.
For system bandwidths in the range of [5, 10]MHz, the mapping between DSAC and KSB
is based on Fig 2.6, and the maximum valid value of KSB is NP RU/4 − 2.
The subband PRUs and miniband PRUs are denoted P RUSB and P RUM B, respectively.
The set of P RUSB is numbered from 0 to LSB− 1, and the set of P RUM B is numbered from
0 to LM B− 1. The mapping of PRUs to P RUSB is
P RUSB[j] = P RU[i], j = 0, 1, ..., LSB− 1, (2.1) where i = N1· {{⌈ Nsub KSB⌉ · ⌊ j N1⌋ + ⌊⌊ j N1⌋ · GCD(Nsub, ⌈NKsubSB⌉)
Nsub ⌋} mod Nsub} + j · N1
(2.2) with GCD(x, y) being the greatest common divisor of x and y. And the mapping of PRUs to P RUM B is defined as P RUM B[k] = P RU[i], k = 0, 1, ..., LM B− 1, (2.3) where i = N1 · {⌈NKsubSB·⌉ · ⌊Nk1⌋ + ⌊⌊ k N1⌋ · GCD(Nsub,⌈NsubKSB⌉)
Nsub ⌋} mod Nsub
+(k) mod N1, KSB > 0,
k, KSB = 0.
(2.4) Fig 2.8 illustrates the PRU to P RUSB and P RUM B mapping for a 10MHz bandwidth system
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Figure 2.5: Mapping between SAC and KSB for FFT size 2048 (from [10, Table 783]).
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Figure 2.7: Mapping between SAC and KSB for FFT size 512 (from [10, Table 785]).
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Figure 2.8: PRU to P RUSB and P RUM B mapping for BW = 10 MHz, and KSB=7.
2.2.2
Miniband Permutation
The miniband permutation maps the P RUM Bs to Permuted P RUM Bs (P P RUM Bs) to ensure
that frequency diverse PRUs are allocated to each frequency partition. Fig. 2.8 together with Fig 2.19 shows an example. The following equation provides a mapping from P RUM B to
P P RUM B: P P RUM B[j] = P RUM B[i], j = 0, 1, ..., LM B− 1, (2.5) where i = (q(j) mod D) · P + ⌊q(j)D ⌋, (2.6) P = min(KM B, N1/N2), (2.7) r(j) = max(j − ((KM B mod P ) · D), 0), (2.8) q(j) = j + ⌊ r(j) D − 1⌋, (2.9) D = ⌊KPM B + 1⌋. (2.10)
2.2.3
Frequency Partitioning
The P RUSB and P P RUM B are allocated to one or more frequency partitions. By default,
only one partition is present. The maximum number of frequency partitions is 4. The frequency partition configuration is transmitted in the SFH in a 4 or 3 bit called the Downlink Frequency Partition Configuration (DFPC) depending on FFT size. Frequency Partition Count (FPCT) defines the number of frequency partitions. Frequency Partition Size (F P Si)
defines the number of PRUs allocated to F Pi. FPCT and F P Si are determined from FPC
as shown in Figs.2.10 to 2.12. A 3, 2, or 1-bit parameter called the Downlink Frequency Partition Subband Count (DFPSC) defines the number of subbands allocated to F Pi, i > 0.
Fig. 2.13 continues the examples in Fig. 2.8 and 2.9 and shows how P RUSB and P P RUM B
can be mapped to frequency partitions.
The number of subbands in ith frequency partition is denoted by KSB,F Pi. The number
of minibands is denoted by KM B,F Pi, which is determined by F P Si and DF P SC fields.
When DF P C = 0, DF P SC must be equal to 0. The number of subband PRUs in each frequency partition is denoted by LSB,F P i, which is given by LSB,F Pi = N1· KSB,F P i. The
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Figure 2.9: Mapping from PRUs to P RUSB and P P RUM B for BW = 10 MHz and KSB =
7.
number of miniband PRUs in each frequency partition is denoted by LM B,F P i, which is given
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Figure 2.10: Mapping between DFPC and frequency partitioning for FFT size 2048 (from [10, Table 786]). GHI G JKLM H NJ OPOP QRQ RS T GH UV G H W V GH XV GH Y Z GHI[ G H\ U GH\ ] TP^_Z ` ab`b`b` a c d ef ` a `bababa g ` hijk l c d efm n o pq rrs tcd ef ugv h ij w l pq r rs tcd ef ugv h ij x l pq r rs tcd ef ugv n abababa y cd e fm g o pq rrs tcd ef uyv pq rrs tcd e f uyv g gbababa y cd e fm g o pq rrs tcd ef uzv pq rrs tcd e f uzv y {bababa y cd e fm g o pq rrs tcd ef u|v pq rrs tcd e f u |v { }b{b{b{ y cd e fm g o pq rrs tcd e fo {u n yv pq r rs tcd efo {u n yv z `babab` n ` cd e f u n p rs~l a n ` p rs~l g ababab` g cd e fm n o pq rrs tcd ef ugv pq rrs tcd ef ugvp rs~l a n ` p rs~l g
Figure 2.11: Mapping between DFPC and frequency partitioning for FFT size 1024 (from [10, Table 787]).
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Figure 2.12: Mapping between DFPC and frequency partitioning for FFT size 512 (from [10, Table 788]). ¶ · ¸ ¹ ¶ º ¶» ¶¼ ¶½ »¼ »½ »¶ »· ¼º ¼» ¼¼ ¼½ ¼¾ ¼¿ ½º ½» ½¸ ½¹ ½¾ ½¿ ¶¶ ¶· ¶¸ ¶¹ ¾ ¿ »º »» »¸ »¹ »¾ »¿ ¼¶ ¼· ¼¸ ¼¹ ½¼ ½½ ½¶ ½· º » ¼ ½ ¶ · ¸ ¹ ¶ º ¶» ¶¼ ¶½ »¼ »½ »¶ »· ¼º ¼» ¼¼ ¼½ ¼¾ ¼¿ ½º ½» ½¸ ½¹ ½¾ ½¿ ¶¶ ¶· ¶¸ ¶¹ ¾ ¿ »º »» »¸ »¹ »¾ »¿ ¼¶ ¼· ¼¸ ¼¹ ½¼ ½½ ½¶ ½· º » ¼ ½ ¶º ¶» ¶ ¼ ¶ ½ ¾ ¿ »¸ »¹ ¼¶ ½¼ º » ¶ · ¸ ¹ »¼ »½ »¶ »· »º ¼· ½½ ¼ ¼º ¼» ¼¼ ¼½ ¼¾ ¼¿ ½º ½» »¾ ¼¸ ½¶ ½ ½¸ ½¹ ½¾ ½¿ ¶¶ ¶· ¶¸ ¶¹ »» »¿ ¼¹ ½·
Figure 2.13: Frequency partitioning for BW = 10 MHz, KSB = 7, FPCT = 4, FPS = 12,
by KSB,F P i = KSB− (F P CT − 1) · DF P SC, i = 0, F P CT = 4, DF P SC, i > 0, F P CT = 4, DF P SC, i > 0, F P SC = 3, DF P C = 1, KSB− (F P CT − 1) · DF P SC, i = 0, F P CT = 3, DF P C 6= 1, DF P SC, i = 1, 2, F P CT = 3, DF P C 6= 1, DF P SC, i = 1, 2, F P CT = 2, KSB, i = 0, F P CT = 1, (2.11)
where F P CT = 2 and DF P SC = KSB/2. The number of minibands for each frequency
partition is given by
KM B,F P i = (F P Si− KSB,F P i· N1)/N2, 0 ≤ i < F P CT. (2.12)
The mapping of subband PRUs and miniband PRUs to the frequency partition is given by P RUF P i(j) = P RUSB(k1), 0 ≤ j < LSB,F P i, P P RUM B(k2), LSB,F P i≤ j < (LSB,F P i+ LM B,F P i), (2.13) where k1 = i−1 X m=0 LSB,F P m+ j (2.14) and k2 = i−1 X m=0 LSB,F P m+ j − LSB,F P i. (2.15)
Fig. 2.13 depicts the frequency partitioning for BW = 10MHz, KSB = 7, F P CT = 4, F P S0 =
F P Si = 12, and DF P SC = 2.
2.3
Cell-Specific Resource Mapping [10]
P RUF P is are mapped to LRUs. All further PRU and subcarrier permutations are constrained
2.3.1
CRU/DRU Allocation
The partition between CRUs and DRUs is done on a sector-specific basis. Let LSB−CRU,F Pi
and LM B−CRU,F Pi denote the number of allocated subband CRUs and miniband CRUs for
F Pi (i ≥ 0). The number of total allocated subband and miniband CRUs, in units of a
subband (i.e., N1 PRUs), for F Pi (i ≥ 0) is given by the downlink CRU allocation size,
DCASi. The numbers of subband-based and miniband-based CRUs in F P0 are given by
DCASSB,0 and DCASM B,0, in units of a subband and a miniband, respectively. When
DF P C = 0, DCASi must be equal to 0.
For F P0, the value of DCASSB,0 is explicitly signaled in the SFH as a 5, 4 or 3 bit field
to indicate the number of subbands in unsigned binary format, where DCASSB,0 ≤ KSB,F P.
A 5, 4 or 3 bit downlink miniband based CRU allocation size (DCASM B,0) is sent in the
SFH only for partition F P0, depending on FFT size. The number of subband based CRUs
for F P0 is given by
LSBCRU,F P0 = N1· DCASSB,0. (2.16)
The mapping between DCASM B,0and the number of miniband based CRUs for F P0is shown
in the Figs. 2.14 to 2.16 for FFT sizes of 2048, 1024 and 512, respectively. For those system bandwidths in range of (10, 20], the mapping between DCASM B,0 and number of
miniband-based CRUs for F P0 is based on Fig. 2.14, and the maximum valid value of LM B−CRU,F P0
is less than ⌊88 · NP RU⌋/96. For system bandwidths in the range of [5, 10], the mapping
between DCASM B,0 and number of miniband-based CRUs for F P0 is based on Fig. 2.15,
and the maximum valid value of LM B−CRU,F P0 is less than ⌊42 · NP RU⌋/48.
For F Pi (i > 0, F P CT ≥ 2) only one value for DCASi is explicitly signaled for all i > 0,
in the SFH as a 3, 2 or 1 bit field to signal the same numbers of allocated CRUs for F Pi
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Figure 2.14: Mapping between DCASM B,0 and number of miniband based CRUs for F P0
for FFT size 2048 (from [10, Table 789]).
For F Pi (i > 0, F P CT ≥ 2), the number of subband CRUs (LSB−CRU,F Pi) and miniband
CRUs (LM B−CRU,F Pi) are derived using the two equations.
LSB−CRU,F Pi = N1· min{DCASi, KSB,F Pi}, (2.17) LM B−CRU,F Pi = 0, DCASi ≤ KSB,F Pi, (DCASi− KSB,F Pi) · N1, DCASi > KSB,F Pi, (2.18) When F P CT = 2, DCASSB,i and DCASM B,i for i = 1 and 2 are signaled using the
DCASSB,0 and DCASM B,0fields in the SFH. Since F P0 and F P3 are empty, LSB−CRU,F P0 =
LM B−CRU,F P0 = LDRU,F P0 = 0 and LSB−CRU,F P3 = LM B−CRU,F P3 = LDRU,F P3 = 0. For i = 1
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Figure 2.15: Mapping between DCASM B,0 and number of miniband based CRUs for F P0
for FFT size 1024 (from [10, Table 790]).
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Figure 2.16: Mapping between DCASM B,0 and number of miniband based CRUs for F P0
the mappings in Figs. 2.14 through 2.16 for FFT sizes of 2048, 1024 and 512, respectively. The number of CRUs in each frequency partition is denoted LCRU,F Pi, where
LCRU,F Pi = LSB−CRU,F Pi+ LM B−CRU,F Pi. (2.19)
The number of DRUs in each frequency partition is denoted by LDRU,F Pi, where
LDRU,F Pi = F P Si− LCRU,F Pi. (2.20)
and F P Si is the number of PRUs allocated to F Pi.
The mapping from P RUF Pi to CRUF Pi is given by
CRUF P i[j] =
P RUF P i[j], 0 ≤ j < LSB−CRU,F Pi· N1, 0 ≤ i < F P CT,
P RUF P i[k + LSB−CRU,F Pi · N1], LSB−CRU,F Pi ≤ j < LCRU,F P i, 0 ≤ i < F P CT.
(2.21) where k = s[j − LSB−CRU,F Pi], with s[ ] being the CRU/DRU allocation sequence defined as
s[j] = {P ermSeq(j) + DL P ermBase} mod {F P Si− LSB−CRU,F Pi · N1} (2.22)
where P ermSeq() is the permutation sequence of length (F P Si− LSB−CRU,F Pi) and is
de-termined by SEED = IDcell · 343 mod 210, DL P ermBase is an interger ranging from 0
to 31, which is set to preamble IDcell. The mapping of P RUF P i to DRUF P i is given by
DRUF P i[j] = P RUF P i[k + LSB−CRU,F Pi], 0 ≤ j < LDRU,F P i (2.23)
where k = s[j + LCRU,F Pi − LSB−CRU,F Pi].
2.3.2
Subcarrier Permutation
The DL DRUs are used to form two stream distributed logical resource unit (DLRU)s by subcarrier permutation. The subcarrier permutation defined for the DL distributed resource allocations within a frequency partition spreads the subcarriers of the DRU across the whole
distributed resource allocations. The granularity of the subcarrier permutation is equal to a pair of subcarriers.
After mapping all pilots, the remainder of the used subcarriers are used to define the distributed LRUs. To allocate the LRUs, the remaining subcarriers are paired into contiguous tone-pairs. Each LRU consists of a group of tone-pairs.
Let LSC,l denote the number of data subcarriers in lth OFDMA symbol within a PRU,
i.e., LSC,l = PSC − nl, where nl denotes the number of pilot subcarriers in the lth OFDMA
symbol within a PRU. Let LSP,l denote the number of data subcarrier-pairs in the lth
OFDMA symbol within a PRU and is equal to LSC,l/2. A permutation sequence PermSeq()
is defined in section 2.3.3, performs the DL subcarrier permutation as follows. For each lth OFDMA symbol in the subframe:
1. Allocate the nl pilots within each DRU as described in Section 2.3.3. Denote the data
subcarriers of DRUF P i[j] in the lth OFDMA symbol as
SCDRU,j,lF P i [k], 0 ≤ j < LDRU,F P i, 0 ≤ k < LSC,l. (2.24)
2. Renumber the LDRU,F P i · LSC,l data subcarriers of the DRUs in order, from 0 to
LDRU,F P i· LSC,l − 1. Group these contiguous and logically renumbered subcarriers
into LDRU,F P i· LSP,l pairs and renumber them from 0 to LDRU,F P i· LSP,l − 1. The
renumbered subcarrier pairs in the lth OFDMA symbol are denoted as
RSPF P i,l[u] = {SCDRU j,lF P i [2v], SCDRU j,lF P i [2v + 1]}, 0 ≤ u < LDRU,F P iLSP,l, (2.25)
where j = ⌊u/LSP,l⌋, v = {u} mod (LSP,l).
3. Apply the subcarrier permutation formula to map RSPF P i,l into the sth distributed
LRU, s = 0, 1, . . . , LDRU,F P i− 1, where the subcarrier permutation formula is given by
where
k = LDRU,F P i· f(m, s, l) + g(P ermSeq(), s, m, l). (2.27)
In the above, 1. SCF P i
LRU s,l[m] is the mth subcarrier pair in the lth OFDMA symbol in the sth distributed
LRU of the tth AAI subframe;
2. m is the subcarrier pair index, 0 to LSP,l− 1;
3. l is the OFDMA symbol index, 0 to Nsym− 1;
4. s is the distributed LRU index, 0 to LDRU,F P i− 1;
5. P ermSeq() is the permutation sequence of length LDRU,F P i and is determined by
SEED = {IDcell · 1367} mod 210; and
6. g(P ermSeq(), s, m, l) is a function with value from the set [0,LDRU,F P i-1], which is
defined according to
g(P ermSeq(), s, m, l) = {P ermSeq[{f(m, s) + s + l} mod {LDRU,F P i}]
+DL P ermBase} mod LDRU,F P i (2.28)
where DL P ermBase is an integer ranging from 0 to 31 (Section 2.3.3), which is set to preamble IDcell, and f (m, s, l) = (m + 13 · (s + l))mod LSP,l.
2.3.3
Random Sequence Generation
The permutation sequence generation algorithm with 10-bit SEED (Sn−10, Sn−9, ..., Sn−1)
shall generate a permutation sequence of size M according to the following process: • Initialization
1. Initialize the variables of the first order polynomial equation with the 10-bit seed, SEED. Set d1 = ⌊SEED/25⌋ + 619 and d2 = SEED mod 25.
2. Initialize the maximum iteration number, N = 4.
3. Initialize an array A with size M to contents 0, 1, . . . , M − 1(i.e.,A[i] = i, for0 ≤ i < M).
4. Initialize the counter i to M − 1. 5. Initialize x to −1.
• Repeat the following steps if i > 0 1. Increment x by i.
2. Calculate the output variable of y = {(d1· x + d2) mod 1031} mod M.
3. If y ≤ i, set y = y mod (i + 1). 4. Swap A[i] and A[y].
5. Decrement i by 1.
• P ermSeq[i] = A[i],where 0 ≤ i < M.
2.4
Test Case Generation
We set some system parameters to build a frame as a test case. The parameters are as given below. We will walk through some of the derived mappings in subsequent subsections.
• NPRU = 48 • SAC = 6 • KSB = 7
Table 2.2: Mapping Between PRU Index and P RUSB Index for the Test Case P RUSB Index 0 1 2 3 4 5 6 7 8 9 10 11 12 13 PRU Index 40 41 42 43 4 5 6 7 12 13 14 15 20 21 P RUSB Index 14 15 16 17 18 19 20 21 22 23 24 25 26 27 PRU Index 22 23 28 29 30 31 36 37 38 39 44 45 46 47 • KM B = 20 • LSB = N1· KSB = 28 • Nsub = 12 • FPC = 1 • FPCT = 4 • FP0:FP1:FP2:FP3 = 1:1:1:1 • FPSC = 2 • ID Cell = 2 • SEED = 343 • DL PermBase = 0
2.4.1
Subband Partitioning
The 48 PRUs map to the subbands according the formulas described in Section 2.3.1. Fig. 2.17 illustrates the PRU to P RUSB mapping for a 10 MHz bandwidth with KSB = 6.
) * + , -. / 0 1 )( )) )* )+ ), )-). )/ )0 )1 *( *) ** *+ *, *-*. */ *0 *1 +( +) +* ++ +, +-+. +/ +0 +1 ,( ,) ,* ,+ ,, ,-,. ,/ , -. / ,( ,) ,* ,+ )* )+ ), )-*( *) ** *+ *0 *1 +( +) +. +/ +0 +1 ,, ,-,. ,/
Table 2.3: Mapping Between P RU Index and P P RUM B Index for the Test Case
P RUM B Index 0 1 2 3 4 5 6 7 8 9
P RU Index 0 1 2 3 8 9 10 11 16 17 P RUM B Index 10 11 12 13 14 15 16 17 18 19
P RU Index 18 19 24 25 26 27 32 33 34 35
Table 2.4: Mapping Between P RUM B Index and P P RUM B Index for the Test Case
P P RUM B Index 0 1 2 3 4 5 6 7 8 9
P RUM B Index 0 8 16 24 32 1 9 17 25 33
P P RUM B Index 10 11 12 13 14 15 16 17 18 19
P RUM B Index 2 10 18 26 34 3 11 19 27 35
2.4.2
Miniband Partitioning
The remainder of the PRUs are allocated to minibands according the formulas given pre-viously. Fig. 2.18 illustrates the PRU to P RUM B mapping for a 10 MHz bandwidth with
KM B = 20. Table 2.3 shows the resulting mapping between the PRU index and the P RUM B
index.
2.4.3
Miniband Permutation
The miniband permutation maps the P RUM Bs to Permuted P RUM Bs (P P RUM Bs) to ensure
that frequency diverse PRUs are allocated to each frequency partition. The mapping rule is as described previously. Fig. 2.19 illustrates the P RUM B to P P RUM B mapping for a
10 MHz bandwidth with KM B = 20. Table 2.4 shows the resulting mapping between the
2 3 45 44 46 47 42 43 89 8: 86 87 ;8 ;; ;9 ;: 5 4 8 ; 4 8 ; 9 : 6 7 2 3 45 44 48 4; 49 4: 46 47 42 43 85 8 4 88 8; 89 8: 86 87 82 83 ;5 ; 4 ;8 ;; ;9 ;: ;6 ;7 ;2 ;3 95 94 98 9; 99 9: 96 97
< = >? >> >@ >A >< >= BC BD B@ BA EB EE EC ED > B E < = >? >> >@ >A > < > = BC BD B@ BA EB EE EC ED > B E
Figure 2.19: P RUM B to P P RUM B mapping for the test case.
Table 2.5: Mapping Between P RUSB Index, P P RUM B Index, and P RUF P 1 Index for the
Test Case
P RUF P 1 index 0 1 2 3 4 5 6 7 8 9 10 11
P RUSB index 0 1 2 3 x x x x x x x x
P P RUM B index x x x x 0 1 2 3 4 5 6 7
2.4.4
Frequency Partitioning
The P RUSB and P P RUM B are allocated to the frequency partitions. There are 4 frequency
partitions used because F P CT = 4. The P RUSB and P P RUM B map to frequency partitions
1, 2, 3, and 4 according the formulas given previously. Fig. 2.20 illustrates the P RUSB and
P P RUM B to frequency partitions mapping for a 10 MHz bandwidth. Tables 2.5 through 2.8
show the resulting mapping between the P RUSB index, P P RUM B index, and P RUF P iindex
F G H I FK FL FM KL KM KF KG LJ LK LL LM LN LO MJ MK MH MI MN MO FF FG FH F I N O KJ KK KH KI KN KO LF LG LH LI ML MM MF MG J K L M FK FL FM N O KH KI LF M L J K F G H I KL KM KF KG KJ L G M M L LJ LK L L L M LN LO MJ MK KN LH MF M MH MI MN MO FF FG FH FI KK KO LI MG
Table 2.6: Mapping Between P RUSB Index, P P RUM B Index, and P RUF P 2 Index for the
Test Case
P RUF P 2 index 0 1 2 3 4 5 6 7 8 9 10 11
P RUSB index 4 5 6 7 8 9 10 11 x x x x
P P RUM B index x x x x x x x x 8 9 10 11
Table 2.7: Mapping Between P RUSB Index, P P RUM B Index, and P RUF P 3 Index for the
Test Case
P RUF P 3 index 0 1 2 3 4 5 6 7 8 9 10 11
P RUSB index 12 13 14 15 16 17 18 19 x x x x
P P RUM B index x x x x x x x x 12 13 14 15
Table 2.8: Mapping Between P RUSB Index, P P RUM B Index, and P RUF P 4 Index for the
Test Case
P RUF P 4 index 0 1 2 3 4 5 6 7 8 9 10 11
P RUSB index 20 21 22 23 24 25 26 27 x x x x
P P RUM B index x x x x x x x x 16 17 18 19
Table 2.9: Random Sequence for the Test Case k 0 1 2 3 4 5 6 7 PermSeq[k] 2 1 0 3 4 5 6 7
2.4.5
Random Sequence
We generate a random sequence with ID Cell = 2, DL P ermBase = 0. Then SEED = {ID Cell × 1367} mod 210 = 343. The random sequence is generated according to the
Table 2.10: Mapping Between P RUF P 1 Index and CRUF P 1/DRUF P 1 Index P RUF P 1 index 0 1 2 3 4 5 6 7 8 9 10 11 CRUF P 1 index 0 1 2 3 4 5 6 7 x x x x DRUF P 1 index x x x x x x x x 0 4 3 2 P Q R S T U V W X Y U T UU P Q R S T U V W X Y UT UU
Figure 2.21: The mapping of P RUF P 1s to CRU/DRU for the test case.
2.4.6
CRU/DRU Allocation
The P RUF P 1s map to the CRUs and DRUs according to the formulas given previously.
Fig. 2.21 illustrates the P RUF P 1to CRU/DRU mapping for a 10 MHz bandwidth. Table 2.10
show the resulting mapping between the P RUF P 1 index and the CRU/DRU index.
2.5
MIMO Midamble [10]
Again this section is mainly taken from [10]. MIMO midamble is used for preferred matrix index (PMI) selection in CL MIMO. For open loop (OL) MIMO, midamble can be used to calculate channel quality index (CQI). MIMO midamble shall be transmitted every frame on the second DL AAI subframe. The midamble signal occupies the first OFDMA symbol in a
DL type-1 or type-2 AAI subframe. For type-1 AAI subframe, the remaining 5 consecutive symbols form a type-3 AAI subframe. For type-2 AAI subframe, the remaining 6 consecutive symbols form a type-1 AAI subframe. The MIMO midamble signal transmitted by the ABS antenna is defined s(t) = Re ( ej2πfct k=NXused−1 k=0 k6=Nused− 1 2 bke j2π(k−Nused− 1 2 )∆f (t−Tg) ) (2.29)
where bk is a coefficient modulating a subcarrier in the midamble symbol
bk =
2.18 · {1 − (2G([k + u + offset(fft)] mod fft))}, k 6= Nused2− 1and (k − s) mod (3 × Nt) = 3g + (⌊ IDcell 256 ⌋ + ⌊ k − s N1− NSC)⌋ mod 3, 0, otherwise. (2.30) where k is the subcarrier index (0 ≤ k ≤ Nused− 1), Nused is the number of used subcarriers,
G(x) is the Golay sequence defined in Fig. 2.22 (0 ≤ x ≤ 2047), fft is the FFT size used, u is a shift value given by u = mod (IDcell, 256), of f set(f f t) is an FFT size specific offset as defined in Fig. 2.23, g is an advanced base station (ABS) transmit antenna index (0 ≤ g ≤ Nt − 1), Nt is the number of ABS transmit antennas, parameter s = 0 for
k ≤ Nused2− 1 and s = 1, for k > (Nused− 1)
2 . An example of the physical structure of the MIMO midamble is shown in Fig. 2.24 for the case with 4 transmit (TX) antennas and IDcell = 0. In Fig. 2.22, the hexadecimal series should be read as a sequence of bits where each 16-bit word starts at the most significant bit (MSB) and ends at the least significant bit (LSB) where the second word’s MSB follows. The first bit of the sequence is referenced as having offset 0.
Figure 2.22: OFDMA parameters (Table 793 in [10]). ZZ[\] ^ _ ` a ab _c def g he iedf je kid f e
Figure 2.23: OFDMA parameters (from [10, Table 794]).
Figure 2.24: Example of MIMO midamble structure for the case of 4 transmit antennas (Figure 502 in [10]).
2.6
Usage of Downlink Pilots [10]
Again this section is mainly taken from [10]. The demodulation pilots in a given PRU on a given pilot stream shall be precoded the same way as the data transmitted on the same stream in that PRU. In DLRU the data transmitted in a given PRU on a given stream may be sent to several advanced mobile station (AMS)s but in different tones using the same precoder.
Two pilot streams shall always be transmitted in the DLRUs, whether inside or outside the OL region type 0, and whether or not data is being transmitted by the ABS in all DLRUs. If no data are transmitted by the ABS on all or some contiguous logical resource unit (CLRU)s in the OL region type 1 or type 2, then max Mtpilots shall still be transmitted
across all CLRUs in that OL region. If no data are transmitted by the ABS on all or some CLRUs outside any OL region, then pilots shall not be transmitted on the CLRUs where no data are sent.
The precoder may be adaptive (user-specific) or non-adaptive (non user-specific) depend-ing on the DL MIMO mode. Non-adaptive precoders are determined accorddepend-ing to the DL MIMO mode, the number of streams, the type of LRU, operation inside or outside the OL region, and the physical index of the subband or miniband where the precoder is applied.
In MU-MIMO transmissions in CLRU each pilot stream is dedicated to one AMS. The AMS shall use its dedicated pilot stream for channel estimation within the allocation. Other pilot streams may be used for inter-stream interference estimation. The total number of streams in the transmission and the index of the dedicated pilot stream are indicated in the DL Basic Assignment A-MAP IE, DL Persistent Allocation A-MAP IE or DL Subband Assignment A-MAP IE.
follows:
• In DLRU: the 2-streams non-adaptively precoded common pilots across the DLRU should be used for channel estimation by all AMSs allocated a burst in the DLRU. Within each frequency partition, all pilots are shared by all AMSs for demodulation in DLRU. Only the pilots located within a physical subband should be used for channel estimation within that subband.
• In CLRU: The AMS should use its dedicated pilot streams for channel estimation in the allocation. Pilots are not shared by AMSs for demodulation in CLRU, whether they are non-adaptively or adaptively precoded.
MIMO feedback measurements at the AMS should be performed as follows:
• For MIMO feedback reports requested with a MIMO feedback mode for operation in an OL region, measurements should be taken on the max Mt streams non-adaptively
precoded pilots in that OL region. All pilots are shared by all AMSs for MIMO feedback measurements in each OL region.
Wideband CQI reports inside OL region should be averaged over OL region pilots of the PRUs in the frequency partition 0.
• For MIMO feedback reports requested with a MIMO feedback mode for operation out-side the OL region, measurements should be taken on the downlink MIMO midamble. Wideband CQI reports outside OL region (measured on MIMO midamble) should be averaged over the frequency partition indicated by Frequency Partition Indicator (PFI) in Feedback Allocation A-MAP IE or according to FPCT for feedback allocated by Feedback Polling A-MAP IE.
For reports requested with a MIMO feedback mode for OL MIMO operation, the AMS should adjust the non-precoded MIMO channel estimated from the midamble by applying it with the non-adaptive precoder according to the MIMO feedback modes (MFM), the subband index and assumption on space time code(STC) rate.
For reports requested with a MIMO feedback mode for CL MIMO operation, the AMS should adjust the non-precoded MIMO channel estimated from the midamble with an estimated adaptive precoder.
Subband CQI reports (inside and outside OL region) should be reported for sub-bands in subband logical resource unit (SLRU)s indicated by superframe header(SFH).
2.7
Downlink control structure [10]
Again this section is mainly taken from [10].
2.7.1
Advanced Preamble
There are two types of Advanced Preamble (A-Preamble): primary advanced preamble (PA-Preamble) and secondary advanced preamble (SA-(PA-Preamble). One PA-Preamble symbol and three SA-Preamble symbols exist in a superframe. An A-Preamble symbol is located at the first symbol of a frame. The PA-Preamble is located at the first symbol of the second frame in a superframe while a SA-Preamble is located at the first symbol of each of the remaining three frames. Fig. 2.25 depicts the location of A-Preamble symbols.
2.7.2
Primary Advanced Preamble (PA-Preamble)
The length of sequence for PA-Preamble is 216 regardless of the FFT size. PA-Preamble carries the information of system bandwidth and carrier configuration. With the subcarrier
Figure 2.25: Location of the A-Preamble symbols (Figure 506 in [10]). index 256 reserved for DC, the allocation of subcarriers is given by
P AP reambleCarrierSet = 2 · k + 41 (2.31) where P AP reambleCarrierSet specifies all subcarriers allocated to the PA-Preamble and k is a running index from 0 to 215. Fig. 2.26 depicts the symbol structure of the PA-Preamble in the frequency domain.
Fig. 2.27 lists the sequences of the PA-Preamble in hexadecimal format. A series is
![Figure 2.11: Mapping between DFPC and frequency partitioning for FFT size 1024 (from [10, Table 787]).](https://thumb-ap.123doks.com/thumbv2/9libinfo/7643339.138278/42.918.137.750.451.874/figure-mapping-dfpc-frequency-partitioning-fft-size-table.webp)
![Figure 2.12: Mapping between DFPC and frequency partitioning for FFT size 512 (from [10, Table 788])](https://thumb-ap.123doks.com/thumbv2/9libinfo/7643339.138278/43.918.146.739.62.327/figure-mapping-dfpc-frequency-partitioning-fft-size-table.webp)
