Chapter 3 Passive Quadrature Signal Generations and Their Applications on BJT-
3.2 Applications on BJT Gilbert Mixers
A Gilbert mixer, especially with a bipolar mixing core [31], has a region of flat gain as a function of the LO input power because around four times the thermal voltage (4VT≈0.1V) is needed to make current commutate in emitter-coupled differential pairs. Typically, there is a flat-gain region of around 5 to 10 dB. Fig. 3-6 illustrates the conversion gain versus LO input power for two identical Gilbert mixers with different LO path loss. The curve B in Fig. 3-6 is the right-shifted version of the curve A because the LO path loss of the curve B is larger than that of the curve A. It shows that the Gilbert mixer is tolerant of different LO path loss if the LO input
power is in the common overlapped range shown in Fig. 3-6.
This advantage of a Gilbert mixer is more significant in BJT switching core than CMOS switching core because a CMOS differential pair requires 2VOVto make the current fully switch. Thus, a wider flat-gain regain is obtained using BJT core and thus much amplitude imbalance is tolerable.
Small Signal
Perfect Switching
(Large Signal) Saturation
Common Region
A B
LO Power (dBm)
Conver sion G ain (dB )
Fig. 3-6 Conversion gain versus LO power of two identical Gilbert mixers (A and B) with different LO path loss.
The followings are two interesting applications using this concept of a flat-gain region in a BJT-based Gilbert mixer. Section 3.2.1 introduces a 5.7 GHz I/Q downconversion mixer is demonstrated using 0.35-μm SiGe BiCMOS technology. A quarter-wavelength coupled line and two center-tapped transformers are utilized to generate differential quadrature LO signals. The wide flat-gain region of the BJT Gilbert mixer tolerates the LO amplitude imbalance due to the substrate loss of the quadrature coupler. On the other hand, an ultra-wideband (UWB) I/Q downconverter with an LR-CR quadrature generator is demonstrated using the same technology and is described in Section 3.2.2. The I/Q outputs of this generator are always in quadrature phase at any frequency while the BJT-type active mixer inherently
tolerates much LO power difference for a flat gain response. Consequently, the amplitude imbalance and phase error of the I/Q outputs are less than 1 dB and 2°
when the RF frequency covering 3-11 GHz.
3.2.1 5.7-GHz I/Q Downconversion Mixers With an LO Quadrature Coupler
In the past, quadrature signals generated with reactive passive components were implemented on the GaAs semi-insulating substrate and high-resistive silicon substrate [32]. There is a need to integrate the quadrature coupler in the standard silicon process for the silicon RFIC era. Thus, the quadrature coupler has been demonstrated by using interconnect metals with ground shielding plane to avoid the substrate loss in the standard silicon process [33]. However, the low dielectric constant of the interconnect dielectrics results in a large size quadrature coupler. It is benign to take advantage of the high silicon dielectric constant. However, the high substrate loss in a standard silicon process leads to amplitude imbalance between the coupling port and through port in a quadrature coupler. It is difficult to employ a quadrature coupler with large amplitude imbalance. Recently, a Marchand balun consisting of two quadrature couplers has been demonstrated directly on lossy silicon substrate [31]. The demonstrated Marchand balun has balanced output signals even though each constituent quadrature coupler has unbalanced outputs.Here, the use of a quadrature coupler directly on the lossy silicon substrate is employed in the LO ports of a Gilbert I/Q downconverter. The impact of quadrature generator amplitude imbalance is minimized by proper choice of the LO input power.
Applying the advantage of BJT mixer with a wide flat-gain region to the I/Q mixer design, we can achieve perfect current switching in both I/Q paths by properly choosing LO input power while a quadrature signal generator is employed in the LO stage with perfect phase relations at the desired frequency but unequal amplitude
caused by different LO path loss. Here, the quadrature generator is composed of a quarter-wavelength coupled line and two transformers as shown in Fig. 3-7.
V
LOFig. 3-7 LO quadrature signal generator using a quarter-wavelength coupled line and two center-tapped transformers.
Fig. 3-8 Schematic of the SiGe BiCMOS I/Q downconverter with a reactive passive LO quadrature signal generator and an RF Marchand balun. The LO quadrature generator is shown in Fig. 3-7.
The LO signal is injected at the incident port of the quarter-wavelength coupled line. The LO ports of the I mixer are connected to the coupling port through the subsequent transformer while the through port is connected to the LO ports of the Q mixer through the other transformer. This type of quadrature generator has been implemented in the RF stage, but the intrinsic loss imbalance caused by the lossy silicon substrate in the differential quadrature generator makes IF I/Q outputs obviously unequal.
The schematic of the I/Q downconverter utilizing an LO differential quadrature signal generator and an RF Marchand balun is shown in Fig. 3-8. The Marchand balun is employed in the RF stage to convert an unbalanced signal into two balanced signals in spite of the lossy silicon substrate [31], [34]. A planar Marchand balun, consisting of two back-to-back quarter-wavelength coupled lines, has both coupling ports connected with short ends, the incident port in the opposite quarter-wavelength coupled line left open while the signal is incident in the input incident port and two balanced signals appear at the two isolated ports as shown in Fig. 3-8. The Marchand balun is followed by the common-base-configured transistors, Q1-Q4, for I/Q channels because of their excellent frequency response and convenience for broadband impedance matching. The short ends in the coupling ports of the Marchand balun are also utilized as the DC return ground of the common-base-configured transistors, Q1-Q4. Each quarter-wavelength coupled line in the Marchand balun is replaced by its shunt C- series L- shunt C lumped versions [35] as shown in Fig. 3-8 to further reduce the Marchand balun size at the cost of narrower bandwidth. The center frequency of each lumped quarter-wavelength coupled line used in the Marchand balun of the RF stage is designed around 5.7 GHz. This lumped-type quadrature coupler can also be employed in the LO port to further reduce the chip size at the cost of bandwidth.
Fig. 3-9 Die photo.
The die photo of the proposed I/Q downconversion mixer is shown in Fig. 3-9.
The die size is 1×1 mm2 and is dominated by the passive elements consisting of a Marchand balun, a quarter-wavelength coupled line and two transformers. The emitter size of all the SiGe HBTs in common-base-configured transistors (Q1-Q4) and Gilbert mixer core (Q5-Q12) are 0.3 μm in width and 1.9 μm in length, respectively. A PMOS current combiner load is employed in each I/Q downconversion mixer to combine two differential signals into a single-ended output. A common-collector output buffer in each IF port is designed to facilitate the on-wafer measurement.
In order to shrink the size of the quarter-wavelength coupled line employed in the LO differential quadrature generator, the interleave transformer type quarter-wavelength coupled lines are employed as shown in Fig. 3-7. The LO quadrature coupler has a 7-μm line width, a 3-μm line spacing and an outer diameter of 266 μm to generate the quadrature phase in coupling and through ports at around 5.7 GHz. There are two 2:3 transformers following the quadrature coupler. Each transformer consists of two constituent inductors with line width, line spacing, and
outer diameter of 2.6 μm, 1.8 μm, and 140 μm, respectively. The dc bias voltage for the LO port is fed from the center-tapped point in the secondary coil of the transformer. On the other hand, the size of 3-stage 5-6 GHz PPF is about 180×180 μm2 with 9.3 dB loss by calculating the RC values in [36].
Fig. 3-10 I/Q-channel conversion gain versus LO power.
-60 -50 -40 -30 -20 -10
Fig. 3-11 Power performance.
The fabricated SiGe BiCMOS quadrature downconverter with the single-ended LO, RF and IF ports is convenient for on-wafer measurements. The supply voltage is 2.5 V and the total power consumption is 3.875 mW. The measured IF I/Q outputs have flat gain regions for LO power ranging from −10 dBm to 1 dBm, and −7 dBm to
3 dBm, respectively when RF=5.7 GHz, LO=5.665 GHz and IF=35 MHz. In other words, the coupling port has about 3 dB more loss than the through port has in the quadrature coupler. The conversion gain difference between I and Q channels varies within 1 dB for LO power from −7 dBm to 1 dBm as shown in Fig. 3-10.
1 10 100
2 3 4 5 6 7 8 9 10
Conversion Gain (dB)
IF Frequency (MHz)
I Channel Q Channel
-3dB BW=120MHz LO: 5.665 GHz/-3dBm
Fig. 3-12 IF bandwidth.
Fig. 3-13 I/Q output waveforms.
Fig. 3-11 shows the power performance of the downconverter for each I/Q channel. 7 dB conversion gain, −26 dBm IP1dB, and −18 dBm IIP3 are achieved for both I/Q outputs. The IF bandwidth of the SiGe BiCMOS I/Q downconverter is 120 MHz as shown in Fig. 3-12. The measured I/Q downconverter output waveforms are shown in Fig. 3-13. The average phase error is below 2° and amplitude error is below
0.3 dB. The RF input return loss is better than 10 dB from 4 GHz to 7 GHz. The measured double sideband noise figure is 20 dB. The I/Q channel performance of the demonstrated downconverter is well balanced in spite of the unbalanced LO path loss caused by the substrate loss. The overall performance is summarized in TABLE. 3.1.
TABLE.3.1PERFORMANCE SUMMARY OF THE 5.7-GHZ I/QDOWNCONVERTER
RF Frequency (GHz) 5.7
Conversion Gain (dB) 7
I/Q Amplitude Imbalance (dB) 0.3
I/Q Phase Error (°) <2
LO Power (dBm) −4
DSB Noise Figure (dB)` 20
IP1dB (dBm) −26
IIP3 (dBm)
−18
Supply Voltage (V) 2.5
Power Consumption (mW) 3.875
Technology 0.35-m SiGe BiCMOS
On the other hand, the use of a quadrature generator in the RF path results in a 2 dB amplitude imbalance for I/Q channel output as shown in [37]-[38]. The insertion loss of 7 dB, magnitude imbalance of 4 dB and phase error of 2° from 5-6 GHz were measured for the quadrature coupler in [38].
3.2.2 UWB I/Q Downconversion Mixers With an LR-CR Quadrature Generator
The block diagram of the UWB I/Q downconverter is shown in Fig. 3-14. The ratio of VQ (CR-path) and VI (LR-path) can be described as Eqn. (3.14), as described in Section 3.1.3. Besides, the input impedance of the LR-CR quadrature generator is always equals to the load impedance (R) under the balanced condition by Eqn. (3.15).Fig. 3-14 Block diagram of the UWB I/Q downconverter and schematic of the micromixer employed in this downconverter.
Eqn. (3.14) indicates that the outputs are always 90° out of phase under the balanced condition (L/C=R2) as long as the load impedance (R=50) and the center frequency [f0 1 (2 LC) ]are specified. However, the amplitude imbalance is proportional to the operating frequency with 6 dB/octave. For a UWB application, the center frequency (f0) is designed at 5.5 GHz with L=1.447 nH and C=0.58 pF. As a result, the I/Q signals have the maximum amplitude imbalance of 6 dB within the 4:1 bandwidth, i.e., from f0/2 (2.75 GHz) to 2fo (11 GHz).
Such amplitude imbalance seems impossible for wideband applications at first sight; however, a BJT-type active mixer only needs a small LO voltage swing for a full current commutation and there is an LO input power range of around 10 dB for a flat gain response [31].
0.5
0region in bipolar is larger than that in MOS.
Fig. 3-15 Conversion gain as a function of LO power while an LO LR-CR quadrature generator is used.
When the operating frequency is lower than f0, the LO voltage swing of at Q channel (VQ) is smaller than the LO voltage swing at I channel (VI) by Eqn. (3.14).
Thus, the Q-mixer needs larger LO input power to reach the flat gain region than the I-mixer as shown in Fig. 3-15. When the operating frequency is at f0, the conversion gain curve of both channels are overlapped because VI=VQ. On the contrary, the
Q-mixer needs less LO input power than the I-mixer when the LO frequency is higher
than f0. Consequently, the suitable LO power can be selected so that the IF I/Q output amplitude imbalance can be minimized with an excellent quadrature accuracy.Fig. 3-16 Schematic of the micromixer employed in this downconverter.
The micromixer topology [6] is chosen in this work because it achieves an RF wideband matching and provides balanced RF currents by the input transconductance balun stage as shown in Fig. 3-16. A broadband Marchand balun [34] consisting of two coupled-line couplers is employed to generate differential signals at the LO port of each I/Q mixer. The dc bias (Vbias) of the mixer core is fed from the ac-ground node of the Marchand balun as shown in Fig. 3-14. A 50- resistor in series with a dc-blocking capacitor is utilized at each output node to achieve a wideband 50-
input impedance of each balun. It is worth mentioning that an active balun can also be applied with a compact die size [39] but the linearity of an active balun should be designed carefully since the limited voltage swing degrades the operating bandwidth.
The die photo of the UWB I/Q downconverter is shown in Fig. 3-17 and the die size is 1.05×0.95 mm2. The supply voltage is 3.3 V with the current consumption of 4.7 mA for each mixer.
Fig. 3-17 Die photo.
Fig. 3-18(a) shows the conversion gain of the I/Q downconverter with respect to the LO power when f=5.5 GHz=f0 and Fig. 3-18(b) shows the results when f=3.282 GHz<f0 and f=10.146 GHz>f0. The I/Q mixers need the same LO power for a full current commutation when the LO frequency is at the center frequency. On the other hand, the I-mixer needs less (more) LO input power to reach the flat gain region than the Q-mixer when the LO frequency is lower (higher) than the center frequency due to the amplitude imbalance of the LR-CR topology as described in Eqn. (3.14). However, the I/Q mixers still have a wide common region of the LO power for balanced quadrature outputs. Thus, an 8 dBm LO power is chosen for all the following measurements.
As shown in Fig. 3-19, the RF 3-dB bandwidth ranges from 2 GHz to 11 GHz while the IP1dB and IIP3 are better than −9 dBm and 6 dBm, respectively. Fig. 3-20 shows the amplitude imbalance <1 dB and the quadrature phase error <2° with respect to the input RF frequency when IF frequency is 150 MHz. The input return loss for RF and LO ports are better than 10 dB when frequency ranging from dc to 20 GHz and from 1.6 GHz to 13 GHz, respectively, as shown in Fig. 3-21.
-20 -10 0 10 20
Fig. 3-18 (a)Conversion gain with respect to the LO power when LO frequency is 5.5 GHz (b)LO frequency is 3.282 and 10.146 GHz, respectively.
The LO matching is achieved due to the 50-Ω impedance of the LR-CR topology as described in Eqn. (3.15) while the micromixer topology facilitates the RF impedance matching. Fig. 3-22 shows the IF 1-dB bandwidth of 500 MHz with the noise figure below 16.5 dB as IF frequency ranging from 200 kHz to 100 MHz thanks to the inherently low flicker noise corner of the SiGe HBT devices.
0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15
Fig. 3-19 Conversion gain and power performance (including IP1dB and IIP3).
2 4 6 8 10 12 14
Fig. 3-20 Amplitude imbalance and phase difference of I/Q outputs.
0 2 4 6 8 10 12 14 16 18 20
Fig. 3-21 LO and RF input return loss.
0.1 1 10 100 1000 Double Sideband Noise Figure
LO=7.128 GHz Noise Figure (dB)
Fig. 3-22 Conversion gain and double-sideband noise figure.
The overall performance is summarized in TABLE. 3.2.
TABLE.3.2PERFORMANCE SUMMARY OF THE UWBDOWNCONVERTER USING LR-CR QUADRATURE GENERATOR
RF Frequency (GHz) 2-11
IF 1-dB Bandwidth (MHz) 500
Conversion Gain (dB) 7
I/Q Amplitude Imbalance (dB) 0.3
I/Q Phase Error (°) <2
Power Consumption (mW) 31
Technology 0.35-m SiGe BiCMOS