國立交通大學
電子工程學系 電子研究所碩士班
碩 士 論 文
應用於數位電視廣播規格之同步化系統設計
Synchronization System
for DVB-T/H Standard
學生 ： 李家豪
指導教授 ： 李鎮宜 教授
中華民國九十五年七月
應用於數位電視廣播規格之同步化系統設計
Synchronization System
for DVB-T/H Standard
研 究 生：李家豪 Student：Chia-Hao Lee
指導教授：李鎮宜 Advisor：Chen-Yi Lee
國 立 交 通 大 學
電子工程學系 電子研究所 碩士班
碩 士 論 文
A ThesisSubmitted to Institute of Electronics
College of Electrical Engineering and Computer Science National Chiao Tung University
in Partial Fulfillment of the Requirements for the Degree of
Master of Science in
Electronics Engineering July 2006
Hsinchu, Taiwan, Republic of China
應用於數位電視廣播規格之同步化系統設計
學生：李家豪 指導教授：李鎮宜 教授
國立交通大學
電子工程學系 電子研究所碩士班
摘要
在本論文中，我們提出了一個應用於數位電視廣播規格的同步化系統設計。此同步 化系統設計由一組載波相位校準、取樣頻率漂移同步以及快速同步設計組成。我們團隊 於 2006 年 ISSCC 期刊中發表的 DVB-T/H 基頻接收機設計為本篇論文的參考設計。我們 提出的載波相位校準可以克服在 Rayleigh 衰減通道中，取樣頻率漂移 20ppm、都卜勒頻 率 70Hz、載波頻率漂移 10.3，、訊雜比 34 分貝，其中，傳輸模式為 2K 模式，Viterbi 編碼率 2/3。第二部份，與傳統設計相比，取樣頻率漂移估計的正確度相比可達到 2~5 倍；取樣頻率漂移追蹤也可達到 3 倍的收斂速度。快速同步的設計也比傳統設計的同步 速度快上 2~4.5 倍，而在 timing-slicing 的架構下，也使得我們的電路功率消耗節省 約莫百分之 10，接收機記憶體減少 1~3Mbits。在不同的傳輸模式以及通到效應下，整 體的系統效能只比完美同步系統損失 0~0.3 分貝。並且在硬體上我們整合了同步系統電 路中重複的部分，與 ISSCC2006 的設計相比，同步化設計電路的部份可以解省百分之 46 的硬體。Synchronization System
for DVB-T/H Standard
Student：Chia-Hao Lee Advisor：Dr. Chen-Yi Lee
Institute of Electronics Engineering
National Chiao Tung University
ABSTRACT
In this thesis, a synchronization system is proposed for Digital Video Broadcasting-Terrestrial/Handheld (DVB-T/H) standard. This synchronization system comprises 3 parts: carrier phase alignment, sampling clock synchronization, and fast synchronization schemes. In this paper, we take the DVB-T/H baseband receiver [] proposed in ISSCC2006 as the reference design. First, the proposed carrier phase alignment can overcome sampling clock offset (SCO) 20ppm, Doppler frequency 70Hz, carrier frequency offset (CFO) 10.3 at Rayleigh fading channel with SNR 34dB in 2K mode and Viterbi coderate 2/3. Second, comparing with conventional design, the SCO estimation accuracy can reach 2~5 times; SCO tracking can obtain the convergent time 3 times fast. Then, the fast synchronization can increase the speed of synchronization 2~4.5 times. Under time-sling architecture, the saving of power consumption can improve about 10% and receiver buffer is decreased 1~3Mbits. The overall performances are only lose 0~0.3dB in different transmission modes. Finally, on the vision of hardware, we integrate the similar part of synchronization algorithms and compare with [] in ISSCC2006. We can save the hardware 46%.
誌
謝
兩年的時光轉眼間就過去了，在Si2 這個大家庭學習讓我感到相當充實。不但學到 了許多專業知識，為人處事方面更是受益良多。 能完成這本論文，我要特別感謝 李鎮宜教授不厭其煩的指導與研究方向的指引。 以及實驗室的所有學長姐、同學、學弟妹之間無私的指導與討論，每每讓我在研究時找 到新的思路和方法。 在這裡，特別感謝DVB group 的黎峰學長、昱偉學長、陳元學長、成偉學長、盧忠 學長、英豪以及義閔學弟，感謝大家這兩年來的腦力激盪與相互討論，不但使我在相關 的研究領域有所精進，更學習到團隊合作的可貴。還要感謝與我同屆的康正、毅宏、婉 君、俊彥與志龍，在這兩年內，我們一起經歷過許多，有你們的陪伴，使我這兩年的碩 士生涯充滿了多彩多姿的回憶。 最後，我要由衷的感謝我的父母及家人，感謝你們多年來的栽培及細心，讓我能順 利完成碩士的學業。更要感謝我的女友瑞文，這幾年來有妳的一路相伴，使我的生活更 豐富有趣，心裡有個寄託的地方。僅將此論文獻給你們，以表達我最深的感激。
Contents
CHAPTER 1 . INTRODUCTION... 1
1.1 MOTIVATION...1
1.2 INTRODUCTION TO DVB-T/H SYSTEM...2
1.3 INTRODUCTION TO TIME-SLICING TECHNOLOGY... 11
1.4 ORGANIZATION OF THIS THESIS...13
CHAPTER 2 . SYNCHRONIZATION ALGORITHMS... 15
2.1 INTRODUCTION TO UNSYNCHRONOUS PROBLEMS...15
2.1.1 Effect of Carrier Frequency Offset and Sampling Clock Offset ...16
2.1.2 Effect of Symbol Timing Offset ...20
2.1.3 Effect of Carrier Phase Mismatch ...2
2.2 CARRIER PHASE ALIGNMENT...27
2.3 FAST SYNCHRONIZATION SYSTEM...32
2.3.1 TPS Decode ...34
2.3.2 RS Packet Synchronization ...37
2.4 SAMPLING CLOCK OFFSET SYNCHRONIZATION...42
2.4.1 Conventional SCO Estimation...42
2.4.2 Proposed SCO Estimation ...44
2.4.3 Conventional Tracking Loop ...45
2.4.4 Proposed Tracking Loop...45
CHAPTER 3 . SIMULATION AND PERFORMANCE ANALYSIS ... 49
3.2.1 Multipath Fading Channel Model ...55
3.2.2 Doppler Spread Model ...58
3.2.3 Carrier Frequency Offset and Sampling Clock Offset model...59
3.3 PERFORMANCE ANALYSIS...59
3.3.1 Carrier Phase Alignment...59
3.3.2 Fast Synchronization ...63
3.3.3 Sampling Clock Synchronization...65
3.3.4 Overall System Performance...70
CHAPTER 4 . HARDWARE INTEGRATION ... 74
4.1 ACQUISITION...75
4.2 TRACKING...79
4.3 INTEGRATION RESULT...80
CHAPTER 5 . CONCLUSION AND FUTURE WORK... 83
List of Figures
FIG 1.1FUNCTIONAL BLOCK DIAGRAM OF THE ADDITIONAL FEATURES...4
FIG 1.2 FRAME STRUCTURE...7
FIG 1.3GENERATION OF PRBS SEQUENCE...8
FIG 1.4:TIMING-SLICING TECHNOLOGY IN DVB-H SYSTEM... 11
FIG 1.5THE TIME SLICING/MPE-FEC BUFFER IN THE RECEIVER...13
FIG 2.1PHASE ROTATION IN TIME DOMAIN FOR LONG TIME RECEPTION WHEN Ε=0.01 ...18
FIG 2.2SPECTRUM OF FIVE SUBACRRIERS IN CARRIER FREQUENCY OFFSET ENVIRONMENT...18
FIG 2.3PHASE ROTATION DUE TO TIMING DRIFTS...20
FIG 2.4ISI-FREE REGION...21
(A)SYMBOL TIMING OFFSET IN THE ISI-FREE REGION...22
FIG 2.5PHASE ROTATION OF SYMBOL TIMING OFFSET Ε =2 AND Ε =5 ...22
FIG 2.6MAPPING CONSTELLATION...2
FIG 2.7 THE INFLUENCE OF FRAME POSITION ERROR TO INTERPOLATIONS...24
FIG 2.8RELATIVE CIR DUE TO INACCURATE FFT WINDOW...24
FIG 2.9INTERPOLATION OF 2D CHANNEL ESTIMATION...25
FIG 2.10 THE PHASE ROTATION OF PILOTS’CFR CAUSED BY DIFFERENT SYMBOL TIMING OFFSET ...26
FIG 2.11RELATIVE CHANNEL IMPULSE RESPONSE WITH MULTI-PATH...27
FIG 2.12LOCATION OF PHASE ALIGNMENT IN RECEIVER PLATFORM...28
FIG 2.13COMPARISON OF PERFORMANCE BETWEEN 2-DCE AND 1-DCE...29
FIG 2.14 THE PHASE ROTATION OF PILOTS’CFR CAUSED BY DIFFERENT SYMBOL TIMING OFFSET ...30
FIG 2.16PROPOSED TIME AXIS PHASE ALIGNMENT ARCHITECTURE...32
FIG 2.17 COMPOSITION OF COMPLETED SYNCHRONIZATION TIME...33
FIG 2.18LOCATION OF FAST SYNCHRONIZATION IN RECEIVER PLATFORM...34
FIG 2.19 THE OPERATION OF CONVENTIONAL TPS WORD...36
FIG 2.20 THE OPERATION OF PROPOSED TPS WORD...37
FIG 2.23LOCATION OF FAST SYNCHRONIZATION IN RECEIVER PLATFORM...42
FIG 2.24ARCHITECTURE OF SCO ESTIMATION [1]...43
FIG 2.25 THE ARCHITECTURE OF PROPOSED SCO ESTIMATION...44
FIG 2.26TIMING DIAGRAM OF CONVENTIONAL SCO TRACKING ARCHITECTURE...45
FIG 2.27 THE ARCHITECTURE OF PROPOSED SCO TRACKING LOOP...46
FIG 2.28 TIMING DIAGRAM OF PROPOSED SCO TRACKING LOOP ARCHITECTURE...47
FIG 2.29 THE ARCHITECTURE OF PROPOSED SCO TRACKING LOOP WITH SP METHOD...48
FIG 3.1BLOCK DIAGRAM OF SIMULATION PLATFORM...49
FIG 3.2OVERVIEW OF RECEIVER DESIGN...50
FIG 3.3STRUCTURE OF INNER RECEIVER...51
FIG 3.4ISI EFFECT ON CFO ACQUISITION...53
FIG 3.52-D INTERPOLATION IN CHANNEL ESTIMATION UNIT DESIGN...54
FIG 3.6CHANNEL MODEL OF DVB-T/H SYSTEM...54
FIG 3.7CHANNEL RESPONSE OF RAYLEIGH AND RICEAN (K=10DB) CHANNEL...57
FIG 3.8DOPPLER FREQUENCY SPREAD MODEL...58
FIG 3.9BER VS SNR AFTER VITERBI DECODER WITH ONE FINE TUNE EVERY 8 SYMBOL...60
FIG 3.10BER VS SNR AFTER VITERBI DECODER WITH ONE FINE TUNE EVERY 64 SYMBOL...61
FIG 3.11TOLERANCE RANGE OF SCO@RAYLEIGH,SNR=34DB, AND DOPPLER 70HZ...62
FIG 3.12BER AFTER VITERBI VS DOPPLER FREQUENCY...63
FIG 3.13 PERFORMANCE OF THE RESIDUAL SCO CONVERGENCE IN CP-LLS ALGORITHM BETWEEN WITH AND WITHOUT FEEDBACK FORWARD ARCHITECTURE...68
FIG 3.14 PERFORMANCE OF THE RESIDUAL SCO CONVERGENCE IN SP-LLS ALGORITHM
BETWEEN WITH AND WITHOUT FEEDBACK FORWARD ARCHITECTURE...68
FIG 3.15 PERFORMANCE OF THE RESIDUAL SCO CONVERGENCE BETWEEN CP-LLS AND SP-LLS ALGORITHM WITH FEEDBACK FORWARD ARCHITECTURE...69
FIG 3.16OVERALL SYSTEM PERFORMANCE IN STATIC GAUSSIAN CHANNEL...70
FIG 3.17OVERALL SYSTEM PERFORMANCE IN STATIC RICEAN CHANNEL...71
FIG 3.18OVERALL SYSTEM PERFORMANCE IN STATIC RAYLEIGH CHANNEL...71
FIG 3.19OVERALL SYSTEM PERFORMANCE IN RAYLEIGH CHANNEL WITH DOPPLER FREQUENCY 70HZ...73
FIG 4.1 OVERALL SYNCHRONIZATION SCHEME...74
FIG 4.2 LOCATIONS OF INTEGRATION SYNCHRONIZATION SYSTEM IN RECEIVER PLATFORM...75
FIG 4.3NORMAL MAXIMUM CORRELATION ALGORITHM FOR GI/MODE DETECTOR...76
FIG 4.5THE PROPOSED GUARD BAND POWER DETECTION BASED APPROACH...79
FIG 4.6 THE ARCHITECTURE OF PROPOSED RESIDUAL CFO TRACKING LOOP WITH SP METHOD..80
FIG 4.7 TIMING DIAGRAM OF PROPOSED RESIDUAL CFO TRACKING LOOP ARCHITECTURE WITH SP METHOD...80
List of Tables
TABLE 1-1PARAMETERS FOR 8MHZ CHANNEL IN DVB-H STANDARD...5
TABLE 1-2PARAMETERS FOR 7MHZ CHANNEL IN DVB-H STANDARD...6
TABLE 1-3PARAMETERS FOR 6MHZ CHANNEL IN DVB-H STANDARD...6
TABLE 1-4PARAMETERS FOR 5MHZ CHANNEL IN DVB-H STANDARD...6
TABLE 1-5 CARRIER INDICES FOR CONTINUAL PILOT CARRIERS FOR 8K MODE...9
TABLE 1-6CARRIER INDICES FOR TPS CARRIERS FOR 8K MODE...10
TABLE 2-1TPS SIGNALING INFORMATION AND CONTENT...35
TABLE 2-2 NUMBER OF RS204 BYTES PACKETS PER OFDM SUPER-FRAME FOR ALL COMBINATIONS OF CODE RATES AND MODULATION FORMS...38
TABLE 2-3 THE PACKETS IN ONE CARRIER WITH VARIED CODE RATE IN QPSK MAPPING...39
TABLE 2-4 THE PACKETS IN ONE CARRIER WITH VARIED CODE RATE IN 16-QAM MAPPING...39
TABLE 2-5 THE PACKETS IN ONE CARRIER WITH VARIED CODE RATE IN 64-QAM MAPPING...39
TABLE 2-5 RESIDUAL CARRIERS FOR RS PACKET SYNCHRONIZATION FOR 2K MODE...40
TABLE 2-6 RESIDUAL CARRIERS FOR RS PACKET SYNCHRONIZATION FOR 4K MODE...40
TABLE 2-7 RESIDUAL CARRIERS FOR RS PACKET SYNCHRONIZATION FOR 8K MODE...41
TABLE 3-1SYNCHRONIZATION TIME (MS) IN 8MHZ CHANNEL...64
TABLE 3-2SYNCHRONIZATION TIME (MS) IN 7MHZ CHANNEL...64
TABLE 3-3SYNCHRONIZATION TIME (MS) IN 6MHZ CHANNEL...64
TABLE 3-4SYNCHRONIZATION TIME (MS) IN 5MHZ CHANNEL...64
TABLE 3-5 THE ESTIMATION ACCURACY OF THE SCO IN PPM...66
TABLE 3-6 CHANNEL MODEL OF THE SIMULATION IN SCO ESTIMATION ALGORITHM...66
TABLE 3-7CHANNEL MODEL OF THE SIMULATION IN SCO TRACKING LOOP ARCHITECTURE...67
TABLE 3-9SYNCHRONIZATION LOSS AND TOTAL SNR LOSS IN MOBILE CHANNEL...73
TABLE 4-1 THE HARDWARE GATE COUNT OF SYNCHRONIZATION FUNCTION IN [1]...81
TABLE 4-2 THE HARDWARE GATE COUNT OF SYNCHRONIZATION FUNCTION FOR THE PROPOSED DESIGN...81
Chapter 1 .
Introduction
1.1
Motivation
Coded Orthogonal Frequency Division Multiplexing (COFDM) technique has developed for a while time to resist mobile channel effect in wireless communication. As European Telecommunication Standard Institute (ETSI) Digital Video Broadcasting-Handheld (DVB-H) system [], the wireless channel environment, like fast frequency selective fading, time variant, Doppler effect, sampling clock offset (SCO) and carrier frequency offset (CFO) is concerned. Under such severe channel effect, the synchronization schemes are especially important in receiver design for COFDM system. In such severe channel, the synchronization is very difficult. However, future DVB-H terminals will most probably make intensive use of a TDM system called “Time-Slicing” [4] to cut power consumption to reasonable number for handheld environment.
In order to fully exploit the potential power reduction, synchronization times of DVB-H receiver must keep a minimum. In this paper, we propose fast synchronization architecture including fast TPS decoder, fast RS header decoder and fast SCO/CFO tracking in the integrated synchronization system.
And then in such severe channel environment, the phase mismatch would happen in post-FFT symbols. If we use the 2-D linear equalization, the problem will make performance degradation much. We need a phase alignment in time axis to solve it. Although the phase mismatch is not caused channel effect but it is the problem followed after synchronizers, fine symbol synchronizer and resampler. Proposed phase alignment in time axis can also be placed
in the proposed integrated synchronization system.
After we solve the synchronization problems in the system, the integrated synchronization will be introduced. The motivation of integration of synchronization system is from this problem that algorithm cores of different synchronization schemes are too similar. Integrating the whole synchronization system and obtaining the most hardware reuse can reduce cost and get the best solution. Fast synchronization architecture, phase alignment and integrated synchronization system can also be integrated into [1], the published paper in ISSCC 2006.
1.2
Introduction to DVB-T/H system
DTV (Digital TV) is popularly used as the next-generation video broadcasting transmission technology in recent years. DTV provides much higher A/V quality and less transmission noise than conventional analog TV. Nowadays, the developed DTV standards consist of DVB (Digital Video Broadcasting) in Europe, ATSC (Advanced Television Systems Committee) in U.S., ISDB (Integrated Services Digital Broadcasting) in Japan and DMB (Digital Multimedia Broadcasting) in China. The transmission modes of DTV include direct satellite broadcasting, cable and terrestrial broadcasting (over-the-air). In terrestrial broadcasting, particularly, video signal is transmitted against severer channel distortions such as multipath fading, co-channel interference and adjacent-channel interference. Since broadcasting transmission system is usually designed to operate within the UHF spectrum allocation for analogue transmissions, it has to provide sufficient protection against high levels of co/adjacent-channel interference emanating from existing PAL (Phase Alternative Line) / SECAM (SEquentiel Couleur Avec Memoire or sequential color with memory) services. Therefore, it is clearly that the terrestrial broadcasting has more challenges in research.
DVB-T standard, one kind of the most popular standards, has been produced by European Telecommunication Standard Institute (ETSI) in Aug, 1997. It has been applied in many countries in the world. In Taiwan, DVB-T standard is also applied as the broadcasting standard. In order to provide the high data rate required for video transmission and resist severe channel distortion in DVB-T, concatenated-coded Orthogonal Frequency Division Multiplexing (COFDM) has been adopted into DVB-T in particular. COFDM is a very popular technology today due to its high data rate transmission capability with high bandwidth efficiency and its robustness to multipath distortion. It has been also chosen as the transmission technique of other communication systems such as ADSL, VDSL, XDSL, DAB and IEEE802.11a/g.
For resisting all kinds of propagation conditions encountered in the wireless broadcasting channel, many parameters of COFDM for DVB-T can be dynamically changed according to channel conditions. The number of COFDM subcarriers can either be 2048 (2K) or 8192 (8K) so that the desired trade-off can be made between inter-symbol-interference (ISI) and Doppler spread. In 2K mode, wider subcarrier spacing can significantly reduce the distortion caused by Doppler frequency spread. In 8K mode, longer OFDM symbol duration can overcome larger multipath fading. Other parameters like guard interval length, constellation mapping mode and coding rate of Viterbi can be also properly decided up to the broadcasting channel condition of the local area. Like guard interval, longer ones have more powerful capability in severe multipath channel than shorter ones. QPSK can resist more noise distortion than 16-QAM and 64-QAM. Less coding rate can detect out more incorrect bit, it is suitable in the severe error channel.
Although the DVB-T reception can also be applied in mobile environment, the ability of reception for handheld terminals is still not good enough because of its high operation power. Therefore, Digital Video Broadcasting-Handheld (DVB-H) was also proposed based on the DVB-T technology to provide broadcast services for handheld devices such as PDAs or mobile phones [6].
receivers, was formally adopted as an ETSI standard in November 2004. This is the offcial DVB-H website maintained by the DVB Project Office. The DVB-H technology is a spin-off of the DVB-T standard. It is large extent compatible to DVB-T but takes into account the specific properties of the addressed terminals- small, lightweight, portable, battery-powered devices in mobile environment. Unlike the DVB-T transport stream adopted from the MPEG2 standard, the DVB-H system is IP (Internet Protocol)-based, therefore the outer DVB-H interface is the IP interface. The IP data are embedded into the transport stream by means of the MPE (Multi Protocol Encapsulation) frame, an adaptation protocol defined in the DVB Data Broadcasting Specification [4]. One MPE frame contains one or more IP datagrams and has a maximum number of 1024 rows and a constant number of 255 columns. The transmission system for DVB-T/H standard is shown in Fig 1.1. It contains the blocks for source coding, outer coding and interleaving, inner coding and interleaving, mapping and modulation, frame adaptation and COFDM transmission. The additional features of DVB-H are also shown in Fig 1.1.
As we can see the DVB-H codec is additional composed of the MPE, MPE-FEC, and time-slicing. Time-slicing architecture will be introduced in section 1.3. For mobile channels reception and long delay spread conditions, an enhanced error protection scheme on the link layer is needed. This scheme is called MPE-FEC and employs powerful channel coding and time interleaving. The MPE-FEC scheme consists of an RS code in conjunction with an extensive block interleaving. The RS (255, 191, 64) code is utilized to perform MPE-FEC error protection. Besides, a virtual block interleaving effect is also performed by reading from and writing to the MPE frame in column direction whereas coding is applied in row direction.
The parameters for 8MHz channel bandwidth in DVB-H standard are listed in Table 1-1. Table 1-1 Parameters for 8MHz channel in DVB-H standard
Parameter 8k mode 4k mode 2k mode
Number of subcarriers K 6817 3409 1705
Value of carrier number Kmin 0 0 0
Value of carrier number Kmax 6816 3408 1704
FFT size N 8192 4096 2048
Symbol duration TU 896μs 448μs 224μs
Subcarrier spacing 1/TU 1.116KHz 2.232KHz 4.464KHz
Spacing between Kmin and Kmax 7.61MHz 7.61MHz 7.61MHz
Guard interval Ng/N 1/4,1/8,1/16,1/32 1/4,1/8,1/16,1/32 1/4,1/8,1/16,1/32
Table 1-2, Table 1-3, and Table 1-4 shows the different parameters for 7MHz, 6MHz, and 5MHz channel in DVB-H standard. The DVB-H also supports 5MHz transmission channel bandwidth in addition.
Table 1-2 Parameters for 7MHz channel in DVB-H standard
Parameter 8k mode 4k mode 2k mode Symbol duration TU 1024μs 512μs 256μs
Subcarrier spacing 1/TU 0.977KHz 1.953KHz 3.906KHz
Spacing between Kmin and Kmax 6.66MHz 6.66MHz 6.66MHz
Table 1-3 Parameters for 6MHz channel in DVB-H standard
Parameter 8k mode 4k mode 2k mode Symbol duration TU 1194.67μs 597.33μs 298.67μs
Subcarrier spacing 1/TU 0.837KHz 1.674KHz 3.348KHz
Spacing between Kmin and Kmax 5.71MHz 5.71MHz 5.71MHz
Table 1-4 Parameters for 5MHz channel in DVB-H standard
Parameter 8k mode 4k mode 2k mode Symbol duration TU 1433.60μs 716.80μs 358.40μs
Subcarrier spacing 1/TU 0.697KHz 1.395KHz 2.790KHz
Spacing between Kmin and Kmax 4.75MHz 4.75MHz 4.75MHz
In the case of two-level hierarchy, the functional block diagram of the system must be expanded to include the modules shown in dashed in. The splitter separates the incoming data stream into the high-priority and the low-priority stream. These two bitstreams are mapped onto the signal constellation by the mapping and therefore the modulator has a corresponding. This system uses COFDM transmission. All data carriers in one COFDM symbol are mapped either as QPSK, 16-QAM, 64-QAM, non-uniform-16-QAM or non-uniform-64-QAM. In
addition to the transmitted data, a COFDM symbol contains scattered pilots, continual pilots and TPS (Transmission Parameter Signaling) pilots. These reference signals can be used for synchronization, channel estimation and transmission mode verification. The COFDM frame consists of 68 COFDM symbols and four frames constitute one super-frame. The frame structure involving distribution of scattered pilots is shown in Fig 1.2. Scattered pilots insert every 12 subcarriers and have an interval of 3 subcarriers in the next adjacent symbol.
Fig 1.2 frame structure
The carrier indices of scatter pilots are shown as
min min max
{ 3 ( mod 4) 12 | integer, 0, [ ; ] }
SP= k=K + × l + p p p≥ k∈ K K (1-1) The corresponding modulation of scattered pilots is expressed as
m,l,k k m,l,k Re{c } = 4 / 3 2 (1/2 - w ) Im{c } = 0 × (1-2)
where wk means Pseudo Random Binary Sequence (PRBS), and cm,l,k means k-th subcarrier in
l-th symbol in m-th frame. PRBS sequence (X11+X2+1) determines the values of scattered pilots, continual pilots and TPS pilots. The PRBS generator is shown as in Fig 1.3.
Fig 1.3 Generation of PRBS sequence
Continual pilots locate at fixed indices of subcarrier, which contains 177 pilots in 8K mode, 89 pilots in 4K mode, and 45 pilots in 2K mode. The corresponding modulation is expressed as m,l,k k m,l,k Re{c } = 4 / 3 2 (1/2 - w ) Im{c } = 0 × (1-3)
The subcarrier indices of continual pilots in 8K mode are shown in Table 1-5. The carrier indices of pilots in 2K mode are 0 to 1704 in Table 1-5, and that in 4K mode are 0 to 3408.
Table 1-5 carrier indices for continual pilot carriers for 8K mode 0 48 54 87 141 156 192 201 255 279 282 333 432 450 483 525 531 618 636 714 759 765 780 804 873 888 918 939 942 969 984 1050 1101 1107 1110 1137 1140 1146 1206 1269 1323 1377 1491 1683 1704 1752 1758 1791 1845 1860 1896 1905 1959 1983 1986 2037 2136 2154 2187 2229 2235 2322 2340 2418 2463 2469 2484 2508 2577 2592 2622 2643 2646 2673 2688 2754 2805 2811 2814 2841 2844 2850 2910 2973 3027 3081 3195 3387 3408 3456 3462 3495 3549 3564 3600 3609 3663 3687 3690 3741 3840 3858 3891 3933 3939 4026 4044 4122 4167 4173 4188 4212 4281 4296 4326 4347 4350 4377 4392 4458 4509 4515 4518 4545 4548 4554 4614 4677 4731 4785 4899 5091 5112 5160 5166 5199 5253 5268 5304 5313 5367 5391 5394 5445 5544 5562 5595 5637 5643 5730 5748 5826 5871 5877 5892 5916 5985 6000 6030 6051 6054 6081 6096 6162 6213 6219 6222 6249 6252 6258 6318 6381 6435 6489 6603 6795 6816
Both scattered pilots and continual pilots are transmitted at a boosted power level of 16/9 whereas the power level of other symbols is normalized to 1.
The TPS carriers are used for the purpose of signaling parameters related to the transmission scheme, i.e. to channel coding and modulation. The TPS is defined over 68 consecutive OFDM symbol and transmitted in parallel on 17 TPS carriers for the 2K mode and on 68 carriers for the 8K mode. Each OFDM symbol conveys one TPS bit which is differentially encoded in every TPS carriers. The TPS information contains frame number, constellation, hierarchy, code rate, guard interval, transmission mode and BCH error protection code. There is 17-bit word as synchronization word in one frame. Unlike scattered and continual pilots, TPS pilots are transmitted at the normal power level of 1 with DBPSK modulation. The modulation is expressed as
m,l,k m,l-1,k m,l,k m,l,k m,l-1,k m,l,k
if sl = 0, then Re{c } = Re{c }; Im{c } = 0;
if sl = 1, then Re{c } = -Re{c }; Im{c } = 0. (1-4)
The absolute modulation of the TPS carriers in the first symbol in a frame is derived from the reference sequence wk as follows:
m,l,k k m,l,k
Re{c } = 2 (1/2 - w )
Im{c } = 0 (1-5)
The carrier indices for TPS carriers in 8k mode are listed in Table 1-6, and that is in 2k mode and 4k mode are from 0 to 1687 and from 0 to 3048. It concludes 17, 34 and 68 TPS carriers in 2K, 4K and 8K mode respectively.
Table 1-6 Carrier indices for TPS carriers for 8K mode
34 50 209 346 413 569 595 688 790 901 1073 1219 1262 1286 1469 1594 1687 1738 1754 1913 2050 2117 2273 2299 2392 2494 2605 2777 2923 2966 2990 3173 3298 3391 3442 3458 3617 3754 3821 3977 4003 4096 4198 4309 4481 4627 4670 4694 4877 5002 5095 5146 5162 5321 5458 5525 5681 5707 5800 5902 6013 6185 6331 6374 6398 6581 6706 6799
The guard interval may have four values, i.e. 1/4, 1/8, 1/16 and 1/32. Guard interval 1/4 would occupy 25% of the usable transmission capacity and hence only be used in case of SFN operation with long distances between transmitter sites. In the case of smaller transmitter distances (local SFN) or non-SFN operation the smaller values of guard interval can be selected. In conclusion, DVB-T/H system has good flexibility for various transmission conditions, so that it becomes a successful technology for video broadcasting.
1.3
Introduction to Time-Slicing Technology
In order to satisfy the low power issue in battery-powered terminals, a time-multiplexed transmission of different service is exploited. This technique, called time slicing, allows for selective access to desired data and results in a large battery power saving effect. The burst duration of time slicing is in the range of several hundred ms whereas the off-time may amount to several seconds. The lead time for power-on and resynchronization is assumed to be less than 250ms. Depending on the duty/turn-off ratio, the resulting power saving may be more than 90%. Timing-Slicing technology in DVB-H system [4], as Fig.1.4, provides a low power consumption methodology. The saving of power consumption depends on each kind of parameter in Time Slice identifier descriptor in DVB-H data [4]. The main spirit of Time-Slicing technology is to utilize the unused bandwidth. Because that the resolution of DVB-H system in handheld device is smaller than household system, the residual bandwidth is waste. Hence, stack the data in burst time can reduce the computing power and the bandwidth will not waste. And the cost of Time-Slicing architecture is to store the symbol data in buffers.
Bu
rs
t b
an
dw
id
th
Constant bandwidth
Burst duration
Burst size
Off-time
Bu
rs
t b
an
dw
id
th
Constant bandwidth
Burst duration
Burst size
Off-time
Synchronization time
The related formulas of power saving are listed below. Bd Burst Duration (seconds) Bs Burst Size (bits)
Bb Burst Bandwidth (bits per second) Cb Constant Bandwidth (bits per second) Ot Off-time (seconds)
St Synchronization Time (seconds) Ps Power Saving (per cent)
Dj Delta-t Jitter (seconds)
Bs Bd = Bb 0.96 Bs Ot = Bd Cb 0.96 (Bd+St+(3/4 Dj)) Cb 0.96 Ps = (1- ) 100% Bs × − × × × × × (1-6)
where most of burst parameters are shown in Fig. 1.3 and delta-t means the period of on-off (Bd+Ot) . These parameters are decided from identifier descriptor in MAC layer in DVB-H system [4]. And we list several key parameters, identifier descriptors, relative to power consumption in Time-Slicing technology below.
Burst Duration (variable of 8-bit number) = (8-bit number) × 20 (ms) (1-7) Burst Size (fixed):
128kbits; 256kbits; 384kbits; 512kbits; 640kbits; 768kbits; 896kbits; 1024kbits; 1152kbits; 1280kbits; 1408kbits; 1536kbits; 1664kbits; 1792kbits; 1920kbits; 2048kbits
Constant Bandwidth (fixed):
16 kbps; 32 kbps; 64 kbps; 128 kbps; 256 kbps; 512 kbps; 1024 kbps; 2048 kbps
Base on the above mentioned equations and identifier descriptors, we can attempt to calculate the saving ratio of power consumption. For instance, in 8k mode, guard interval 1/4, Bd=260ms, St=240ms, Dj=10ms, Cb=256kbps and Bs=1280kbits, the saving of power consumption reaches 90% than DVB-T system. Another example, in 2k mode guard interval 1/32, Bd=120ms, St=160ms, Dj=10ms, Cb=512kbps, Bs=1024kbits, the saving of power
consumption reaches 86% than DVB-T.
However, the cost of saving power is the buffers in receiver. In Fig 1.4, it illustrates a simplified model for the Time Slicing/MPE-FEC buffer which is used in the Receiver to store the time slicing burst and to offer constant bit stream for streaming services during off time. The data is received at the rate of Bb and the leakage rate, i.e. the rate at the output of the
buffer, is Rout. The buffer has a certain size and when the data is written into the buffer, there
is a certain processing delay (including, e.g. MPE-FEC decoding time) before the data can be read out. For each elementary stream, the maximum average bit rate over one time slicing cycle denoted by Cb is signalled in the time_slice_fec_identifier descriptor, and it is defined
as: s s b d t c B B C B O T = = + (1-8) where Bs and Bd are the size and the duration of the burst, respectively, Ot is the off-time
between the bursts, and Tc is the cycle time for the burst ( Bd + Ot ). All parameters are here
defined with respect to layer 3 datagrams. By knowing Cb and its own processing time, the
Receiver can, for instance, check if the leakage rate Rout is high enough to successfully
receive the particular elementary stream.
Fig 1.5 The Time Slicing/MPE-FEC buffer in the Receiver
1.4
Organization of This Thesis
This thesis is organized as follows. In chapter 2, the signal models and the detailed algorithms of the proposed CFO synchronization scheme will be introduced. The simulation
result and performance analysis will be discussed in chapter 3. Chapter 4 will introduce the design methodology, hardware architecture, and the chip summary of the proposed design. Conclusion and future work will be given in chapter 5.
Chapter 2 .
Synchronization Algorithms
2.1
Introduction to Unsynchronous Problems
In OFDM system, unsynchronous problems can be divided into two parts: timing and frequency. Timing unsynchronous problems concludes sampling clock offset (SCO) and symbol timing offset. The symbol timing offset occurs when symbol synchronization finds incorrect OFDM symbol boundary, and sampling clock offset is caused by the difference between the sampling frequencies of the digital-to-analog converter (DAC) and the one of the analog-to-digital converter (ADC). Sampling clock offset can also lead to symbol timing drift. Unlike other packet based communication system such as 802.11a, DVB-T system is a continuous-data transmission. Therefore, sampling clock offset is a critical problem to be solved.
Frequency unsynchronous problems are carrier frequency offset (CFO) and phase mismatch in time axis. Phase mismatch problem is happened when symbol timing offset changed in the system with 2-D linear interpolation in channel estimation. The phase rotation difference in the same carrier index between adjacent symbols causes interpolation error in time axis is called phase mismatch. And in OFDM system, the spectrum of the individual subcarrier mutually overlaps and exhibits orthogonality to achieve optimum spectrum efficiency. However, CFO would make inter carrier interference (ICI). CFO is also introduced by the mismatch of oscillator frequency between transmitter and receiver. Once CFO exists, the orthogonality between subcarriers will be destroyed and the degradation of the system performance will be serious. Compared with other OFDM based system such as IEEE 802.11a, the subcarrier space of DVB-T/H system is relatively narrower and the tolerance of carrier frequency offset is also worse [3][7]. Hence the CFO synchronization is a very critical
problem to be solved in DVB-T/H system.
2.1.1 Effect of Carrier Frequency Offset and Sampling Clock Offset
Consider an OFDM system using an inverse fast Fourier transform (IFFT) of size N for modulation. Each OFDM symbol is composed of K (K<N) data subcarriers al,k, where l
denotes the OFDM symbol index and k (0 ≤ k < K) denotes the subcarrier index. After IFFT, a
cyclic prefix composed of Ng samples is inserted to avoid the influence of multipath channel
delay spread. So a transmitted symbol has Ns =N+ Ng samples with sample period T. The
transmitted complex baseband signal of the l-th symbol can be expressed as
, 2 '( ( ) ) 1 2 , 0 1 ( ) g s c tx j k t N l N T K j tf NT l l k k s t e a e N π π − − + ⋅ = ⎧ ⎫ ⎪ ⎪ = ⎨ ⋅ ⎬ ⎪ ⎪ ⎩
∑
⎭ (2-1) where fc tx, is the central frequency of the transmitter RF oscillator, and k' is the subcarrier index relative to the centre frequency, k'= −k (K−1) / 2.Since the CFO Δ (f Δ =f fc tx, − fc rx, ) between transmitter and receiver RF oscillator can be expressed as a time-variant phase error, ej2πΔft, the l-th received symbol after sampling
with period T’ at time instants tn=(lNs+Ng+n)T’ and removing guard interval can be expressed
as ( ) j2 ftn ( ) ( , ) ( ) l l n n l r n =e πΔ ⋅s t ∗h t τ +w n 2 '( ( ) ) 1 2 , 0 1 ( , ) ( ) n g s n j k t N l N T K j ft NT l k n l k e a e h t w n N π πΔ − − + ⋅ τ = = ⋅
∑
⋅ ∗ + (2-2) where ( , )h tn τ is the channel impulse response with delay spread τ , ( )w n is the lcomplex-valued additive white Gaussian noise (AWGN).
After demodulation via a fast Fourier transform (FFT), the l-th OFDM symbol at subcarrier k, R is as follows l k,
1 2 ' / , 0 ( ) N j k n N l k l n R − r n e− π = =
∑
2 ' ( ) 2 ( )(1 ) / , , , , s g s g k j lN N j lN N N N l k l k l k l k e e H a ICI W π ζ πε + +ζ + α = ⋅ ⋅ ⋅ + + (2-3) where ε = ΔfNT is the CFO value normalized with the subcarrier space, ζ is the sampling clock offset (SCO) (ζ =( 'T T T− ) / ), α is an attenuation factor which is close to 1, and,
l k
ICI is the inter-carrier interference noise due to carrier frequency offset. Likewise, H is l k,
the channel frequency response on the k-th subcarrier of the l-th OFDM symbol with the assumption that the channel is stationary within at last one symbol, W is a zero-mean l k,
stationary complex process as well.
As previous section shows, CFO introduces various imperfect effects to the received signal. From the viewpoint of time domain, the CFO can be expressed as a time-variant phase error. The rotated phase error is in proportion to the received sample time instants tn and can
be expressed as
( ) 2 2 ( ) /
l n ftn lNs Ng n N
θ = Δ =π πε + + (2-4) where θ is the phase rotation caused by CFO. Unlike other packet-based communication systems such as IEEE 802.11a, DVB-T/H is a continuous-data transmission system and the receiving of data continues until the receiver is turned off. So the phase error will still be large even in very weak CFO environment when the receiver operates for a long time as shown in Fig 2.1.
0 2 4 6 8 10 x 105 -4 -3 -2 -1 0 1 2 3 4 sample index phase r otati on
Fig 2.1 Phase rotation in time domain for long time reception when ε=0.01
-4 -3 -2 -1 0 1 2 3 4 -0.4 -0.2 0 0.2 0.4 0.6 0.8 1 f (k) -0.4-4 -3 -2 -1 0 1 2 3 4 -0.2 0 0.2 0.4 0.6 0.8 1 f(k) (a) ε=0.1 (b) ε=1.1
Fig 2.2 Spectrum of five subacrriers in carrier frequency offset environment
CFO results in different effects in frequency domain. It not only reduces the amplitude but also shifts the phase of the demodulated signal. Further more, the second term of (2-3)
,
l k
ICI degrades the system performance strongly because it destroys the orthogonality within
each subcarrier in OFDM symbols, and can be expressed as
~ ~ ~ 1 ( 1) / ( ) / , , , ~ 0 sin( ) sin( ( ) / ) K j N N j k k N l k l k l k k k k ICI H a e e N k k N πε π πε π ε − − − − = ≠ = ⋅ − +
∑
. (2-5)expressed ass
I F
ε ε= +ε (2-6) From Fig. 2.2, we can find that CFO causes inter-carrier interference noise within each subcarrier and makes the orthogonality of spectrum lost. Once the integral part of CFO εI is not zero, all of the subcarriers will shift circularly. The shift of subcarrier index will make the channel estimator receive wrong pilot sequence at the pre-defined pilot index and then the calculated channel frequency response will be not reliable. Also the TPS decoder can not receive correct TPS pattern to decode the correct system parameter. All of these imperfect effects in different domain should be corrected by the aid of CFO synchronization to obtain good receiving performance.
The difference of rotatedphases between two adjacent symbols is represented as:
1 ' ( ) ( ) ( ) 2 2 2 2 2 l l l s s s s s k k k N k fN T fN T N N k fN T N ϕ ϕ ϕ π ζ π π ζ π ζ π − = − = Δ + Δ + ≈ Δ + (2-7)
We can ignore the term 2πΔfNTζ, since the SCO is usually less than 1.0x10-4. As (2-7) indicates, CFO causes mean phase error as well as SCO causes linear phase error between two adjacent symbols.
Fig 2.3 Phase rotation due to timing drifts
In Fig 2.3, it demonstrates the phase rotation of timing drift due to sampling clock offset. In the former symbols, the total amount of phase rotation is limited in 2π (rads) since the drift point is less than one sample. After symbol timing drift exceeding one sample, phase rotation becomes severer increasingly. Regardless of the case of symbol timing drifting into ISI region, the violent phase variation still reduce the performance of channel estimation. If symbol timing drifts out of ISI-free region, inter-symbol interference is produced and hence system performance degrades much.
2.1.2 Effect of Symbol Timing Offset
The symbol synchronization of the OFDM system is to find the start of OFDM symbol, i.e. the FFT window position. Just as what is shown in Fig 2.4, we call Δ the ISI-free region. If the estimated start position of OFDM symbol is located within the ISI-free region, data will not be affected by inter-symbol interference (ISI). The effect of phase rotation caused by symbol timing offset can be easily corrected after FFT.
Fig 2.4 ISI-free region
Assume x(n) represents received data in time domain, X(k) is subcarrier after FFT operation
for x(n) with perfect symbol timing, and ˆ ( )X k denotes subcarrier after FFT operation with
symbol timing offset ε in the ISI-free region. The detail equations are demonstrated as follows. 1 2 0 1 ^ 2 0 1 ^ 2 2 0 ^ 2 / ( ) ( ) ( ) ( ) ( ) ( ) ( ) ( ) n N i k N n n N i k N n n N i k i k N N n i k N X k x n e X k x n e X k x n e e X k X k e π ε π ε π π π ε − − = + − − = − − − = = = = =
∑
∑
∑
(2-8)where k represents the subcarrier index, n denotes sample index in time domain, and N is the number of subcarriers in an OFDM symbol. Note the last term ei2πkε/N in (2-8), which exhibits the phase rotation. Therefore, we can conclude that the effect of symbol timing offset in the ISI-free region is phase rotation and unchanged magnitude of subcarrier, which can be compensated by equalizer completely. The phase rotation effect is shown in Fig 2.5. Fig 2.5(b) depicts the condition of symbol timing offset ε = 2 while Fig 2.5(c) shows the condition of ε = 5. As symbol timing offset ε is lager, the phase variation is severer. The additional variance of channel response due to timing error will increase the difficulty of channel estimation. In order to
ease the load of channel estimation unit, the symbol timing effect should be as small as possible even the phase rotate effect can be completely corrected in theory.
(a) Symbol timing offset in the ISI -free region
(b) Phase rotation due to symbol timing offset=2
(c) Phase rotation due to symbol timing offset=5
Fig 2.5 Phase rotation of symbol timing offset ε = 2 and ε = 5
On the other hand, if the estimated start position locates out of ISI-free region, the sampled OFDM symbol will contains some samples that belong to previous symbol or following symbol, which leads to the dispersion of signal constellation (ISI) and reduce system performance much. Therefore, the objective of symbol synchronization, first of all, is to avoid the estimated symbol boundary lying in ISI region and subsequently reduce the
symbol timing offset as far as possible. The relative mapping constellations are depicted in Fig 2.6. Fig 2.6(a) shows the phase rotation effect due to symbol timing offset of 5 samples while Fig 2.6(b)shows the ISI effect which destroys the signal constellation heavily.
(a) Symbol offset 5 samples in the ISI-free region
(b) Symbol offset 5 samples in the ISI region
Fig 2.6 Mapping constellation
2.1.3 Effect of Carrier Phase Mismatch
The problem of carrier phase mismatch comes from the different phase rotation of post-FFT symbols by using 2-D channel equalization (2-D CE). To observe the carrier phase mismatch problem, we should discuss from fine symbol synchronization, SCO tracking and 2-D channel interpolation.
First, we discuss fine symbol synchronization. Fine symbol synchronization scheme in DVB-T/H system is the function of precisely monitoring the time symbol boundary to prevent ISI occurring in serious frequency selective fading channel. If we don’t use fine symbol synchronization, the carrier phase rotation would be too large and the phase between pilots to pilots would make the interpolation error, as shown in Fig 2.7. The performance would decrease very seriously.
Re Im ˆ p θ H kp( ) ( 1) p H k+
{
H k( )}
{ }
H kˆ ( )Fig 2.7 the influence of frame position error to interpolations
In our simulation, we choose IFFT method to be the algorithm of fine symbol synchronization. IFFT based method is the most popular algorithm in DVB-T/H system as shown in [1], [2]. Observing the channel impulse response (CIR) in time domain we can obtain the residual symbol timing offset information as shown in Fig 2.8.
N
N-2
N-5
Channel Impulse Response FFT Window
Cyclic prefix
As we know, fine symbol synchronization of IFFT based method uses the channel frequency response (CFR) of pilots in frequency domain. It goes through IFFT operation and then gets CIR. The source of CFR is calculated from 2-D linear interpolation as shown in Fig 2.9.
Frequency axis
Time axis
Fig 2.9 Interpolation of 2D channel estimation
2-D channel interpolation [8], usually used to resist severe multi-path and mobile environment, includes time and frequency axis interpolations. The first step is to store the previous 3 post-FFT symbols and use the pilot information producing the time axis interpolation. Then, in frequency axis it interpolates the total CFR Hl,k of previous 3-th
post-FFT symbol. Before the frequency axis interpolation and after time axis interpolation, we take the interpolated (K/3) pilots to IFFT for obtaining CIR. And we must pad zeros to (N/2). The decision equation express as
{
max}
2 2 ˆ min | /16, 0,...,( 1) / 3 1 3 ˆ | | x x x n S S x K S h δ = > = − − = (2-9)where hx is CIR of pilots, x is its index and M is K/3.
After fine symbol updating, it remains phase rotation between pilots’ CFR, as shown in Fig 2.10. And the previous and current post-FFT symbols have different phase rotation. In 2-D linear interpolation algorithm, the previous symbols and pilots are stored in memory in order to time axis interpolation. At the same subcarrier index between previous and current
symbols, the different phase rotation makes phase mismatch. It would cause the performance degradation. The carrier phase mismatch is shown as
2 / 2 / 2 ( ) / 2 / ˆ ( ) ( ) ˆ ( ) ( ) ˆ ( ) ( ) ( ) pre cur cur pre i k N p pre p i k N p cur p i k N i k N H k H k e H k H k e H k H k e H k e π ε π ε π ε ε π ε − − − Δ = = = = (2-10)
where Hp-pre(k) and Hp-cur(k) respectively means CFR of k-th pilot in previous symbols and
current symbol in f-domain. H(k) means CFR of k-th subcarrier in f-domain symbol. Ĥ(k) means the estimated value.
Fig 2.10 the phase rotation of pilots’ CFR caused by different symbol timing offset
When the multi-path fading spread changes seriously, the channel impulse response would also changes, as shown in Fig 2.11. This situation is obvious in SCO effect in mobile environment. In order to simulate the influence of SCO in carrier phase mismatch, we plus one term to modify the equation and is expressed as
2 ( ) / 2 /
ˆ ( ) ( ) i k cur pre N ( ) i k N
N-5
Cyclic prefix
Root mean square delay spread
Fig 2.11 Relative channel impulse response with multi-path
Summary of this section, we understand the carrier phase mismatch in time axis is happened in a specific situation. It is in mobile environment with SCO effect and 2-D CE must be used in receiver platform. However, 2-D CE is the most common architecture in state-of-the-art and SCO effect in mobile environment is the significant issue for synchronization in DVB-H system. Thus, the problem of carrier phase mismatch has to be solved.
2.2
Carrier Phase Alignment
Phase alignment locates after fine symbol synchronization scheme and passes information to channel estimator and equalizer, as shown in Fig 2.12. In resent research, carrier phase rotation problem caused by residual symbol timing offset is brought up to discuss. Large phase rotation will produce significant errors in linear channel interpolation, as mentioned in Section 2.1.4. To solve this problem, in [5], it provides a phase compensation method. It can deal with the interpolation error by compensating the phase rotation of pilots and symbols. And the compensating equation is expressed as
ˆ
ˆ ( ) ( ) j kp
p p
Fig 2.12 Location of phase alignment in receiver platform
where ˆ
p
θ is detected out from channel frequency response of post-FFT symbols. Note that it doesn’t move the symbol boundary and it uses phase compensator to replace fine symbol scheme. It will cause a problem in fast frequency selective channel when the system is without fine symbol synchronization, the samples drift. Also, we show the comparison of performance between 2-D CE and 1-D CE in Fig 2.13.
Fig 2.13 Comparison of performance between 2-D CE and 1-D CE
However, two problems remain in [5]. The system of [5] is not scattered pilot based. 2-D CE is not discussed in [5] but the carrier phase mismatch is happened with 2-D CE. Moreover, in [5], the influence of SCO with Doppler effects is not considered. However, these two conditions are significant issues for synchronization in DVB-H system.
The proposed carrier phase alignment in time axis is to align the phase rotation of previous post-FFT symbols in 2-D interpolator buffers. The block diagram is shown in Fig 2.14.
Fig 2.14 the phase rotation of pilots’ CFR caused by different symbol timing offset
Frequency axis
Time ax
is
Stored in Memory
Fig 2.15 2-D channel interpolation
In Fig 2.15, it shows 2-D CE. Interpolation in time axis acts before in frequency axis. The proposed phase alignment in time axis is to align the phase rotation of previous post-FFT symbols in 2-D interpolator buffers.
First step of carrier phase alignment, we take out the CFR of post-FFT symbol pilots by interpolation in time axis. Then, CFR is passed through fine symbol synchronization and we get the updating symbol bound. When the symbol boundary updated, the current symbol is passed to frequency domain and cause the phase mismatch to previous symbols in buffers.
axis will multiply the alignment factor in 3 parts, one is pilots before 3 previous symbols and the other parts are the estimated pilots and the symbol data before divider. Here the equations are expressed as 2 / 2 / 2 / ˆ ( ) ( ) ( ) pre cur i k N i k N p align p i k N p H k H k e e H k e π ε π ε π ε Δ − = × = (2-13) where Δε means the difference timing symbol offset εcur – εpre – εΔδ. εΔδ means the error terms
at resampler out. After aligning the pilots in time axis, we interpolate the CFR of pilots in every 3 subcarriers in time axis, it can be expressed as
, , ,
ˆ ( ) (1 ) ( ) ( )
4 4
p align l p align l i p align l i
i i
H − k = − ×H − − k + ×H − + k , i=1, 2,3 (2-14) Then interpolating the frequency axis subcarrier CFR to get total CFR in each subcarrier, it is expressed as
, ,
ˆ ( ) (1 ) ( ) ( )
4 4
align p align l p align l
j j
H k = − ×H − k− + ×j H − k+ j , Kmin ≤ ≤k Kmax, i=1, 2,3, (2-15) Before subcarrier data are sent into divider, we align the phase as CFR of carriers. And the equations are expressed as
2 / 2 / 2 / ˆ ( ) ( ) ( ) pre cur i k N i k N align i k N R k R k e e R k e π ε π ε π ε Δ = × = (2-16) ˆ ( ) ( ) ˆ ( ) ˆ ( ) ( ) align align align R k R k Y k H k H k = = (2-17)
S-1(20K Bytes) S-2 (20K Bytes) S-3 (20K Bytes) Input Pilots of S-4~S-6(6.8K Bytes) Z-3 x1 x2 x3 x1 x2 x3 + x1 x2 x1 x2 + DIV Output R_sh2 R_sh2 R_sh1 Pilots of S0 pilots Data of S-3 data & Pilots of S0 Z-2 12 12 2D Linear Interpolator Z-1 =
×
e
−i2π εkΔ /NFig 2.16 Proposed time axis phase alignment architecture
After multiplying phase alignment factor e-i2πΔε⁄N, the CFR of pilots and symbol data are both on the same phase criterion. And the phase alignment is completed.
2.3
Fast Synchronization System
For the reason of power saving, Time-Slicing methodology in DVB-H system is used. The saving of power consumption is in (1-6), where Bd means burst duration, St means synchronization time, Dj means delta-t jitter Cb means constant bandwidth, Bs means burst size and Ps means power saving. In this equation, Bd, Cb and Bs are defined in the DVB-H data, [4]. Hence, note that the synchronization time dominates the saving of power consumption in fixed mode. In Fig 2.17, it shows the composition of completed synchronization time. According to Fig 2.17, we can get the information as
St = max{(Ta + Ttps + Trs); (Ta + Tt)} (2-18) where St means synchronization time, Ta means acquisition time, Ttps means transimission parameter signaling (TPS) decode time, Trs is Reed-Solomon (RS) packet synchronization time and Tt means the SCO/CFO tracking time.
Burst Duration
Off-time
Synchronization Time
RS packet
sync.
Acquisition
Time
TPS decode
SCO/CFO tracking
Fig 2.17 composition of completed synchronization time
The determination of synchronization time depends on SCO/CFO tracking is longer or the summation of TPS decoded time and RS packet synchronization time is longer. SCO/CFO tracking time depends on channel effects. However, the TPS decoder and RS packet synchronization needs a fixed time to receive enough symbols. Hence, in this section, we discuss the TPS decoded and RS packet synchronization. In Fig 2.18, it shows the location of these two designs in the receiver platform.
Fig 2.18 Location of fast synchronization in receiver platform
Acquisition time is about 10 symbols, TPS decoded time is about 2 frames (136 symbols), RS sync. Time is about 4/2/1 frames (272/136/68 symbols) and the total synchronization time in costs 417/281/213 symbols in 8k/4k/2k mode in conventional method. And we can see the symbol duration time in this equation:
symbol duration=N T (1+GI) × × (2-19) where N is 2k/4k/8k, T is elementary period 7/64, 1/8, 7/84 and 7/40 μs in 8MHz, 7MHz, 6MHz and 5MHz channel and GI is 1/4, 1/8, 1/16 and 1/32.
2.3.1 TPS Decode
The structure of TPS listed in DVB-T/H standard [3] is composite of 68-bit word (one frame) and concludes 16-bit synchronization word and 1 initialization bit. The transmission parameter information shall be transmitted as shown in table 2-1. Here, one symbol includes one bit TPS and one frame includes 68 bits TPS information.
The mapping of each of the transmission parameters: constellation characteristics, α value, code rate(s), super-frame indicator and guard interval onto the bit combinations is performed according to clauses appendix A. The left most bit is sent first.
Table 2-1 TPS signaling information and content Bit number Purpose/Content
s0 Initialization s1 to s16 Synchronization word s17 to s22 Length indicator s23, s24 Frame number s25, s26 Constellation s27, s28, s29 Hierarchy information
s30, s31, s32 Code rate, HP stream
s33, s34, s35 Code rate, LP stream
s36, s37 Guard interval
s38, s39 Transmission mode
s40 to s47 Cell identifier
s48, s49 DVB-H signaling
s50 to s53 Reserved for future use
s54 to s67 Error protection
The first bit, s0, is an initialization bit for the differential 2-PSK modulation. The modulation
of the TPS initialization bit is derived from the PRBS sequence defined in [3]. Bit 1 to 16 of the TPS is a synchronization word. The first and third TPS block in each super-frame haves the following synchronization word:
s1 - s16 = 0011010111101110.
The second and fourth TPS blocks have the following synchronization word: s1 - s16 = 1100101000010001.
The conventional design [1] of TPS decoded costs 135 (68 + 67) symbols time in the worst situation to receive the completed TPS word and then synchronizes the header. This is the primitive method, that it uses more than one frame to find the synchronization word and initialization bit and then decodes TPS. The conventional operation of TPS word distributes in Fig 2.19.
Fig 2.19 the operation of conventional TPS word
The proposed method can decode out TPS faster than conventional method. The proposed method is to buffer the previous 68-bit TPS word. When finding the 16-bit synchronization word and initialization bit, we read the previous 51 words information as TPS decoded word. Hence, we can advance the time of 51 symbols. The worst case of the decoded time is 84 (67+17) symbols. We can see the operation of proposed TPS word distribution in Fig 2.20.
16
51
16
51
Worse case
searching time
84-bit
Synchronization
word
Decoder
buffer
Initialization
bit
Fig 2.20 the operation of proposed TPS word
2.3.2 RS Packet Synchronization
As Fig 2.21 shown, we know the outer decoder is RS decoder. Because that the input of RS decoder must be a completed packet. In other word, we must synchronize the header of the RS packet. The impact of RS packet is shown as Fig. 8. If we don’t consider the header and the completed packet for the RS decoder input, the decoder must waste a lot of time for waiting synchronization word.
Fig 2.21 RS packet structure
The worst condition of RS packet synchronization time in conventional method is 4/2/1 frame (272/136/68 symbols) in 2k/4k/8k mode. The conventional method is to synchronize the packet in completed frames, but the proposed method just needs to calculate the carrier index to synchronize the packet in the completed carriers. The proposed method will help us to reduce the synchronization time a lot. We can reduce the synchronization time to 5/3/2
symbols in 2k/4k/8k modes.
To introduce the conventional design, we discuss how many RS packets in one super-frame (4 frames). It shows the RS packets in one super-frame in Table. 2. It is accomplished by the formula:
(mapping bits) (data carriers) (symbols) (code rate) (packets/super-frame)=
204 8
× × ×
× (2-20)
For instance, in QPSK mapping, 2k mode and code rate 7/8, we can calculate the RS packets in a super-frame as 2 1512 68 4 7 / 8 441
204 8 packets
× × × × =
× . According to Table 2-2 we can know one super-frame concludes how many packets in all kinds of transmission mode, mappings and code rates. The conventional design of RS packet of synchronization in [1] directly synchronizes in 4/2/1 frame (272/136/68 symbols) in 2k/4k/8k mode. Because that the RS packets in 4k/8k mode can be divided by 2/4 with no remainder in all kinds of mappings and code rates. Hence, RS packets synchronization in 4k/8k mode can be synchronized in 2/1 frame. The worst RS packet synchronization time is 272/136/68 symbols in 2k/4k/8k mode in conventional method.
Table 2-2 number of RS 204 bytes packets per OFDM super-frame for all combinations of code rates and modulation forms
QPSK 16-QAM 64-QAM Code rate 2k 4k 8k 2k 4k 8k 2k 4k 8k 1/2 2/3 3/4 5/6 7/8 252 336 378 420 441 504 672 756 840 882 1008 1344 1512 1680 1764 504 672 768 840 882 1008 1344 1512 1680 1764 2016 2688 3024 3360 3528 756 1108 1134 1260 1323 1512 2016 2268 2520 2646 3024 4032 4536 5040 5292
including packets. Before finding how many subcarriers constituting with completed packets, we calculate the how many packets in a subcarrier. There is a formula to calculate that how many packets are included in a subcarrier:
(code rate) (mapping) (packets/subcarrier)=
(204 8) ×
× (2-21) The solution of this equation is listed in Table 2-3, Table 2-4, and Table 2-5. The denominator of packets/subcarrier is the answer of synchronization subcarriers.
Table 2-3 the packets in one carrier with varied code rate in QPSK mapping Code rate 1/2 2/3 3/4 5/6 7/8
Packets/carrier 1/1632 1/1224 1/1088 5/4896 7/6528 Sync carrier 1632 1224 1088 4896 6528
Sync packet 1 1 1 5 7
Table 2-4 the packets in one carrier with varied code rate in 16-QAM mapping Code rate 1/2 2/3 3/4 5/6 7/8
Packets/carrier 1/816 1/612 1/544 5/2448 7/3264 Sync carrier 816 612 544 2448 3264
Sync packet 1 1 1 5 7
Table 2-5 the packets in one carrier with varied code rate in 64-QAM mapping Code rate 1/2 2/3 3/4 5/6 7/8
Packets/carrier 1/544 1/408 3/1088 5/1632 7/2176 Sync carrier 544 408 1088 1632 2176
Sync packet 1 1 3 5 7
The RS packet synchronization starts after TPS decoding the synchronization word. Hence, the synchronization carriers can be calculated after 17 symbols in 1st-4th frame. The
synchronization carriers are fixed in Table 2-5, Table 2-6 and Table 2-7. The equation is shown as:
residual carriers = (sync carriers) - remainder{[(frame_idx-1) 68+17]/(sync carriers)}× (2-22)
Table 2-5 residual carriers for RS packet synchronization for 2K mode Code rate 1/2 2/3 3/4 6/5 7/8 QPSK 1st frame 408 0 408 3672 408 QPSK 2nd frame 408 0 952 3672 2040 QPSK 3rd frame 408 0 408 3672 3672 QPSK 4th frame 408 0 952 3672 5304 16-QAM 1st frame 408 0 408 1224 408 16-QAM 2nd frame 408 0 408 1224 2040 16-QAM 3rd frame 408 0 408 1224 408 16-QAM 4th frame 408 0 408 1224 2040 64-QAM 1st frame 408 0 408 408 408 64-QAM 2nd frame 408 0 952 408 2040 64-QAM 3rd frame 408 0 408 408 1496 64-QAM 4th frame 408 0 952 408 952
Table 2-6 residual carriers for RS packet synchronization for 4K mode Code rate 1/2 2/3 3/4 6/5 7/8
QPSK 1st frame 816 0 816 2448 816 QPSK 2nd frame 816 0 816 2448 4080 QPSK 3rd frame 816 0 816 2448 816 QPSK 4th frame 816 0 816 2448 4080
16-QAM 1st frame 0 0 272 0 816 16-QAM 2nd frame 0 0 272 0 816 16-QAM 3rd frame 0 0 272 0 816 16-QAM 4th frame 0 0 272 0 816 64-QAM 1st frame 272 0 816 816 816 64-QAM 2nd frame 272 0 816 816 1904 64-QAM 3rd frame 272 0 816 816 816 64-QAM 4th frame 272 0 816 816 1904
Table 2-7 residual carriers for RS packet synchronization for 8K mode Code rate 1/2 2/3 3/4 6/5 7/8 QPSK 1st frame 0 0 544 0 1632 QPSK 2nd frame 0 0 544 0 1632 QPSK 3rd frame 0 0 544 0 1632 QPSK 4th frame 0 0 544 0 1632 16-QAM 1st frame 0 0 0 0 1632 16-QAM 2nd frame 0 0 0 0 1632 16-QAM 3rd frame 0 0 0 0 1632 16-QAM 4th frame 0 0 0 0 1632 64-QAM 1st frame 0 0 544 0 1632 64-QAM 2nd frame 0 0 544 0 1632 64-QAM 3rd frame 0 0 544 0 1632 64-QAM 4th frame 0 0 544 0 1632
We can use a counter to calculate the carrier index and then get completed packets and find the packet synchronization header. The additional complexity to complete this design is one
13-bit counter (>5304, the biggest number is 5304). In Table 2-5, Table 2-6 and Table 2-7, it shows the needed amount of subcarriers for RS packet synchronization.
2.4
Sampling Clock Offset Synchronization
This section introduces proposed SCO estimation algorithm and proposed SCO tracking loop architecture both. The proposed estimation algorithm and tracking loop architecture provide different improvements. The proposed estimation algorithm improves the estimation accuracy and the proposed tracking loop architecture reduces the time tracking. The blocks location is shown in Fig 2.23.
Fig 2.23 Location of fast synchronization in receiver platform
2.4.1 Conventional SCO Estimation
The first conventional SCO estimation algorithm is mentioned in [1] and [2], and it is a method of jointed SCO and CFO estimation (CFD/SFD) by calculating the phase difference between continual pilots of two consecutive symbols and then do average to the left part and
right part and calculate. The method has low mean error but unavoidable variance of SCO estimation. Then the linear least square is published [9]. Combine these two different methods, we obtain a jointed SCO and CFO LLS estimation method [10] implemented in [1]. The principle of the method is to calculate the phase difference between continual pilots of two consecutive symbols and then multiply a linear weight. Finally, calculate the slope of the line. The equations are listed as:
(
)
/ 2 1 ^ 2, , / 2 1 2 1 / M k l k k M g B y N N ζ π − =− = ⋅ ⋅ +∑
(2-23) * , , 1, 1 1 2 ( ) 1 1 | 1 l k l k l k T T i M y Arg z z B A A A k k A k CP k − − ⎡ ⎤ = ⎣ ⋅ ⎦ = ⎡ ⎤ ⎢ ⎥ ⎢ ⎥ = ∈ ⎢ ⎥ ⎢ ⎥ ⎢ ⎥ ⎣ ⎦ # #where ζ^ is estimated SCO, N means guard interval length, g N means 2/4/8k, z is l k,
carrier data, l is symbol index, k is continual pilot index, M is numerous of continual pilots in a symbol, y is the phase difference between continual pilots of two consecutive l k,
symbols, and B is the linearization matrix. The architecture of jointed SCO/CFO LLS estimation is shown in Fig 2.24.
1 2 1π⎛⎜⎝ +N Ng/ ⎞⎟⎠
Fig 2.24 Architecture of SCO estimation [1]
2.4.2 Proposed SCO Estimation
The proposed SCO estimation method has a key difference between the conventional designs: using the scatter pilots to replace the continual pilots. The equations become:
(
)
/ 2 1 ^ 2, , / 2 1 2 1 / 4 N x w x k N g D y N N ζ π − =− = ⋅ ⋅ + ⋅∑
(2-24) * , , 1, 1 1 2 ( ) 1 1 | 1 w x w x w x T T i N y Arg z z D C C C x x C x SP x − − ⎡ ⎤ = ⎣ ⋅ ⎦ = ⎡ ⎤ ⎢ ⎥ ⎢ ⎥ = ∈ ⎢ ⎥ ⎢ ⎥ ⎢ ⎥ ⎣ ⎦ # #Where z is carrier data, w is symbol index, x is scatter pilot index, N is numerous of scatter pilots in a symbol, y is the phase difference between scatter pilots of 4 symbol distance location, and D is the linearization matrix. The advantage is obvious, because the numerous of scatter pilots is 142/284/568 in 2/4/8k mode but the numerous of continual pilots is 45/89/177 in 2/4/8k mode. The architecture of proposed SCO estimation is shown in Fig 2.25.
1
4 2 1× π⎛⎜⎝ +N Ng/ ⎞⎟⎠
Fig 2.25 the architecture of proposed SCO estimation
Hence the proposed scatter pilot method is more accurate than continual pilot method. And the disadvantage of proposed method is the additional 4 buffers and throughput of estimation outcome up to 5 symbols. The buffers have already existed in 2-D channel estimation methodology [8] and the problem of throughput of estimation outcome will be solved in followed contents. We can also see the simulation results in chapter 3.
