A Novel High Step-Up DC-DC Converter for Microgrid System

IEEE Transaction on Power Electronics

Title: A Novel High Step-Up DC-DC Converter for Microgrid System Manuscript ID: TPEL-Reg-2010-06-0460

Authors: Yi-Ping Hsieh, Email: [email protected]

Jiann-Fuh Chen, Member, IEEE, Email: [email protected] Tsorng-Juu Liang, Member, IEEE, Email: [email protected] Lung-Sheng Yang, Email: [email protected]

Address:

Green Energy Electronics Research Center, Department of Electrical Engineering, National Cheng Kung University, Tainan, Taiwan, R.O.C.

Footnote of the first page:

This work made use of Shared Facilities supported by the Research Center of Ocean Environment and Technology , Ocean Energy Research Center, and the National Science Council, Taiwan, under Award Numbers NSC 97-2221-E-006-278-MY3.

Copyright © 2010 IEEE. Personal use of this material is permitted. However, permission to use this material for any other purposes must be obtained from the IEEE by sending a request to [email protected].

The authors are with the Green Energy Electronics Research Center

,

Abstract

A novel high step-up DC-DC converter for distributed generation (DG) system is proposed in

this paper. The concept is composed of two capacitors, two diodes, and one coupled-inductor. Two

capacitors are charged in parallel, and are discharged in series by the coupled-inductor. Thus, high

step-up voltage gain can be achieved with an appropriate duty ratio. The voltage stresses on the

main switch and output diode are reduced by a passive clamp circuit. Therefore, low resistance

R

for the main switch can be adopted to reduce conduction loss. In addition, the

reverse-recovery problem of the diode is alleviated, and thus, the efficiency can be further improved.

The operating principle and steady-state analyses of the voltage gain are also discussed in detail.

Finally, a 24-V input voltage, 400-V output voltage, and 400-W output power prototype circuit of

the proposed converter is implemented in the laboratory to verify the performance.

Index Terms – High step-up, microgrid, DG system, coupled-inductor

I. I

The distributed generation (DG) systems based on the renewable energy sources (RES) have

rapidly developed in recent years [1], [2]. These DG systems are powered by microsources such as

fuel cells, photovoltaic systems, and batteries [3]-[7]. Fig. 1 shows a photovoltaic (PV) distributed

system in which the solar source is low DC input voltage. PV sources can also connect in series to

obtain sufficient DC voltage for generating AC utility voltage; however, it is difficult to realize a

series connection of the PV source without incurring a shadow effect [8], [9]. High step-up DC-DC

converters are generally used as the front-end converters to step from low voltage (12-40 V) up to

high voltage (380-400 V) [10]. High step-up DC-DC converters are required to have a large

conversion ratio, high efficiency, and small volume [11]-[16].

The isolated converters such as forward, flyback, push-pull, half-bridge, and full-bridge types

can adjust the turns ratio of the transformer to achieve high step-up voltage gain. However, the main

switches of these converters will suffer a high voltage spike and high power dissipation from the

leakage inductor of the transformer [17]. To reduce these drawbacks, the non-dissipative snubber

circuits and active-clamp circuits are employed. However, the cost increases accordingly due to the

extra power switch and high side driver [18].

Theoretically, the non-isolated converters can be adopted to provide high step-up voltage gain

with extremely high duty cycle [19]. However, the step-up voltage gain is limited by the effect of

the power switches, rectifier diodes, and equivalent series resistance (ESR) of the inductors and

capacitors. The extreme duty cycle operation may also result in serious reverse-recovery problem

and electromagnetic interference (EMI) problem [20].

To improve the conversion efficiency and achieve high step-up voltage gain, many topologies

have been proposed [19]-[28]. High step-up gain can be achieved by the use of the switched

capacitor technique [22]-[25]. However, the main switch will suffer high transient current, and the

conduction loss is high. Another method for achieving high step-up gain is the use of the

voltage-lift technique [26]-[28]. However, it has the same drawback.

The converters employ the coupled-inductor technique to achieve high step-up gain by

adjusting the turns ratio [29]. However, the leakage inductor of the coupled inductor incurs a

voltage spike on the main switch and affects the conversion efficiency. For this reason, the

converters using the coupled-inductor technique with an active clamp circuit have also been

proposed [30], [31]. An integrated boost-flyback converter is presented, in which the secondary side

of the coupled-inductor is used as a flyback type [32], [33]. The leakage-inductor energy of the

coupled inductor is recycled into the load during the switch-off period, thus the voltage spike on the

main switch can be limited. Additionally, the voltage stress of the main switch can be adjusted by

the turns ratio of the coupled inductor. To achieve high step-up voltage gain, it has been proposed

that the secondary side of the coupled-inductor can be used as a flyback and a forward type

[34]-[36]. Also, several converters that combine output-voltage stacking to increase voltage gain are

proposed [37], [38]. The sepic-flyback converter with the coupled inductor and output stacking

techniques has been proposed [39]. Additionally, a high step-up boost converter that uses multiple

coupled-inductor with output stacking have been proposed [40], [41].

This paper proposes a high efficiency, high step-up voltage gain, and clamp-mode converter.

The proposed converter adds two pairs of additional capacitors and diodes to achieve high step-up

voltage gain. The coupled-inductor is used as both a forward and flyback type, thus the two

capacitors can be charged in parallel and discharged in series via the coupled-inductor. The transit

current does not flow through the main switch compared with earlier studies [22]-[28]. Thus, the

proposed converter has low conduction loss. Additionally, this converter allows significant weight

and volume reduction compared with other converters [29]-[40]. Another benefit is that the voltage

stresses on the main switch and output diode are reduced. However, the leakage inductor of the

coupled-inductor may cause high power loss and voltage spike. Thus, a passive clamping circuit is

needed to recycle the leakage-inductor energy of the coupled inductor and to clamp the voltage

across the main switch. The reverse-recovery problems in the diodes are alleviated, and thus high

efficiency can be achieved.

II. O

P

P

C

Fig. 2 shows the circuit topology of the proposed converter. This converter consists of DC

input voltage V

, power switch S, coupled-inductor N

and N

, one clamp diode D

, clamp capacitor

C

, two blocking capacitors C

and C

, two blocking diodes D

and D

output diode D

, and output

capacitor C

. The coupled inductor is modeled as the magnetizing inductor L

and leakage inductor

L

.

To simplify the circuit analysis, the following conditions are assumed:

1) Capacitors C

, C

, and C

are large enough that V

, V

and V

are considered to be constant

in one switching period.

2) The power MOSFET and diodes are treated as ideal, but the parasitic capacitor of the power

switch is considered.

3) The coupling-coefficient of coupled-inductor k is equal to L

/(L

+L

) and the turns ratio of

coupled-inductor n is equal to N

/N

.

(A) Continuous-Conduction Mode (CCM) Operation

In CCM operation, there are six operating modes in one switching period of the proposed

converter. Fig. 3 shows the typical waveforms and Fig. 4 shows the current-flow path of the

proposed converter for each modes. The operating modes are described as follows:

1) Mode I [t

, t

]: During this time interval, S is turned on. Diodes D

, D

, and D

are turned off, and

D

is turned on. The current-flow path is shown in Fig. 4(a). The primary-side current of the

coupled inductor i

is increased linearly. The magnetizing inductor L

stores its energy from

DC-source V

. Due to the leakage inductor L

, the secondary-side current of the coupled

inductor i

is decreased linearly. The voltage across the secondary-side winding of the coupled

inductor V

, and blocking voltages V

and V

are connected in series to charge the output

capacitor C

and to provide the energy to the load R. When the current i

becomes zero,

DC-source V

begins to charge capacitors C

and C

via the coupled inductor. When i

is equal

to i

at t = t

, this operating mode ends.

2) Mode II [t

, t

]: During this time interval, S is still turned on. Diodes D

and D

are turned off,

and D

and D

are turned on. The current-flow path is shown in Fig. 4(b). The magnetizing

inductor L

is stored energy from DC-source V

. Some of the energy from DC-source V

transfers to the secondary side of the coupled inductor to charge the capacitors C

and C

.

Voltages V

and V

are approximately equal to nV

. Output capacitor C

provides the energy to

load R. This operating mode ends when switch S is turned off at t = t

.

3) Mode III [t

, t

]: During this time interval, S is turned off. Diodes D

and D

are turned off, and

D

and D

are turned on. The current-flow path is shown in Fig. 4(c). The energies of leakage

inductor L

and magnetizing inductor L

are released to the parasitic capacitor C

of switch S.

The capacitors, C

and C

, are still charged by the DC-source V

via the coupled inductor. The

output capacitor C

provides energy to load R. When the capacitor voltage V

+V

is equal to V

at t = t

, diode D

conducts and this operating mode ends.

4) Mode IV [t

, t

]: During this time interval, S is turned off. Diodes D

, D

, and D

are turned on

and D

is turned off. The current-flow path is shown in Fig. 4(d). The energies of leakage

inductor L

and magnetizing inductor L

are released to the clamp capacitor C

. Some of the

energy stored in L

starts to release to capacitors C

and C

in parallel via the coupled inductor

until secondary current i

equals to zero. Meanwhile, current i

is decreased quickly. Thus,

diodes D

and D

are cut off at t = t

, and this operating mode ends.

5) Mode V [t

, t

]: During this time interval, S is turned off. Diodes D

and D

are turned on, and D

and D

are turned off. The current-flow path is shown in Fig. 4(e). The energy of leakage

inductor L

and magnetizing inductor L

are released to the clamp capacitor C

. The

primary-side and secondary-side windings of the coupled inductor, DC sources V

, and

capacitors, C

and C

, are series to transfer their energies the output capacitor C

and load R.

This operating mode ends when capacitor C

starts to discharge at t = t

.

6) Mode VI [t

, t

]: During this time interval, S is still turned off. Diodes D

and D

are turned on,

and D

and D

are turned off. The current-flow path is shown in Fig. 4(f). The primary-side and

secondary-side windings of the coupled inductor, DC sources V

, and capacitors, C

, C

and C

,

transfer their energies the output capacitor C

and load R. This mode ends at t = t

when S is

turned on at the beginning of the next switching period.

(B) Discontinuous-Conduction Mode (DCM) Operation

In order to simplify the analysis for DCM operation, leakage inductor L

of the

coupled-inductor is neglected. Fig. 5 shows the typical waveforms when the proposed converter is

operated in DCM, and Fig. 6 shows the current-flow path of the proposed converter for each modes.

There are three modes in DCM operation. The operating modes are described as follows:

1) Mode I [t

, t

]: During this time interval, S is turned on. The current-flow path is shown in Fig.

6(a). The part energy of DC-source V

transfers to magnetizing inductor L

. Thus, i

is

increased linearly. The DC-source V

also transfers another part energy to charge capacitors C

and C

via the coupled inductor. The energy of the output capacitor C

is discharged to load R.

This mode ends when S is turned off at t = t

.

2) Mode II [t

, t

]: During this time interval, S is turned off. The current-flow path is shown in Fig.

6(b). The energy of the magnetizing inductor L

is released to the capacitor C

. Similarly,

capacitors C

and C

are discharged in a series with DC source V

and magnetizing inductor L

to the capacitor C

and load R. This mode ends when the energy stored in L

is depleted at t = t

.

3) Mode III [t

, t

]: During this time interval, S remains turned off. The current-flow path is shown

in Fig. 6(c). Since the energy stored in L

is depleted, the energy stored in C

is discharged to

load R. This mode ends when S is turned on at t = t

.

III. S

-S

A

P

C

(A) CCM Operation

At modes IV and V, the energy of the leakage inductor L

is released to the clamped capacitor

C

. According to previous work Ref. [15], the duty cycle of the released energy can be expressed as

1 1

2(1 )

1 ,

c c

s

t D

D T n

  



(1)

where T

is the switching period, D

is the duty ratio of the switch, and t

is the time of modes IV

and V.

By applying the voltage-second balance principle on L

, the voltage across the capacitor C

can be represented by

1

(1 ) (1 )

1 2 .

c in

D k

V V

D

  

  



k n

(2)

Since the time durations of modes I, III, and IV are significantly short, only modes II, V, and

VI are considered in CCM operation for the steady-state analysis.

In the time period of mode II, the following equations can be written based on Fig. 4(b).

1

1

II m ,

L in in

m k

v L V

L L

 

 kV

2 1

(3)

II II .

L L in

2 3 2

c c L in

v nv nkV

(4)

Thus, the voltage across capacitors C

and C

can be written as

.

(5)

V V vII nkV

During the time duration of modes V and VI, the following equation can be formulated based on

Fig. 4(f).

2= 2 1 2 3 .

V VI

L L in c c c o

v v V V V V V

(6)

Thus, the voltage across the magnetizing inductor L

can be derived as

1 2 3

2 1= 1

VI

V VI L in c c c o

L L

V V V V V

v v v

n n

   

  .



(7)

Using the volt-second balance principle on L

, the following equation is given

1 1

0 ^{s s} 0.

s

DT II T VI

L DT L

v dt v dt

  (8)

Substituting (2), (3), (5), and (7) into (8), the voltage gain is obtained as

1 (1

1 1 2 .

CCM

nk D k n

M nk

D D

 

   

 

)( 1)

(9)

The schematic of the voltage-gain versus the duty-ratio under various the coupling coefficients of

the coupled-inductor is shown in Fig. 7. It is seen that the voltage gain is not very sensitive to the

coupling-coefficient. When k is equal to 1, the ideal voltage gain is written as

1

CCM 1

M n n

D

  



(10)

Fig. 8 shows the voltage gain versus the duty ratio of the proposed converter as compared with

the converters in previous work [35] and [36] at CCM operation under k =1 and n =3. One can see

that the voltage gain of the proposed converter is higher than the converters in [35] and [36].

According to the description of the operating modes, the voltage stresses on the active switch S and diodes D

, D

, D

, and D

are given as

1 ,

1 1

o in

DS in

V nV

V V

D n

  

 

(11)

1

1 ,

1 1

o in

D in

V nV

V V

D n

  

 

(12)

2 3 ( )

1 1 .

D D Do in o in

n n

V V V V V nV

D n

    

 

1

(13)

Equations (11), (12), and (13) mean that with the same specifications, the voltage stresses on

the main switch and diodes can be adjusted by the turns ratio of the coupled inductor.

(B)DCM Operation

In DCM operation, three modes are discussed. The key waveform is shown in Fig. 5. During

the time of mode I, switch S is turned on. Thus, the following equations can be formulated based on

Fig. 6(a)

I ,

L in

2

v V

(14)

I .

L in

v nV

(15)

The peak value of the magnetizing-inductor current is given as

in .

Lmp s

m

I V DT

L

(16)

Furthermore, the voltage across capacitors C

and C

can be written as

2 3 2

c c L in

1 1

.

(17)

V V vI nV

In the time interval of mode II, the following equations can be expressed based on Fig. 6(b):

II ,

L c

2 1 2 3

v  V

(18)

II .

L in c c c o

1 2

L L

v V V V V V

(19)

During the time of mode III, the following equation can be derived from Fig. 6(c):

III III 0.

v v 

(20)

Applying the voltage-second balance principle on N

, N

of the coupled inductor, the following

equations are given as

( )

1 1 1

0 ^{s L s} (^s 0.

s L s

DT I D D T II T III

L DT L D D T L

v dt ^ v dt v dt

  

  

) 

(21)

( )

2 2 2

0 ^{s L s} (^s 0.

s L s

DT I D D T II T III

L DT L D D T L

v dt ^ v dt v dt

  

   (22)

Substituting (14), (15), (17), (18), (19), and (20) into (21) and (22), the voltage gain is obtained as

follows:

1 ,

c L

V DV

D _in

(23)

( 1) (2 1) .

(24)

o in

L

V D n n V

D

 

    

 

According to (24), the duty cycle D

can be derived as

(1 )

(1 2 ) .

in L

o i

D n DV

V n

 

  V_n

(25) From Fig. 5, the average current of i

is computed as

1 .

2 1

Lmp

co L o

I D I

 n

 I

(26) Since I

is equal to zero under steady state, Equations (16), (25), and I

= 0 into (26) yields

 

2 2

2 (1 2 ) .

in s o

o in m

D V T V

V n V L 

  R

(27) Then, the normalized magnetizing-inductor time constant is defined as

m m s,

Lm s

L L f

RT R

  

(28)

where f

is the switching frequency.

Substituting (28) into (27), the voltage gain is given by

2 2

1 2 (1 2 )

2 4 2 .

o DCM

in Lm

V n n

M V

D



 

   

(29)

The curve of the voltage gain, shown in Fig. 9, illustrates the voltage-gain versus the duty-ratio

under various τ

.

(C) Boundary Operating Condition between CCM and DCM

If the proposed converter is operated in boundary-condition mode (BCM), the voltage gain of

CCM operation is equal to the voltage gain of DCM operation. From (10) and (29), the boundary

normalized magnetizing-inductor time constant, τ

, can be derived as

(1 )2

2(1 )(1 2 ).

LmB

D D

n n nD

  ^

  

(30)

The curve of τ

is plotted in Fig. 10. If τ

is larger than τ

, the proposed converter is operated

in CCM.

IV. D

E

P

C

To verify the performance of the proposed converter, a prototype circuit is implemented in the

laboratory. The specifications are as follows:

1) input DC voltage V

: 24 V

2) output DC voltage V

: 400 V

3) maximum output power: 400 W

4) switching frequency: 50 kHz

5) MOSFET S: IRFB4410ZPBF

6) Diodes D

: SBR20A100CTFP, D

/D

: DESI30, and D

: BYR29

7) Coupled inductor: ETD-59, core pc40, N

: N

= 1 : 4

L

= 48 H; L

= 0.25 H

8) Capacitors C

: 56 F/ 100 V, C

/C

: 4.7 F/ 200 V, and C

: 180 F/ 450 V

Fig. 11 shows the measured waveforms for full-load P

= 400 W and V

= 24 V. The

proposed converter is operated in CCM under full-load condition. The waveforms demonstrate that

the steady-state analysis is correct. In the measured waveforms, v

is clamped at appropriately 90

V during the switch-off period. Therefore, a low-voltage-rated switch is adopted to achieve high

efficiency for the proposed converter.

The waveform of the secondary-side current of the coupled-inductor i

in Fig. 11(a) shows

that the proposed converter is operated in CCM because the current is not equal to zero when the

switch is turned on. In Fig. 11(b), the waveforms of i

and i

show that capacitors C

and C

are

charged in parallel. Fig. 11(c) shows that the energy of leakage inductor L

is released to capacitor

C

from i

. Fig. 11(d) reveals that V

and V

are able to satisfy Equations (2) and (5). In addition,

output voltage V

is consistent with Equation (10). Fig. 11(e) shows the voltage stresses of the main

switch and diodes, and demonstrates the consistency of Equations (11), (12), and (13). Fig. 12

shows the light-load waveforms. The output voltage is about 400 V and the analysis of the DCM of

the proposed converter is demonstrated.

Fig. 13 shows the experimental conversion efficiency of the proposed converter, which the

maximum efficiency is around 96.8% at P

= 80 W and the full-load efficiency is appropriately 94.2

% at P

= 400 W. The efficiency of converter in Ref. [41] is also showed. The maximum efficiency

is 94.3% and full-load efficiency is 92.25%. The results show that the proposed converter has

promoted the efficiency about 2%.

The results verify the high efficiency of the proposed converter. Because the low input voltage

is applied in this converter, the proposed converter should experience extremely high input current

at full-load, which would leads to high conduction loss during the switch turn-on period. To further

improving the conduction loss, two of the proposed converter can be used in interleaved operation

where the input current is shared by two switches to achieve high efficiency.

V. C

This paper proposed a novel, high efficiency and high step-up DC-DC converter. By using the

capacitor charged in parallel and discharged in a series by the coupled-inductor, high step-up

voltage gain and high efficiency are achieved. The steady-state analyses of voltage gain and

boundary operating condition are discussed in detail. A prototype circuit of the proposed converter

is built in the laboratory. Experimental results confirm that high efficiency and high step-up voltage

gain can be achieved. The efficiency is 96.8%. The voltage stress on the main switches is 90 V, thus

low voltage ratings and low on-state resistance levels R

switch can be selected. Moreover, the

proposed converter has simple structure. It is suitable for renewable energy systems in microgrid

applications.

R

[1] Y. Li and Y. W. Li, “Decoupled power control for an inverter based low voltage microgrid in autonomous

operation,” in Proc. IEEE IPEMC, pp. 2490-2496, 2009.

[2] Y. Li, D. M. Vilathgamuwa, and P. H. Loh, “Design, analysis, and real-time testing of a controller for multibus

microgrid system,” IEEE Trans. Power Electron., vol. 19, no. 5, pp. 1195-1204, 2004.

[3] C. L. Chen, Y. W, J. S. Lai, Y.S. Lee, and D. Martin, “Design of parallel inverters for smooth mode transfer

microgrid applications,” IEEE Trans. Power Electron., vol. 25, no. 1, pp. 6-15, Jan. 2010

[4] A. Timbus, M. Liserre, R. Teodorescu, P. Rodriguez, and F. Blaabjerg, “Evaluation of current controllers for

distributed power generation systems,” IEEE Trans. Power Electron., vol. 24, no. 3, pp. 654-664, Mar. 2009.

[5] Y.A.-R.I. Mohamed and E.F. El Saadany, “Hybrid variable-structure control with evolutionary optimum-tuning

algorithm for fast grid-voltage regulation using inverter-based distributed generation,” IEEE Trans. Power Electron.,

vol. 23, no. 3, pp. 1334-1341, May. 2008.

[6] Y.A.-R.I. Mohamed and E.F. El Saadany, “Adaptive decentralized droop controller to preserve power sharing

stability of paralleled inverters in distributed generation microgrids,” IEEE Trans. Power Electron., vol. 23, no. 6,

pp. 2806-2816, Nov. 2008.

[7] Y.W. Li and C.-N. Kao, “An accurate power control strategy for power-electronics-interfaced distributed

generation units operating in a low-voltage multibus microgrid,” IEEE Trans. Power Electron., vol. 24, no. 12, pp.

2977-2988, Dec. 2009.

[8] H. Karimi, A. Yazdani, and R. Iravani, “Negative-sequence current injection for fast islanding detection of a

distributed resource unit,” IEEE Trans. Power Electron., vol. 23, no. 1, pp. 298-307, Jan. 2008.

[9] T. Shimizu, K. Wada, and N. Nakamura, “Flyback-type single-phase utility interactive inverter with power

pulsation decoupling on the dc input for an ac photovoltaic module system,” IEEE Trans. Power Electron., vol. 21,

no. 5, pp. 1264-1272, Sep. 2006.

[10] L. Palma, M. H. Todorovic, and P. Enjeti, “A high gain transformer-less DC-DC converter for fuel-cell

applications,” in Proc. IEEE PESC, pp. 2514-2520, 2005.

[11] P. Biczel, “Power electronic converters in dc microgrid,” in Proc. IEEE CPE Conf., pp. 1-6, 2007.

[12] A.M. Salamah, S.J. Finney, and B.W. Williams, “Single-phase voltage source inverter with a bidirectional

buck–boost stage for harmonic injection and distributed generation,” IEEE Trans. Power Electron., vol. 24, no. 2, pp.

376-387, Feb. 2009.

[13] A. Cid-Pastor, L. MartÕnez-Salamero, C. Alonso, A. El Aroudi, and H. Valderrama-Blavi, “Power distribution

based on gyrators,” IEEE Trans. Power Electron., vol. 24, no. 12, pp. 2907-2909, Dec. 2009.

[14] L. S. Yang, T. J. Liang, and J. F. Chen, “Transformer-less DC-DC converter with high voltage gain,” IEEE Trans.

Ind. Electron., vol. 56, no. 8, pp. 3144-3152, Aug. 2009.

[15] Q. Zhao and F. C. Lee, “High-efficiency, high step-up dc-dc converters,” IEEE Trans. Power Electron., vol. 18,

no. 1, pp. 65-73, Jan. 2003.

[16] R. J. Wai, C. Y. Lin, C. Y. Lin, R. Y. Duan, and Y. R. Chang, “High-efficiency power conversion system for

kilowatt-level stand-alone generation unit with low input voltage,” IEEE Trans. Ind. Electron., vol. 55, no. 10, pp.

3702-3714, Oct. 2008.

[17] N. P. Papanikolaou and E. C. Tatakis, “Active voltage clamp in flyback converters operating in CCM mode

under wide load variation,” IEEE Trans. Ind. Electron., vol. 51, no. 3, pp. 632-640, Jun. 2004.

[18] J. M. Kwon and B. H. Kwon, “High step-up active-clamp converter with input-current doubler and

output-voltage doubler for fuel cell power systems,” IEEE Trans. Power Electron., vol. 24, no. 1, pp. 108-115, Jan.

2009.

[19] B. Axelrod, Y. Berkovich, and A. Ioinovici, “Transformerless DC-DC converters with a very high DC

line-to-load voltage ratio,” Proc. IEEE ISCAS, pp. III435-III438, 2003.

[20] R. J. Wai and R. Y. Duan, “High-efficiency DC/DC converter with high voltage gain,” IEE Proc. Inst. Elect.

Eng.-Electric Power Applications, vol. 152, no.4, pp. 793-802, Jul. 2005.

[21] J.A. Carr, D. Hotz, J.C. Balda, H.A. Mantooth, A. Ong, and A. Agarwal, “Assessing the impact of SiC

MOSFETs on converter interfaces for distributed energy resources,” IEEE Trans. Power Electron., vol. 24, no. 1,

pp. 260-270, Jan. 2009.

[22] Fan Zhang, Lei Du, Fang Zheng Peng, and Zhaoming Qian, “A new design method for high-power high-efficiency

switched-capacitor dc–dc converters,” IEEE Trans. Power Electron., vol. 23, no. 2, pp. 832-840, Mar. 2008.

[23] O. Abutbul, A. Gherlitz, Y. Berkovich, and A. Ioinovici, “Step-up switching-mode converter with high voltage

gain using a switched-capacitor circuit,” IEEE Trans. Circuits and Systems I, vol. 50, no. 8, pp. 1098-1102, Aug.

2003.

[24] B. Axelrod, Y. Berkovich, and A. Ioinovici, “Switched-capacitor/ switched-inductor structures for getting

transformerless hybrid DC-DC PWM converters,” IEEE Trans. Circuits and Systems I, vol. 55, no. 2, pp. 687-696,

Mar. 2008.

[25] B. Axelrod, Y. Berkovich, and A. Ioinovici, “Switched-capapcitor (SC)/Switched-inductor (SL) structures for

getting hybrid step-down CUK/ZETA/SEPIC converters,” in Proc. IEEE ISCAS, pp. 5063-5066, 2006.

[26] F. L. Luo, “Six self-lift DC–DC converters, voltage lift technique,” IEEE Trans. Ind. Electron., vol. 48, no. 6, pp.

1268-1272, Dec. 2001.

[27] F. L. Luo and H. Ye, “Positive output super-lift converters,” IEEE Trans. Power Electron., vol. 18, no. 1, pp.

105-113, Jan. 2003.

[28] F. L. Luo and H. Ye, “Positive output multiple-lift push–pull switched-capacitor Luo-converters,” IEEE Trans. Ind.

Electron., vol. 51, no. 3, pp. 594-602, Jun. 2004.

[29] B. Axelrod, Y. Berkovich, and A. Ioinovici, “Switched coupled-inductor cell for DC-DC converters with very

large conversion ratio,” in Proc. IEEE IECON Conf., pp. 2366-2371, 2006.

[30] B. R. Lin and F. Y. Hsieh, “Soft-switching zeta–flyback converter with a buck–boost type of active clamp,” IEEE

Trans. Ind. Electron., vol. 54, no. 5, pp. 2813-2822, Oct. 2007.

[31] T. F. Wu, Y. S. Lai, J. C. Hung, and Y. M. Chen, “Boost converter with coupled inductors and buck–boost type of

active clamp,” IEEE Trans. Ind. Electron., vol. 55, no. 1, pp. 154-162, Jan. 2008.

[32] K. C. Tseng and T. J. Liang, “Novel high-efficiency step-up converter,” IEE Proc. Inst. Elect. Eng.-Electric Power

Applications, vol. 151, no. 2, pp. 182-190, Mar. 2004.

[33] K. C. Tseng and T. J. Liang, “Analysis of intergrated boost-flyback step-up converter,” IEE Proc. Inst. Elect.

Eng.-Electric Power Applications, vol. 152, no. 2, pp. 217-225, 2005.

[34] R. J. Wai and R. Y. Duan, “High step-up converter with coupled-inductor,” IEEE Trans. Power Electron., vol. 20,

no. 5, pp. 1025-1035, Sep. 2005.

[35] R. J. Wai, L. W. Liu, and R. Y. Duan, “High-efficiency voltage-clamped DC–DC converter with reduced

reverse-recovery current and switch-voltage stress,” IEEE Trans. Ind. Electron., vol. 53, no. 1, pp. 272-280, Feb.

2005.

[36] J. W. Baek, M. H. Ryoo, T. J. Kim, D. W. Yoo, and J. S. Kim, “High boost converter using voltage multiplier,” in

Proc. IEEE IECON, pp. 567-572, 2005.

[37] T. Dumrongkittigule, V. Tarateeraseth, and W. Khan-ngern, “A new integrated inductor balanced switching

technique for common mode EMI reduction in high step-up DC/DC converter,” in Proc. International Zurich

Symposium on Electromagnetic Compatibility, pp. 541-544, 2006

[38] Y. J. A. Alcazar, R. T. Bascope, D. S. de Oliveira, E. H. P. Andrade, and W. G. Cardenas, “High voltage gain

boost converter based on three-state switching cell and voltage multipliers,” in Proc. IEEE IECON, pp. 2346-2352,

2008.

[39] K. B. Park, H. W. Seong, H. S. Kim, G. W. Moon, and M. J. Youn, “Integrated boost-sepic converter for high

step-up applications,” in Proc. Power Electronics Specialists Conf., Rohode, Greece, pp. 944-950, 2008.

[40] G. V. T. Bascope, R. P. T. Bascope, D. S. Oliveira Jr., S. A. Vasconcelos, F. L. M. Antunes, and C. G. C. Branco,

“A high step-up DC-DC converter based on three-state switching cell,” in Proc. IEEE ISIE, pp. 998-1003, 2006.

[41] S. K. Changchien, T. J. Liang, J. F. Chen, L. S. Yang, “Novel high step-up DC-DC converter for fuel cell energy

conversion system,” IEEE Trans. Ind. Electron., vol. 57, no. 6, pp. 2007-2017, 2010.

Fig. 1. photovoltaic (PV) distributed system

Fig. 2. Circuit configure of the proposed converter

Fig. 3. Some typical key waveforms of the proposed converter at CCM operation.

Vo

Vin S Np

D2

C_o R Ns

Lk

D3

C₂ C₃

L_m

Do

C1

D1

(a) Mode I

(b) Mode II

(c) Mode III

(d) Mode IV

Vo

Vin S Np

D₂

C_o R Ns

L_k

D₃

C₂ C₃

L_m

D_o C₁

D1

(e) Mode V

(f) Mode VI

Fig. 4. Current flowing path of operating modes during one switching period at CCM operation.

(a) Modes I. (b) Modes II. (c) Mode III. (d) Mode IV. (e) Mode V. (f) Mode VI.

0

i

i

t

0

0 t

t

DT_s DLTs

Ts

i

v

0

i

t

t0 t₁ t2 t3

ModeI ModeII ModeIII

0 t

i

I

0

i

t

ILmp/n+1

-Io

I_Lmp/n+1

ILmp/n+1 - Io

Fig. 5. Some typical key waveforms of the proposed converter at DCM operation.

Vo

Vin S Np

D2

Co R Ns

D3

C2 C3

Lm

Do

C1

D1

(a) Mode I

Vo

Vin S Np

D2

Co R Ns

D3

C2 C3

Lm

Do

C1

D1

(b) Mode II

V_o Vin S

Np

D₂

Co R Ns

D3

C₂ C₃

Lm

D_o C1

D1

(c) Mode III

Fig. 6. Current flowing path of operating modes during one switching period at DCM operation.

(a) Modes I. (b) Mode II. (c) Modes III.

Fig. 7. Voltage-gain versus duty-ratio at CCM operation under n = 3 and various k.

Fig. 8. Voltage-gain versus duty-ratio of the proposed converter, the converters in [35] and [36] at CCM operation under n = 3 and k =1.

Vo / Vin

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 0

5 10 15 20 25 30 35 40 45 50

D

DCM (under ٛLm = 0.001) CCM

DCM (under ٛLm = 0.0015) DCM (under ٛLm = 0.002)

Fig. 9. Voltage-gain versus duty-ratio at DCM operation under various τ

and at CCM operation under n = 3 and k = 1.



Fig. 10. Boundary condition of the proposed converter under n = 3.

vds: 40V/div

iLk: 20A/div

is: 5A/div

Time : 5 s/div

(a)

(b)

(c)

(d)

(e)

Fig. 11. Experiment results under full-load P

= 400 W.

(a)

(b)

(c)

(d)

(e)

Fig. 12. Experiment results under light-load P

= 40 W.

Fig. 13. Experimental conversion efficiency.