### 行政院國家科學委員會專題研究計畫 成果報告

### 子計畫二：以正交分頻多工為基礎之多模式基頻收發器研製

### (3/3)

計畫類別： 整合型計畫 計畫編號： NSC93-2220-E-009-033- 執行期間： 93 年 08 月 01 日至 94 年 07 月 31 日 執行單位： 國立交通大學電子工程學系及電子研究所 計畫主持人： 李鎮宜 報告類型： 完整報告 報告附件： 出席國際會議研究心得報告及發表論文 處理方式： 本計畫可公開查詢### 中 華 民 國 95 年 6 月 1 日

### 行政院國家科學委員會補助專題研究計畫成果報告

### 用於軟體無線電基頻處理之系統晶片設計技術

### 子計劃二：以正交分頻多工為基礎之多模式基頻收發器研製（3/3）

計畫類別：□個別型計畫 ■整合型計畫 計畫編號：NSC 91-2218-Ｅ-009-010 執行期間：91 年 8 月 1 日 至 93 年 7 月 31 日 計畫編號：NSC92－2220－E－009-019－ 執行期間：92 年 8 月 1 日 至 93 年 7 月 31 日 計畫編號：NSC93－2220－E－009-033－ 執行期間：93 年 8 月 1 日 至 94 年 7 月 31 日 計畫主持人：李鎮宜 計畫參與人員： 執行單位：國立交通大學電子工程系所### 中 華 民 國 94 年 10 月 28 日

**中文摘要 **

在此結案報告中將敘述第三年有關於 DVB-T 基頻接收系統中關鍵模組的改進與設 計。相關的研究項目包含頻率同步系統的改善與實現，以及整個 DVB-T 基頻系統的架 構設計、晶片實現。在系統整合過程中，我們將三年內陸續發展的相關關鍵模組，例如： FFT processor, Viterbi decoder, RS decoder, De-interleaver 完全整合於單一晶片中，並且

通過完整的功能、工作速度、功率消耗量測。

關鍵字：數位視訊系統、正交分頻多工、頻率同步演算法、系統整合與實現

**Abstract: **

This final report describes the project progress in third year about developing,

implementing core technologies for OFDM-based digital video broadcasting (DVB) system

but also DVB-H system. The research tasks include frequency synchronization system

improvement, DVB-T/H baseband receiver design, and implementation. We integrate

DVB-T/H baseband receiver by several developed functional block designs, such as 2k/4k/8k

point processor, Viterbi decoder, single memory de-interleaver, and RS decoder. Finally, the

measurement result of single chip DVB-T/H baseband receiver will be reported in the end.

Keywords: DVB-T baseband receiver, OFDM, Frequency Synchronization Algorithm,

**Part I: Low Complexity Carrier Frequency Synchronization for DVB-T/H **

**System **

A low complexity carrier frequency offset (CFO) synchronization scheme is proposed

for Digital Video Broadcasting-Terrestrial/Handheld (DVB-T/H) system, which comprises

two acquisition strategies and a tracking loop. In time-domain, Pre-FFF algorithm, the

proposed fractional CFO acquisition algorithm can overcome the distortion caused by

multipath delay spread and achieves 0.25~7.8dB gain in RMSE compared with the

conventional approach. In frequency-domain, Post-FFT algorithm, a 2-stage scheme is

proposed for the integral CFO acquisition to reduce the search range. In the other hand, we

propose two low complexity algorithms to detect the accurate integral CFO value and save

more than 80% of number of multiplication without any performance loss.

**1. Carrier Frequency Offset Synchronization Scheme **

The objective of CFO synchronization is to establish subcarrier orthogonality as fast and

accurately as possible (acquisition) and then maintain orthogonality as well as possible at all

times during online reception (tracking). However, a CFO acquisition algorithm alone can not

be both fast and sufficiently accurate, because

1. Pre-FFT algorithms allow only fast acquisition of the fractional CFO but no

acquisition of the integral CFO.

2. Post-FFT algorithms allow fast acquisition of the integral CFO but, due to lack of

orthogonality, acquisition of fractional CFO is very complicate.

Both fast and accurate acquisition can be attained by adopting a multi-stage

synchronization strategy with two one-shot acquisition stages (one pre-FFT and the other

post-FFT) followed by tracking. In DVB-T/H system, the data format provides for training is

only for frequency domain (continual and scattered pilots) but not for time domain. Hence,

pre-FFT non-data-aided acquisition and post-FFT data-aided acquisition and tracking

scheme as shown in Fig. 1.
Pre-FFT
CFO
Acquisition
( )
Post-FFT
CFO
Acquisition
( )
FFT
CFO
Compensator
Post-FFTCFO
Tracking ( )
**+** ε*I*
^
*I*

### ε

dataCFO ^*F*

### ε

^*F*ε

*R*ε

*R*ε

*R*

### ε

from ADC to EQFig. 1: Overall CFO synchronization and compensation scheme

The control loops of the three-stage synchronization subsystem operate in a

per-OFDM-symbol basis. When the CFO acquisition or tracking stage has generated an

estimation of CFO value, the CFO compensator will calculate the effective compensation

value before the beginning of the next pre-FFT OFDM symbol, and then start to compensate

the updated CFO value when the next pre-FFT OFDM symbol comes.

**1.1 Fractional Carrier Frequency Offset Synchronization **

The conventional fractional CFO estimation utilizes maximum likelihood estimation

(MLE) of differential phase between two repeated training symbols in frequency domain to

estimate the fractional CFO value. The estimation range is limited within±0.5 subcarrier space, and can be expressed as

/ 2 1
*
1, 2,
^
1 / 2
/ 2 1
*
1, 2,
/ 2
Im
1
tan
2
Re
*K*
*k* *k*
*k* *K*
*K*
*F*
*k* *k*
*k* *K*
*R* *R*
*R* *R*
π

## ε

− − =− − =− ⎡_{⋅}⎤ ⎢ ⎥ ⎢ ⎥ = ⎢

_{⋅}⎥ ⎢ ⎥ ⎣ ⎦

### ∑

### ∑

(1)where *R _{1,k}** and

*R*are the pre-defined training symbols in frequency domain.

_{2,k}In WLAN IEEE 802.11a system, similar idea is exploited but different training patterns

are utilized. The estimation of CFO is accomplished by the aid of pre-defined short preamble

and long preamble in time domain and achieves wider estimation range than Moose’s

scattered pilots in DVB-T/H system. The former two data-aided algorithms are both not

suitable solutions for our application.

From section 2.1.2, we can know that the phase of the received signal in time

*domain is rotated by CFO linearly according to the sample time instant tn* as (2-4) shows.

When the difference of sample time instant between two received signals is equal to FFT

*length N, the phase error difference caused by CFO between them can be expressed as *

( ) ( ) 2 2
*l* *n* *N* *l* *n* *ftn N* *ftn*
θ + −θ = Δπ _{+} − Δ π
2πε(*lN _{s}*

*N*

_{g}*n*

*N*) /

*N*2πε(

*lN*

_{s}*N*

_{g}*n*) /

*N*= + + + − + + 2πε 2 (π ε

*ε*

_{I}*) = = + . (2) Since the phase rotation of multiples of 2π can be ignored, the phase error between ( )*

_{F}*l*

*r n and (r n _{l}* +

*N*) is just equal to 2πε

*and in proportion to the fractional CFO value. This phase error feature will be utilized in our proposed fractional CFO synchronization. In*

_{F}the proposed DVB-T/H system platform, however, no any useful training symbol can be used

in time domain. So if we want to exploit the phase error feature between ( )*r n and _{l}*

( )

*l*

*r n*+*N* , the guard interval based algorithm is the most suitable solution.

In order to prevent the influence of multipath channel spread and inter-symbol

interference (ISI), a cyclical prefix is inserted in front of each symbol. The cyclical prefix

must be composed of partial signal in the back of the symbol, and its length has to be longer

or equal to the multipath delay spread as shown in Fig. 2.

Symbol (N) GI(Ng)

copy channel impulse

response

Fig. 2: Guard interval insertion and multipath channel spread

received sample ( )*r n in guard interval and ( _{l}*

*r n*+

_{l}*N*) in the symbol’s tail are exactly identical when there is no any distortion exists such as multipath delay spread or CFO. As

previous sections mentioned, the difference of rotated phase error between ( )*r n and _{l}*

( )

*l*

*r n*+*N* is in proportion to the fractional CFO value ε* _{F}*. We can conclude that the tail
received sample and its cyclical prefix show the same property except for a phase rotation

error which is exactly 2πε* _{F}*. The estimation of fractional CFO value can be accomplished
with the MLE of differential phase between guard interval and the tail of symbol, and can be

expressed as
1 1
2
* *
, , ( , , )
*s* *s*
*F*
*s* *g* *s* *g*
*N* *N*
*j*
*l n N* *l n* *l n N* *l n N*
*n N* *N* *n N* *N*
*x* *r* *r* *r* *r* *e* πε
− −
− − −
= − = −
=

### ∑

⋅ =### ∑

⋅ 1 2 2 ,*s*

*F*

*s*

*g*

*N*

*j*

*l n N*

*n N*

*N*

*e*πε

*r*− − = − =

### ∑

1 * , , ^ 1 1 * , , Im 1 1 arg( ) tan 2 2 Re*s*

*s*

*g*

*s*

*s*

*g*

*N*

*l n N*

*l n*

*n N*

*N*

*N*

*F*

*l n N*

*l n*

*n N*

*N*

*r*

*r*

*x*

*r*

*r*π π

## ε

− − = − − − − = − ⎡ ⎤ ⋅ ⎢ ⎥ ⎢ ⎥ = =_{⎢}

_{⎥}⋅ ⎢ ⎥ ⎢ ⎥ ⎣ ⎦

### ∑

### ∑

(3) (3) shows that the distinguishable phase error of arg( )*x is within*± , so the π

estimation range of the fractional CFO synchronization is also limited within ±0.5 subcarrier space. In the proposed CFO synchronization scheme, the rough estimation of

fractional CFO is calculated with the first symbol after symbol boundary is decided. And then

the estimated fractional CFO value

^

*F*

## ε

will be sent to the CFO compensator before data being sent to FFT receiver as Fig. 1 shows.If AWGN is the only external distortion, the accuracy of the fractional CFO

synchronization will be very excellent because the correlation of guard interval and tail of

symbol can average the noise induced by AWGN. However, the DVB-T/H system is an

multipath channel is necessary. As Fig. 2 shows, the delay spread of multipath channel will

affect the data of the front portion of the guard interval directly especially when the length of

guard interval is relatively short (2k mode, *N _{g}* /

*N*=1/ 32). In order to reduce the effect of multipath delay spread, several beginning samples of the guard interval must be discarded,

and (3) can be rewritten as

1
*
, ,
^
1
1
*
, ,
Im
1 1
arg( ) tan
2 2
Re
*s*
*s* *g*
*s*
*s* *g*
*N*
*l n N* *l n*
*n N* *N* *y*
*N*
*F*
*l n N* *l n*
*n N* *N* *y*
*r* *r*
*x*
*r* *r*
π π

## ε

− − = − + − − − = − + ⎡ ⎤ ⋅ ⎢ ⎥ ⎢ ⎥ = =_{⎢}

_{⎥}⋅ ⎢ ⎥ ⎢ ⎥ ⎣ ⎦

### ∑

### ∑

(4)*where y is the number of discarded samples. However, discarding too many samples will*

also degrade the averaging performance.

**1.2 Integral Carrier Frequency Offset Synchronization **

From previous section, we can know that the time domain guard interval correlation

algorithm can only deal with the rotated phase error caused by the fractional CFO value. The

imperfect effect caused by the integral CFO should be monitored and synchronized in

frequency domain. Thanks to the compensation of

^

*F*

## ε

, the residual fractional CFO ε*is relatively smaller (ε*

_{R}*≤0.02*

_{R}*) and the ICI noise is also neglected. In essence, the k-th*transmitted subcarrier shows up at FFT output bin with subcarrier index

*k*+ as Fig. 2-3 (b) ε

_{I}shows. The subcarrier index shift, which is just equal to the integral CFO ε* _{I}*, must now be
detected by using the pre-defined training sequence (continual and scattered pilots) or the null

subcarriers. In later sections, some different algorithms of integral CFO synchronization will

be illustrated and discussed.

**1.2.1 Conventional Pilots Based Approach **

The DVB-T/H standard defines continual and scattered pilots for synchronization and

equalization in frequency domain. The signal power of the two kinds of pilots is at boosted

continual and scatter piloted is their subcarrier index. The continual pilots locate at fixed

subcarrier index and do not shift as OFDM symbol number increases. However, scattered

pilots are inserted every 12 subcarriers and have an interval of 3 subcarriers in the next

adjacent symbol. In general, the continual pilots based integral CFO synchronization

algorithms are the most widely used because of its good performance in low SNR and mobile

environment. The main idea of this approach is based on the MLE theory. In the first step, the

correlation between two continual pilots at the same subcarrier index for two successive

symbols in the frequency domain based on shifting the pilot positions is calculated, and can

be expressed as
*
1, , ,
*i*
*i* *l* *k* *l k*
*k P*
*C* *R*_{−} *R* *i* *m*
=
=

### ∑

⋅ ≤ (5) where*C*

_{i}*is the correlation value at the i-th shift location,*

*i*

*P =[p*_{1}+*m p*, _{2}+*m*,...,*p _{P}*+

*m*,] are the positions of the subcarriers to be correlated in two

*successive symbols, and m is the estimation range. The integer CFO value*ε

*is then*

_{I}*estimated by detecting the offset position i where the value*

*C is maximized as*

_{i}^

m a x ( * _{i}*)

*I* _{i}*C*

## ε

= (6) Fig. 3 shows the received signal according to the subcarrier in frequency domain whenthe integral CFO is equal to 1 subcarrier space. In DVB-T/H 2k mode, the positions of

continual pilots should be 0, 48, 54, 87…. Accordingly, if the maximum value of *C is _{i}*

obtained from subcarriers 1, 49, 55, 88…, the estimated integral CFO is 1 because the

position of maximum correlation is achieved one subcarrier position away from the original

continual pilots. Because the continual pilots are transmitted at boosted power level, the

power difference of correlation values is still apparent and not affected by strong noise even

in low SNR and deep delay spread channel condition. The total number of multiplication

(2 1) ( 4 2)

*M* = *m*+ ⋅ *P*⋅ + (7)

*where M is the total number of multiplication, and P is the number of correlated pilots, *

respectively. In DVB-T/H system, P is 45, 89, and 177 for 2k, 4k, and 8k mode. Apply (7) we

can see that as the search range increases, if all of the continual pilots are used for estimation,

the total number of multiplication will increase enormously. For example, if the desired

*search range m is 60 for 2k mode when using all continual pilots, the number of *

multiplication will raise up to 22022. For low power consideration, such large number of

multiplication should be avoided. The tradeoff between estimator performance and power

consumption has become an important task for the integral CFO acquisition.

47 48 49
…
... _{…} _{… ……} _{…}
... _{……} _{………} _{…}
53
54 55 87
88
89
47
48 49 53 54 55 87
88 89
+
1
*C*−
+
0
*C*
+
1
*C*
…
…
…
…
…
…
correlation
subcarrier
index
subcarrier
index
(j-1)-th
symbol
j-th
symbol

MAX: estimated CFO=1

continual pilot data

Fig. 3: Received signal in frequency domain when CFO=1 subcarrier space

Besides the continual pilots based approach, another algorithm based on both continual

and scattered pilots (CP+SP) was also proposed. This algorithm calculates the correlation

between possible 4 types of CP+SP patterns with the shifted received symbol in frequency

domain. By detecting the peak value of the correlation result among the 4 CP+SP patterns,

the integral CFO and the scattered pilot mode can be estimated at the same time, and can be

'
^
*
, , ,
1
max , [0,1, 2,3]
*P*
*l z k i* *z k*
*I* _{i}*k*
*R* *Y* *z*

## ε

+ = =### ∑

⋅ ∈ (8)*where P’ is the total number of CP+SP,*

*Y*

_{z k}_{,}

*is the z type CP+SP sequence, and z is the*

subcarrier index pattern of 4 possible types of CP+SP, respectively. Although this approach

can acquire the scattered pilot mode and the integral CFO at the same time, the computational

complexity also rises to about 4 times of the continual pilots based one and leads to more

power consumption.

**1.2.2 Conventional Guard Band Based Approach **

*In DVB-T/H system, the number of subcarriers K within an OFDM symbol is chosen *

*smaller than the symbol length N to provide that so-called “guard bands” at the edges of the *

transmission spectrum are left free. Hence all the subcarriers within guard-bands are

composed of null subcarriers and the transmitted signal power is zero. According to the

DVB-T standard, the signal power of the useful data subcarrier is normalized to 1, and the

power of the reference pilots is 16/9. By exploiting the feature of power difference, a guard

band power detection based algorithm for integral CFO acquisition was proposed by Kim in

1997. This algorithm utilizes the guard bands in both sides of spectrum as a moving window

to search the subcarrier index shift value caused by the integral CFO. The main idea is that

when the useful signal component (data or pilot subcarriers) is not within the moving window,

the total component power within the moving window includes only noise component. So

when the power of the moving window reaches minimum, the shift value of the window is

equal to the shift value of signal spectrum due to the integral CFO, and can be expressed as

max
min
min max
1
1
^
2 2
, ,
min{ },
*K*
*K*
*l k i* *l k i*
*I* _{i}*k K* *w* *k K* *w*
*R* *R* *i* *m*

## ε

− + + + = − = + =### ∑

+### ∑

≤ (9)*where w is the width of the moving window at both sides of the guard band and is set as*

min
*K* *K*max *N*−1
-3
-2
-1
0
1
2
3
i
perfect
symbol
shifted
symbol

min power in this window subcarrier index

. . . _{. . .}_{. . . .}

0

Fig. 4: Received symbol in frequency domain when CFO is -2

Fig. 4 shows the received symbol spectrum in frequency domain according to subcarrier

index when the integral CFO is -2 subcarrier space. As we can see the minimum power

*appears in the moving window where i is -2 because it does not include any data or pilot *

*component. The total number of multiplication M required for the acquisition of integral CFO *

can be expressed as

( 2 ) 4

*M* = *w*+ *m* ⋅ (10)

*From (4), we can find that the number of multiplication M could be reduced effectively *

*by using small moving window width w. However, small w may lead this algorithm to worse *

performance in low SNR and deep frequency selective fading environment. So the trade-off

*between w and M should be treated very carefully. *

In order to improve the performance of the conventional guard band power detection

based algorithm, another modified guard band power detection method was proposed. This

algorithm modifies the structure of the symbol spectrum and inserts additional null

frequency selective fading. However, the modification conflicts with the DVB-T/H standard

and can’t be applied for our system platform.

**1.3. Proposed 2-stage Approach **

From previous sections, we can conclude that neither the continual pilots based

algorithm nor the guard band power detection based algorithm can satisfy good performance

and low computational complexity at the same time. Besides, the number of multiplication of

all these algorithms is in proportion to the search range. If we want to let the integral CFO

estimator work in low SNR and deep frequency selective fading environment and search

large range CFO with low computational complexity, none of these algorithms is the best

choice. In order to solve this problem, a 2-stage integral CFO acquisition algorithm is

proposed as Fug 5 shows. The objective of the first stage is to recognize whether the integral

CFO value ε* _{I}* is positive or negative (i.e. to find whether the direction of subcarrier shift
due to integral CFO is right or left) with a low complexity guard band based algorithm. Once

the first stage finishes and finds the direction of the subcarrier shift, the search range and the

number of multiplication can be reduced half at the same time. In the second stage, the

accurate integral CFO value ε* _{I}* will be acquired along the direction estimated by the first
stage with the proposed continual pilots based algorithm or guard band based algorithm. The

detailed content of the proposed 2-stage approach will be illustrated in later sections.

Recognize orε* _{I}* >0 ε

*<0*

_{I}Stage 1

Stage 2
Find accurate toward leftε*I*

Stage 2

Find accurate toward rightε*I*

0
*I*
ε >
0
*I*
ε <
Acquire ε* _{I}*
Acquire ε

_{I}**1.3.1. The first Stage of the Proposed Approach **

The main task of this stage is to find whether the integral CFO value ε* _{I}* is positive or
negative fast and efficiently, so a left window and a right window that composed of

*w*

_{1}

guard band null subcarriers and *w data subcarriers at the boundary between guard band and *_{1}

data are exploited. In the first step, the summation of signal power of two successive OFDM

symbols based on the position of left and right window is calculated separately. Once the

integral CFO value ε* _{I}* is not equal to zero, the subcarrier distribution of guard band and data
within the left and right window will be imbalanced. So in the second step, we compare the

calculated correlation power to decide whether the integral CFO value ε* _{I}* is positive or
negative, and can be expressed as

2 2
1, ,
*l* *k* *l k*
*k left*
*L* *R*_{−} *R*
=
⎡ ⎤
= _{⎢} + _{⎥}
⎣ ⎦

### ∑

2 2 1, ,*l*

*k*

*l k*

*k right*

*R*

*R*

_{−}

*R*= ⎡ ⎤ =

_{⎢}+

_{⎥}⎣ ⎦

### ∑

,*0*

_{I}*L*>

*R*ε ≤ or

*L*<

*R*,ε

*≥ (11) 0 where left =[*

_{I}*K*

_{min}−

*w K*

_{1},

_{min}−

*w*

_{1}+1,...,

*K*

_{min}−1,

*K*

_{min},

*K*

_{min}+1,...,

*K*

_{min}+

*w*

_{1}−1] , right

max 1 max 1 max max max max 1

[*K* *w* 1,*K* *w*,...,*K* 1,*K* ,*K* 1,...,*K* *w*]

= − + − − + + , and *w is the window *_{1}

width, respectively. data data

### L

### R

Compare L < R, positive L > R, negative Symbol j-1 Symbol j w_{1}w

_{1}w

_{1}w

_{1}K

_{min}K

_{max}GB GB GB GB

Fig. 6: The first stage of the proposed integral CFO estimator

received subcarrier will shift toward right and the number of guard band signal will be more

than that of the data signal in the left window. Also in the right window, the number of the

guard band signal will be less than that of the data signal. The power difference between the

left and right window will appear and help us to decide whether the integral CFO value ε* _{I}*
is positive or negative. The total number of multiplication of the first stage can be expressed

as

1

8

*M* = ⋅ (12) *w*

From (12) we can know that the number of multiplication of the first stage is not

*affected by the estimation range m and low complexity calculation can be achieved by *

choosing smaller window width. However, too small window width will affect the

performance of the first stage. The optimal window width will be shown by simulation result

in chapter 3.

**1.3.2. The Second Stage of the Proposed Approach **

By the aid of the first stage, the search range of the second stage can be reduced from

*m*

± to *m*. However, the result of the first stage may be incorrect while the integral CFO
value ε* _{I}* is smaller than the window width

*w in deep frequency selective fading channel*

_{1}

environment. In order to prevent estimation error, the search range should be extended from

m to *m+ , implying that we should add more ww*_{1} *1* points to the search range toward the

reverse direction to assure correct acquisition result when the integral CFO value ε* _{I}* is near
zero in deep frequency selective fading channel.

Once the search range of the second stage is decided, there are still various algorithms

can be applied for acquisition the accurate integral CFO value ε* _{I}*. The trade-off between
estimator performance and computational complexity, however, still exists among the

previous mentioned algorithms. Considering acceptable acquisition performance and efficient

computation load, a reduced continual pilot based algorithm and a guard band power

illustrated in later sections.

**(a) Proposed Reduced Continual Pilot Based Approach **

From (7), we can find that the number of multiplication of the conventional continual

pilot based approach is in proportion to not only the search range *m but also the number of *

utilized continual pilot *P. In order to achieve efficient computational load, the number of *

utilized continual pilot should be reduced with the search range at the same time. Hence a

reduced continual pilot based approach is proposed. The main feature of the proposed

reduced continual pilot based algorithm for the second stage integral CFO acquisition is

similar to the conventional continual pilot based one. But the proposed one exploits only a

part of the continual pilot instead of all of them to reduced the number of multiplication, and

can be expressed as
,
^
*
1, ,
max
*r i*
*l* *k* *l k*
*I* _{i}*k P*
*R* *R*

## ε

− = =### ∑

⋅ (13) where*P is the shifted subcarrier index of the selected continual pilots,*

_{r i}_{,}− ≤ ≤

*m*

*i*

*w*

_{1}

while negative value estimated by the first stage, and − ≤ ≤ while positive value, *w*_{1} *i* *m*

respectively. The number of multiplication for the proposed reduced continual pilot approach

can be expressed as

1

( 1) ( * _{r}* 4 2)

*M* = *m*+*w* + ⋅ *P* ⋅ + (14)

where *P is the total number of the correlated continual pilots. Because the power _{r}*

difference between pilot and data subcarrier is very significant, it is not necessary to use all of

the continual pilots and the acquisition performance is still acceptable to meet lower

computational load. As (14) shows, the number of multiplication can be reduced effectively.

**(b) Proposed Guard Band Power Detection Based Approach **

As previous sections mentioned, the conventional guard band power detection based

algorithm requires fewer number of multiplication and performs worse performance in low

symbol. In order to utilize the advantage of lower computational complexity and to improve

the performance in critical channel condition, we propose a new guard band power detection

based algorithm. By the aid of the proposed first stage, the search range of the second stage

can be reduced effectively and more OFDM symbols can be utilized to improve the

acquisition performance. Thus the proposed guard band power detection based algorithm still

keeps the moving window scheme and calculates the summation of signal power within three

successive OFDM symbols, and can be expressed as

max
min
min 2 max 2
1
1
^ _{2} _{2} _{2} _{2} _{2} _{2}
, 1, 2, , 1, 2,
min{ }
*K*
*K*
*l k i* *l* *k i* *l* *k i* *l k i* *l* *k i* *l* *k i*
*I* _{i}*k K* *w* *k K* *w*
*R* *R* *R* *R* *R* *R*

### ε

− + − + − + + + − + − + = − = + ⎡ ⎤ ⎡ ⎤ =### ∑

_{⎢}

_{⎣}+ +

_{⎥}

_{⎦}+

### ∑

_{⎢}

_{⎣}+ +

_{⎥}

_{⎦}(15) where

*w is the width of the moving window at both sides of the guard band,*

_{2}

1

*m* *i* *w*

− ≤ ≤ while negative value estimated by the first stage, and − ≤ ≤*w*_{1} *i* *m* while
positive value, respectively. As Fig. 2.10 shows, by the use of summation within three

successive OFDM symbols, the distortion induced by noise in severe environment can be

decreased effectively. The number of multiplication can be expressed as

2 1

( ) 12

*M* = *w* + +*m* *w* ⋅ (16)

Compared (16) with (10), we can see that the total number of multiplication of the

proposed guard band power detection based approach consumes about 1.5 times of that of the

conventional approach. However, the acquisition performance is improved significantly.

GB data
GB data GB
GB
w_{2}
GB data
GB
w_{2}
GB Moving Wondow
Symbol j-1
Symbol j
Symbol j-2
max
*K*
min
*K*
GB

**1.4. Residual Carrier Frequency Offset Synchronization **

After the acquisition stage estimates the integral and most of the fractional CFO value,

the residual CFO value is usually less than 1 to 2 percent of the subcarrier space. However,

the phase error induced by such small value of CFO in time domain still affects the system

performance for long time receiving operation. As Fig. 2.2 shows, the accumulative phase

error when residual CFO value is 0.01 still exceeds π while the received number of data is more than 10,000. Besides, the Doppler effect in mobile environment also introduces small

drift to CFO. Therefore the tracking of residual CFO is necessary and has to operate

continuously until the reception is turned off.

Generally speaking, the residual CFO value ε* _{R}* is usually very small. Thus only
fractional CFO estimation is sufficient. In particular, the estimation of the residual CFO at

tracking stage requires precise and low variation result. Therefore in our DVB-T/H system

platform, the tracking stage of CFO is divided into two parts. The first part estimates the

residual CFO value symbol by symbol followed by a PI (proportional-integral) loop filter to

reduce the variation. The tracking loop of the CFO synchronization is shown in Fig. 8.

### FFT

### CFO

### Compensator

### PI loop

### filter

### data

### CFO

### from

### ADC

### to

### EQ

### One-shot

### residual CFO

### estimator

1*e*^ 1

*e*~ 1

*e*^ ^

*I*

*F*

### ε ε

+ 2*e*ε

Fig. 8: The tracking loop of the CFO synchronization

As shown in Fig. 2.11, *e is the residual CFO value of the first iteration of the tracking *_{1}

loop. After the estimation of *e , the output of the residual CFO estimator *_{1}

^

1

post-processed by the PI loop filter. When the second iteration starts, the CFO compensator

will compensate the incoming data with the updated CFO value

^ ^ ~

1

*I* *F* *e*

## ε ε

+ + and then get the next residual CFO error*e of the second iteration. As the CFO tracking loop works*

_{2}

iteratively, the residual CFO error will be minimized.

**1.4.1. Residual CFO Estimation **

The objective of the residual CFO estimator is to estimate the residual CFO error value

precisely and fast. As previous section mentioned, only fractional CFO synchronization is

sufficient for this estimator. Considering hardware integration and resource reuse, the

fractional CFO estimator may can be utilized for the residual CFO estimator. However, the

non-data-aided algorithm that exploits the guard interval is very sensitive to the inter-symbol

interference introduced by the multipath delay spread and the estimated result may be not

precise enough for the residual CFO estimation in deep delay fading environment. Only

roughly fractional CFO value can be obtained with this approach. Therefore an efficient

data-aided algorithm that employs the pre-defined continual pilots is applied for the residual

CFO estimator.

After most of the CFO value is estimated and compensated, the residual CFO value is

usually less than 1 to 2 percent of the subcarrier space and the ICI noise is small enough to be

neglected. As (2-3) shows, regardless of the ICI term, the phase error caused by the residual

CFO error and SCO at the *k-th subcarrier of the l-th OFDM symbol in frequency domain can *

be expressed as
2
( ) 2 ( )(1 ) / ( ) ( )
*l* *R* *s* *g* *s* *g* *l*
*k*
*k* *lN* *N* *N* *lN* *N* *k*
*N*
π
ϕ = πε + +ζ + + ζ φ+ (17)
where ( )φ_{l}*k* is the phase of the channel frequency response *H _{l k}*

_{,}. If the channel is a slowly fading channel (φ

*( )*

_{l}*k*≈φ

_{l}_{−}

_{1}( )

*k*), the difference of phase rotation between two successive OFDM symbols is represented as

1

' ( )_{l}*k* * _{l}*( )

*k*

*( )*

_{l}*k*

2 * _{R}N_{s}* 2

*2*

_{R}N_{s}*kN*

_{s}*N*

*N*

*N*πε πε ζ π ζ = + + 2

*2*

_{R}N_{s}*kN*

_{s}*N*

*N*πε π ζ ≈ + (18)

The second term

2 _{R}N_{s}*N*

πε ζ

can be ignored since the product of ε ζ*R*⋅ _{ is usually less }

than 2.0x10-6. From (2-24) we can know that the residual CFO ε*R*_{ causes mean phase error }

and the SCO ζ causes linear phase offset between two consecutive OFDM symbols. If we take two adjacent continual pilots of arbitrary two consecutively received OFDM symbols,

the phase rotation is shown in Fig. 2.12 [17]. The total phase rotation includes the effects of

symbol timing offset, residual CFO and SCO. As we can see from Fig. 2.12, the magnitude of

phase rotation induced by symbol timing offset is identical and in proportion to the subcarrier

index among the two symbols. However, in the current symbol, the effect of residual CFO

and SCO are accumulated in the phase of the previous symbol, where the residual CFO

induces mean phase and SCO generates linear phase. Thus, we must estimate the residual

CFO as well as the SCO by computing the phase rotation between two successive symbols.

**Adjacent Continual Pilot**

**Symbol timing offset**
**CFO**

**Sampling clock offset**

**Previous symbol**
**Current symbol**

**subcarrier index**

Fig. 9: Phase rotation between two successive OFDM symbols

continual pilots which have fixed subcarrier index are exploited to estimate the residual CFO.

In general, the residual CFO and the SCO are estimated jointly because their effects of phase

rotation are uncorrelated. Thus a joint residual CFO and SCO estimation algorithm is applied

as
^
2, 1,
1 1
( )
2 (1 / ) 2 *l* *l*
*R*
*g*
*N* *N* ϕ ϕ
π

## ε

= ⋅ ⋅ + + ^ 2, 1, 1 1 ( ) 2 (1*N*/

_{g}*N*)

*K*/ 2

*l*

*l*ζ ϕ ϕ π = ⋅ ⋅ + + 1|2 * 1|2,

*l*arg[

*l k*,

*l*1,

*k*]

*k C*

*R*

*R*ϕ

_{−}∈ =

### ∑

⋅ (19) where*C denotes the subcarrier index set of continual pilots which locates in the left*

_{1}

half (*k*∈[0, (*K*−1) / 2)), and *C denotes the subcarrier index set of continual pilots which *_{2}

locates in the right half (*k*∈((*K*−1) / 2,*K*_{max}]) of the OFDM symbol spectrum, respectively.
Applying correlation of continual pilots within two successive OFDM symbols and

accumulating the correlation results in two parts lead to the so-called CFD/SFD (carrier

frequency detector / sampling frequency detector) algorithm [18]. The summation of ϕ* _{2,l}*
and ϕ

*can compute mean phase error while subtraction of ϕ*

_{1,l}*and ϕ*

_{2,l}*produces the linear phase error. As a result, the residual CFO and SCO can be estimated jointly by*

_{1,l}multiplying different coefficients.

Besides the continual pilots based approach, some other scattered pilots based

approaches are also presented in [14] and [19]. [14] proposes a residual CFO estimator that

*exploits the continual and scattered pilots between the l-th and the (l-4)-th OFDM symbol. *

The equation of this approach is very similar to (2-25) except the correlated symbols and

pilots. The main feature of this algorithm is to use more pilots to reduce the distortion caused

by AWGN and ICI noise. However, the convergence speed is extended about 2.5 times

*longer than that of the CFD/SFD algorithm because it utilizes the l-th and the (l-4)-th OFDM *

successive OFDM symbols and has similar equation with (2-25). However, the subcarrier

index of scattered pilots of two successive OFDM symbols is not identical and has a

difference of 3. The estimated phase error between two scattered pilots is also distorted by the

symbol timing offset. However, the symbol timing offset is an unknown factor and can not be

estimated precisely by symbol synchronizer. So the estimation result of this approach is not

reliable only if precise symbol offset value is estimated.

**1.4.2. Residual CFO Tracking Loop Filter **

In order to reduce the variation of the estimated residual CFO, a PI loop filter is utilized

in our CFO synchronization design [20]. The PI loop filter is composed of two paths. The

proportional path multiplies the estimated residual CFO by a proportional factor *K . The _{P}*

integral path multiplies the estimated residual CFO by an integral factor *K and then _{I}*

integrates the scaled value by using an adder and a delay element. The block diagram of the

PI loop filter is shown as Fig. 10.

K_{P}

K_{I}

### +

Z-1### +

Fig. 2.10: Block diagram of PI loop filter

The transform function of the PI loop filter can be represented as

1
1
( )
1
*P* *I*
*Z*
*H z* *K* *K*
*Z*
−
−
= +
− (20)
For small loop delay and *K _{I}* −

*K*

_{P}*K*, the standard deviation of the steady-state 1 tracking error is expressed as

_{P}( ')*e* *K _{P}*/ 2 ( )

*e*

σ = ⋅σ (21)
*where e is the estimation error of the residual CFO estimator and e’ is the steady-state *

tracking error. The close-loop tracking time constant is approximately given by

1/

*loop* *P*

*T* ≈ *K* (22)
So from (21) and (22) we can find that there is a tradeoff between steady-state tracking

error and tracking convergence speed. In our proposed DVB-T/H platform, the loop

*parameter KP* is chosen as a larger value to increase the convergence speed in the beginning

of tracking, and then switched to a smaller value to reduce the steady-state tracking error

**Part II: A Single Chip DVB-T/H baseband receiver design **

A DVB-T/H baseband receiver with 2k/4k/8k-point FFT, complete synchronization,

channel equalizer, and channel decoder is implemented with developed designs, such as

2K/4K/8K FFT processor, Path Merging Viterbi decoder, Single Memory De-interleaver, and

RS Decoder. This baseband receiver achieves 70Hz Doppler effect tolerance with multiple

steps CFO compensation, 2D linear channel equalizer in 2k mode. The chip with single port

154Kbytes embedded SRAM only consumes 250mW for highest 31.67Mb/s data rate.

**I. Introduction **

In conventional approaches for DVB-T receiver, they are partial functional design

[8][10][12][13] or non-fully baseband supporting design [11][14], or multiple chip design

approach [3][4][5][7][9]. There is no single design for DVB-T and DVB-H baseband receiver,

except Fechtel’s design [6], but his approach is design for simpler channel environment.

These proposed designs have optimized for functional blocks or partial system. In this paper,

we present one DVB-T/H fully baseband receiver, which included two synchronization

systems, FFT core, equalization, QAM demodulation, and FEC decoders. In following paper

organization, we will introduce proposed system architecture in section II. The detail

architecture of functional blocks will be described in section III. The simulation result,

estimation result and chip photo will be shown in section IV. In the end, we will discuss

conclusion and future work in the section V.

The existing DVB-T receivers are partial functional design [8][10][12][13], or multiple

chip design approach [3][4][5][7][9], otherwise the processor/DSP based approaches [LSI

Logic, L64782] were proposed in past few years. The DVB-H system is based on DVB-T

system and modified for handheld applications.

**II. System Architecture **

In this report, we introduce one single chip DVB-T/H [1][2] baseband receiver, which

demodulation, inner de-interleaver, Viterbi decoder, outer deinterelaver and RS decoder. The

system block diagram is shown in Fig. 1.

Fig. 1: Block diagram of DVB-T/H baseband receiver

In system organization, we separate this DVB-T/H baseband receiver into two main

sections: inner receiver (synchronization/demodulation) and outer receiver (FEC). In the

synchronization systems of inner receiver, there are two sub-systems to perform

synchronization operations, one is timing synchronization system, and another one is

frequency synchronization system. These two subsystems are allocated in pre-FFT

synchronizations and post-FFT synchronizations. In timing synchronization system, there are

two target to optimized, first one is OFDM symbol bound detection, second one is sampling

clock offset (SCO) value. In frequency synchronization system, we have to reduce the carry

frequency offset (CFO) value by three estimation processes: fractional part of CFO

estimation, integral part of CFO estimation, and residue CFO tracking.

In channel equalizer, we are using 2 1D linear interpolators [6] in channel estimation for

mobile environment issue, and zero-forcing method is used in channel equalization. The

structure of channel estimation and channel equalization is shown in Fig. 3.

Inner Deinterleaver Inner Deinterleaver Outer Deinterleaver Outer Deinterleaver Outer Decoder Outer Decoder Inner Decoder Inner Decoder De-scrambler De-scrambler FFT Window FFT Window Pre-FFT AFC Pre-FFT AFC Carrier Frequency Compensator Carrier Frequency Compensator Post-FFT AFC Post-FFT AFC Sampling Frequency Tracking Sampling Frequency Tracking Carrier Frequency Tracking Carrier Frequency Tracking Fine Symbol Synchronization Fine Symbol Synchronization Equalizer Equalizer GI/Mode decision GI/Mode decision Coarse Symbol Synchronization Coarse Symbol Synchronization Interpolator/ Decimation FFT Core FFT Core SP Mode Detection SP Mode Detection QAM Demodulation QAM Demodulation TPS decode & Pilot Remove TPS decode & Pilot Remove Rx RF ADC DVB-T/H baseband receiver Inner Deinterleaver Inner Deinterleaver Outer Deinterleaver Outer Deinterleaver Outer Decoder Outer Decoder Inner Decoder Inner Decoder De-scrambler De-scrambler FFT Window FFT Window Pre-FFT AFC Pre-FFT AFC Carrier Frequency Compensator Carrier Frequency Compensator Post-FFT AFC Post-FFT AFC Sampling Frequency Tracking Sampling Frequency Tracking Carrier Frequency Tracking Carrier Frequency Tracking Fine Symbol Synchronization Fine Symbol Synchronization Equalizer Equalizer GI/Mode decision GI/Mode decision Coarse Symbol Synchronization Coarse Symbol Synchronization Interpolator/ Decimation FFT Core FFT Core SP Mode Detection SP Mode Detection QAM Demodulation QAM Demodulation TPS decode & Pilot Remove TPS decode & Pilot Remove Rx RF ADC DVB-T/H baseband receiver

In the end of inner receiver, there are two operations: QAM demodulation, and inner

De-interleaving. The soft-decision QAM demodulation is used to improve performance gain

in Inner decoder (Viterbi decoder). In inner de-interleaving, we exchange the symbol

de-interleaving operation order before QAM demodulation that can reduce the data overhead

when the soft-decision QAM demodulation result word-length is longer than word-length of

channel equalization output.

A 64 64 states ACSs structure and path merge method is impelemted in Viterbi decoder,

which can reduce feedback timing and memory access time. For Outer DeInterleaver, we

propose one address generator with universal memory structure, which improved the memory

efficiency and minimize the required memory size. The RS (204, 188) decoder is assembling

by several modules: Syndrome Calculator, Key Equation Solver, Chien Search, Error Value

Evaluator, and Error Corrector. In the output of RS decoder, last stage of receiver, the

Descrambler decodes the scrambling data but not including the synchronization word in

original data stream.

**III. Functional block architecture **

Two main regions organized the proposed DVB-T/H receiver, inner receiver and outer

receiver. The inner receiver performs synchronization process, FFT operation, channel

equalization, QAM demodulation, and inner deinterleaver. The block diagram of inner

receiver is shown in Fig. 2. The outer receiver contents inner decoder (Viterbi decoder), outer

deinterleaver, outer decoder (RS decoder), and Descrambler. The block diagram of outer

receiver is shown in Fig. 3. We will introduce detail of inner receiver and outer receiver in

the two following sub-sections separately.

**1. Functional blocks of inner receiver **

The proposed inner receiver is working firstly after system reset signal triggered. When

the receiving data arriving, timing synchronization system will detect the key frame

Guard Interval Ratio (1/4, 1/8, 1/16, 1/32). These two parameters are main operation

parameters of inner receiver, without operation mode and guard interval length (ratio), The

following functional blocks can’t work correctly. After detected operation mode, guard

interval ratio, the coarse symbol bound will be decided by normalize maximum correlation

method. In the same time, the fraction part of CFO will be estimated based on the phase

rotation between guard interval data and partial of symbol data.

Fig. 2: Inner receiver architecture for DVB-T/H system

The FFT core will do Fast Fourrier Transform when receiverd complete OFDM symbol.

Because the require operation clock cycle count of FFT core is almost 3 times of sample

count of one symbol, the FFT core will operate at 4 times of sampling clock rate. The FFT

core is based on radix-8 butterfly unit with 64 points pre-fetch buffer, and using dynamic

scaling method to reduce output word length overhead.

The scatter pilot (SP) order detection and post-FFT CFO estimation will start after FFT

output data. These two functions will spend 3 OFDM symbols to detect SP order and integral

part of CFO. The CFO compensator will compensate the FFT input data to reduce the ICI

effect when the integral part of CFO ready. After detecting SP order, the channel estimator

will extract the SP information in the OFDM symbol. To getting correct, acceptable Channel

Frequency Respond (CFR), the channel estimator will queue four OFDM symbols to get 2-
Channel
Estimator
Channel
Equalizer
Symbol
De-Interleaver
QAM
demaper
Bitwise
memory
P/S
4xÆ6x
CFO
Compensation
Time Sync
CFO fractional
Estimation
FFT
memory
FFT
Core
FFT
memory
Symbol
memory
Symbol
memory
CFO integral
Estimation
CFO Tracking
**Data**
**input**
TPS decoder
CE
memory
CE
memory
CE
memory
CE
memory
CE
memory
CE
memory
Bitwise
memory
Bitwise
memory
Bitwise
memory
Bitwise
memory
Bitwise
memory
Control
unit
**8**
**6**
**Viterbi input**
**36**
**24**

dimension CFR information. After channel estimator collected sufficient CFR information,

the channel equalizer will equalize data and output to symbol deinterleaver memory. The

detail structure of channel estimation and channel equalizer is shown in Fig. 3.

Fig. 3: Architecture of 2D linear channel equalizer

Before symbol deinterleaving operation, we have to decode Transmission Parameter

Signal (TPS), which including operation mode, QAM modulation method, guard interval

ratio, symbol interleaving order, and coding rate of inner decoder. Without decoded TPS

information, the inner deinterleaver, QAM receiver, and Outer receiver can’t work directly.

The complete TPS information is embedded in one OFDM frame (68 OFDM symbols) of

DVB-T/H frame structure. So we have to waiting at least one OFDM frame to collect

complete TPS code and decode TPS by DeBPSK modulation method. When the TPS

information is ready, the Symbol deinterleaving, QAM demodulation, and bitwise

deinterleaving will start working. For each bitwise interleaving section, symbol deinterleaver

will output 126 deinterleved data to 64-level Soft decision QAM receiver. The QAM receiver

output the demodulated data and write into bitwise deinterleaving memory. After each 126

demodulated data fill into bitwise interleaving memory. The Symbol deinterleaving, QAM

receiver will hold until bitwise deinterleaver transfer one complete section data.

Size 6817 RAM Size 6817 RAM Size 6817 RAM Size 3424 RAM Size 3424 RAM Serial In Pilots STORAGE Size 2288 RAM 1x 2x 3x 1x 2x 3x + 1x 2x 1x 2x +

DIV Serial Out

Size 6817 RAM Size 6817 RAM Size 6817 RAM Size 3424 RAM Size 3424 RAM Serial In Pilots STORAGE Size 2288 RAM 1x 2x 3x 1x 2x 3x + 1x 2x 1x 2x +

DIV Serial Out

**2. Functional blocks of outer receiver **

The detail block diagram of outer receiver is shown in Fig. 4.After bitwise

deinterleaving, the Virterbi decoder will receive bitwise deinterleaver output to decode

puntched convolution code. The outer de-interleaver is universal memory structure with

specified address generator. The required memory space in the universal memory structure

can be reduced to minimum size, which is depended on RS(204,188) decoding length. There

are 5 steps in the RS(204,188) decoder, Syndrom Calculator, Key Equation Solver, Chien

Search, Error Value Evaluator, and Error Corrector. At the output of RS decoder, the

Descrambler decode the scrambeled data stream expect the synchronization words.

Fig. 4: FEC architecture for DVB-T/H system

**IV. Simulation and implementation **

The chip implementation is using 0.18um CMOS process with die size is 6.9x5.8 mm2 including IO pad and using 208-pin CQFP package. The chip simulated post-layout power

consumption is 250mw@31.67Mbps of maximum data rate of DVB-T/H system. The real

chip measurement and testing procedure are complete. The Fig. 5 shows the power profile of

simulation, and measurement result. The comparisons with existing design are shown in

branch metric 64 ACS 64 Path metric Survivor memory Universal Outer DeInterleaver Memory Outer DeInterleaver address generator Syndrome Calculator Key Equation Solver Chien Search Error Value Evaluator Error Corrector Descrambler MPEG-2 Stream Bitwse deinterleaving output

Table 1. The detail information, advanced of proposed design will highlight in the

comparison table. The chip photo and supporting system specification are shown in Fig. 6.

0 50 100 150 200 250

Sync. Stage, simulated Sync. Stage, measurement receiving out stage, simulated receiving out stage,

measurement Synchronization FFT core Memory access FECs others overall

Fig. 5: Power consumption profile

Table 1: The Comparison with existing designs

Proposed Jheng‘s[14] Hosemann’s[11] LSI L64782[15]

Technology 0.18um CMOS ASIC 0.18um CMOS ASIC 0.13um CMOS

AS-DSP -

Input clock 109.71M Hz 54.86M Hz *1 250M Hz -

Power 250mW 307mW 300mW 800mW *2

Memory size 1263.4K bits - 1180K bits -

Die size 40.02 mm2 15.6 mm2 9.7 mm2 -

Feature

All functional block included, except ADC Low power

consumption

1D CE method, include FEC blocks Without sync. func, and ADC

Without sync. func. DSP based approach Doesn’t support DVB-H yet Fully baseband integrated with embedded 10bit ADC Doesn’t support DVB-H yet *1: nominal frequency

Fig. 6: chip result

**V. Conclusion and future work **

In related research publish; there is no single chip, DVB-T/H fully baseband receiver

design which including synchronization, demodulation, and channel decodeing. Otherwise,

even in published single chip design, the system environment constrain is simpler such as the

design may not meet the DVB-T/H required system performance. In this paper, we present

one single chip DVB-T/H baseband receiver with 1.8V simulated 250mW average power

consumption for 31.67 Mbps output data rate. Since there are several COFDM applications

announce in the world, for example: ISDB, DMB and DVB systems, and they have similar

system architecture or frame structure; so that the design strategy, algorithm approach may be

reused for these systems. Base on current research result, developing the low power,

universal COFDM processor for multiple COFDM system is our next target.

**Reference **

[1] ETSI EN 300 744 V1.5.1, “Digital Video Broadcasting (DVB): Framing structure, channel coding and modulation for digital terrestrial television”, ETSI, Nov. 2004.

[2] ETSI EN 302 304 V1.1.1, “Digital Video Broadcasting (DVB): Transmission System for Handheld Terminals (DVB-H)”, ETSI, Nov. 2004.

[3] Anikhindi, S., et al., “A commercial DVB-T receiver chipset”, Broadcasting Convention, 1997. International, 12-16 Sept. 1997 Page(s):528 – 533

**1/2, 2/3, 3/4, 5/ 6, 7/8**
**Guar d Interval rat io**

**QPSK, 16QA M, 64QA M**
**Modulatio n**
**2k, 4k, 8k**
**Operatio n mo de**
**DV B- T/ DV B- H**
**Supporting Standard**
**250mw @31.67Mbps ***

**Po wer Co nsumpt ion**

**1.8V Core, 3.3V I/ O**
**Supply Voltage**

**109.71 MHz**
**Inp ut Cloc k Speed**

**6.9 X 5.8 mm2**

**die S ize**

**208-p in CQFP**
**Pac kage**

**154.2 Kbytes**
**Embe dded Me mory S ize**

**371,353**
**Logic Gate Co unt**

**(Exc luding SRA M)**

**UMC 0.18um CMOS, 1P6M**
**Tec hnique**

**1/2, 2/3, 3/4, 5/ 6, 7/8**
**Guar d Interval rat io**

**QPSK, 16QA M, 64QA M**
**Modulatio n**
**2k, 4k, 8k**
**Operatio n mo de**
**DV B- T/ DV B- H**
**Supporting Standard**
**250mw @31.67Mbps ***

**Po wer Co nsumpt ion**

**1.8V Core, 3.3V I/ O**
**Supply Voltage**

**109.71 MHz**
**Inp ut Cloc k Speed**

**6.9 X 5.8 mm2**

**die S ize**

**208-p in CQFP**
**Pac kage**

**154.2 Kbytes**
**Embe dded Me mory S ize**

**371,353**
**Logic Gate Co unt**

**(Exc luding SRA M)**

**UMC 0.18um CMOS, 1P6M**
**Tec hnique**
**FFT**
**RAM**
**FFT**
**CE**
**RAM**
**Viterbi**
**Decoder**
**RS**

**Decoder** **ViterbiRAM**

**CE**
**EQ**
**(I)**
**Inner**
**Deinterleaver**
**EQ**
**(II)**
**DeQAM**
**CFO**

**(I)** **CFO(II)**

**CFO**
**(III)**
**Time (I)**
**Tim**
**e**
**(II)**
**Ot**
**hers**

[4] Makowitz, R., et al., “DVB-T Decoder ICs”, Consumer Electronics, IEEE Transactions on Volume 43, Issue 3, Aug. 1997 Page(s):438 – 442

[5] Tognin, M., et al., “A VLSI solution for a digital terrestrial TV (DVB-T) receiver”, Broadcasting Convention, 1997. International 12-16 Sept. 1997 Page(s):343 – 348

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[8] Frescura, F., et al., “DSP based OFDM receiver and equalizer for professional DVB-T receivers”, Broadcasting, IEEE Transactions on Volume 45, Issue 3, Sept. 1999 Page(s):323 – 332

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[14] Kai-Yuan Jheng, et al., “A DVB-T baseband receiver design based on multimode silicon IPs”, VLSI Design, Automation and Test, 2005. (VLSI-TSA-DAT). 2005 IEEE VLSI-TSA International Symposium on 27-29 April 2005 Page(s):49 – 52